JP2009291019A - Controller for inverter for ac motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce current ripple without unnecessarily increasing a switching loss in an inverter. <P>SOLUTION: A controller 20 for an inverter 16 for an AC motor, which converts DC voltage supplied from a power supply to AC voltage to apply it to an AC motor M, is equipped with a switching signal generating section 58 which generates a switching signal for a switching element in the inverter 16 based on a comparison result between the carrier and a voltage command value and then outputs the switching signal to the inverter 16; and a carrier frequency controlling section 64 for controlling the frequency of the carrier in the inverter 16 according to the voltage command value. The carrier frequency controlling section 64 includes a carrier frequency switching section which sets the carrier frequency high in a region wherein the voltage command value is large and sets the carrier frequency low in a region wherein the voltage command value is small within one control period of the inverter 16. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a control device that controls an inverter for an AC motor used as a traveling motor of an electric vehicle, for example.

  Conventionally, in an electric vehicle such as a hybrid vehicle or an electric vehicle, a DC power supplied from a power source is converted into an AC voltage by an inverter and applied to an AC motor that is a traveling motor, thereby driving the AC motor for traveling. Power is output.

  Generally, in a control device that controls an inverter that performs DC / AC conversion, PWM (Pulse Width Modulation) control is executed. A specific example will be described with reference to FIG. 10A. In an inverter control device including a plurality of switching elements (for example, IGBTs), a sinusoidal voltage command value 1 and a high-frequency carrier or carrier (generally, Compared with (triangular wave) 2, a rectangular pulse in which the region where the voltage command value is equal to or larger than the absolute value of the carrier is an on period, and a large number of PWM pulses in which the on period is increased or decreased at a constant period is generated. The PWM pulse is input to the inverter as a switching signal and the switching element is turned on / off so that the fundamental voltage component of the direct-current voltage input to the inverter is converted into an approximately sinusoidal AC voltage. It has become. In the example shown in FIG. 10A, since the voltage command value is compared with the absolute value of the carrier, two PWM pulses are generated in one cycle of the carrier.

  When the AC voltage converted as described above is applied, a motor current (actual current waveform) 4 flows through the stator coil in the AC motor. Here, the motor current 4 has a phase in which the electrical angle is delayed by 90 degrees due to the influence of the inductance of the stator coil with respect to the motor input voltage having the same phase as the voltage command value 1.

  Since the actual AC voltage output from the inverter does not have a smooth sine wave shape and includes minute fluctuations or ripples, a current ripple 5 is also generated in the motor current 4 as shown in FIG. 10A. When the current ripple 5 is large, iron loss and electromagnetic noise in the AC motor increase, and the motor operating efficiency decreases.

  For example, in Patent Document 1, in a VVVF (variable voltage variable frequency) inverter, the magnitude of a current ripple is detected using a bandpass filter. When the current ripple is large, the carrier frequency is increased, and when the current ripple is small, the carrier frequency is increased. It is described that current noise is reduced and electromagnetic noise is reduced by controlling to lower.

  In addition, in Patent Document 2, in an inverter for driving a brassless DC DC motor, the carrier frequency is lowered when the duty ratio of the PWM pulse is small when the torque is small and the PWM pulse is It is described that the carrier frequency is increased when the duty ratio is large.

JP-A-6-178550 JP 2005-94875 A

  However, neither of Patent Documents 1 and 2 considers the magnitude of the current ripple within one control period of the inverter at all. Therefore, if the carrier frequency is increased uniformly (or macroscopically), the one control period The carrier frequency rises even in the time domain or electrical angle domain where the current ripple is small, thereby unnecessarily increasing the switching loss in the inverter.

  Here, specifically explaining with reference to FIG. 10B, it is clear that the current ripple 5 is relatively small in the example of FIG. 10B in which the carrier frequency is doubled compared to FIG. 10A. However, since the carrier frequency is doubled over the entire range of electrical angle 360 degrees that is one control cycle, the magnitude of current ripple 5 does not become a problem so much, for example, the range of electrical angle 180 degrees ± 30 degrees Then, the problem of switching loss in the inverter becomes larger than the current ripple.

  An object of the present invention is to reduce the current ripple without unnecessarily increasing the switching loss in the inverter, thereby reducing the iron loss and electromagnetic noise in the AC motor and satisfactorily improving the motor efficiency. It is to provide a control device for an inverter.

  The present invention relates to a control device for an inverter for an AC motor capable of converting a DC voltage supplied from a power source to be applied to an AC motor into an AC voltage by switching control of a plurality of switching elements, the carrier having a predetermined waveform and an external A switching signal generator that generates a switching signal of the switching element based on a comparison with a voltage command value input from the inverter and outputs the switching signal to the inverter, and controls the frequency of the carrier in the inverter according to the voltage command value A carrier frequency control unit configured to set a carrier frequency high in a region where the voltage command value is large and a carrier frequency in a region where the voltage command value is small within one control cycle of the inverter. It includes a carrier frequency switching unit that is set low.

  In the AC motor inverter control device according to the present invention, the carrier frequency control unit includes a duty ratio calculation unit that calculates a duty ratio for each switching signal generated by the switching signal generation unit, and a duty ratio calculation unit. A duty ratio determination unit that determines whether to perform carrier frequency switching by comparing an input duty ratio with a predetermined threshold, and the carrier frequency switching unit receives a determination result by the duty ratio determination unit A carrier frequency switching signal may be output to the switching signal generator.

  In addition, the electric vehicle according to the present invention is applied by converting the DC voltage supplied from the inverter and the inverter controlled by the AC motor inverter control device according to the present invention into an AC voltage by the inverter, And an AC motor capable of outputting driving power.

  According to the present invention, within one control cycle of the inverter, the carrier frequency is set high in a region where the voltage command value is large, and the carrier frequency is set low in a region where the voltage command value is small. It can be avoided that the carrier frequency is set high in a region where is originally small. As a result, while suppressing an unnecessary increase in switching loss of the inverter, iron loss and electromagnetic noise in the AC motor can be reduced to improve the motor efficiency.

  Embodiments according to the present invention will be described below in detail with reference to the accompanying drawings. In this description, specific shapes, materials, numerical values, directions, and the like are examples for facilitating the understanding of the present invention, and can be appropriately changed according to the application, purpose, specification, and the like. Moreover, although the apparatus of the inverter for alternating current motors in this embodiment is demonstrated as what is used about the inverter which drive-controls the driving | running | working AC motor of electric vehicles, such as a hybrid vehicle and an electric vehicle, the inverter for alternating current motors which concerns on this invention This apparatus can be widely applied to industrial equipment and household equipment using an AC motor.

  FIG. 1 is an overall schematic configuration diagram of a motor drive control device 10 including an AC motor inverter control device (hereinafter simply referred to as “control device”) 20 according to an embodiment of the present invention. FIG. 3 is a diagram showing functional blocks of the control device 20.

  As shown in FIG. 1, the motor drive control device 10 includes a battery B as a DC power source, system relays SR1 and SR2, and a converter 12 capable of boosting a DC voltage supplied from the battery B via a smoothing capacitor 33. , An inverter 16 that converts a DC voltage supplied from the converter 12 through the smoothing capacitor 14 into an AC voltage for driving the motor, and an AC motor driven by the AC voltage supplied from the inverter 16 (hereinafter simply referred to as “ M) and a control device 20 that outputs a control signal to the converter 12 and the inverter 16 based on a torque command τ * input from an external ECU (Electronic Control Unit).

  The motor drive control device 10 further includes a voltage sensor 22 and a temperature sensor 24 for detecting the output voltage VB and the temperature TB of the battery B, a current sensor 23 for detecting the battery current IB, and a voltage across the smoothing capacitor 14, that is, an inverter 16 A voltage sensor 26 for detecting the system voltage VH supplied to the motor, a current sensor 28 for detecting motor phase currents iu, iv and iw flowing from the inverter 16 to the U, V and W phase terminals of the motor M, and a motor. And a rotation angle sensor 30 made of, for example, a resolver for detecting the rotor rotation angle θ of M. Detection signals from the sensors 22 to 30 are output to the control device 20, respectively.

  The AC motor M is a three-phase synchronous type or three-phase induction type motor, and is a drive motor that generates torque for driving a drive wheel of a vehicle such as a hybrid vehicle or an electric vehicle. It can be configured to function as an output generator. Further, the motor M may be used as one that can provide power for starting an engine in a hybrid vehicle.

  The battery B can be composed of a secondary battery such as a nickel metal hydride battery or a lithium ion battery. System relay SR 1 is connected between the positive terminal of battery B and power line 32, and system relay SR 2 is connected between the negative terminal of battery B and ground line 34. The system relays SR1 and SR2 are turned on / off in response to a signal from the control device 20, and a DC voltage is supplied from the battery B to the smoothing capacitor 33 by turning on the system relays SR1 and SR2. Smoothing capacitor 33 is connected between power line 32 and ground line 34, smoothes the DC voltage supplied from battery B, and supplies the smoothed voltage to inverter 16.

  Converter 12 includes a reactor 37, power switching elements E1, E2, and diodes D1, D2. Switching elements E 1 and E 2 are connected in series between power line 32 and ground line 34. As the switching elements E1 and E2, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, a power bipolar transistor, or the like can be used. The diodes D1 and D2 are connected in antiparallel to the switching elements E1 and E2 so that current flows from the emitter side to the collector side.

  The reactor 37 is connected between the connection line 35 between the switching elements E1 and E2 and the power line 32. The smoothing capacitor 14 is connected between a power line 36 and a ground line 34 that connect the converter 12 and the inverter 16. Smoothing capacitor 14 supplies inverter 16 with system voltage VH obtained by smoothing the DC voltage supplied from converter 12.

  Inverter 16 includes a U-phase arm 38, a V-phase arm 40, and a W-phase arm 42 provided in parallel with each other between power line 36 and ground line 34. Each of the phase arms 38 to 42 includes two switching elements connected in series between the power line 36 and the ground line 34 and two diodes connected in antiparallel to the respective switching elements. More specifically, U-phase arm 38 includes switching elements E3 and E4 and diodes D3 and D4, V-phase arm 40 includes switching elements E5 and E6 and diodes D5 and D6, and W-phase arm 42 includes switching elements E7 and E8. And diodes D7 and D8. For example, an IGBT or the like can be used for each of the switching elements E3 to E8. The on / off states of the switching elements E3 to E8 are controlled by switching signals S3 to S8 from the control device 20.

  An intermediate point of each phase arm 38, 40, 42 is connected to one end of each phase coil (refer to FIG. 2) of U phase, V phase and W phase (hereinafter simply referred to as “three phases”) of motor M. ing. Each other end of each phase coil is commonly connected to a neutral point N (see FIG. 2) in the motor M.

  Converter 12 boosts DC voltage VB supplied from battery B during the boosting operation. The boosted DC voltage is supplied to the inverter 16 as the system voltage VH. More specifically, the ON period of switching element E1 and the ON period of switching element E2 are alternately provided according to switching signals S1 and S2 from control device 20, and the step-up ratio corresponds to the ratio of these ON periods. .

  On the other hand, converter 12 steps down the DC voltage supplied from inverter 16 via smoothing capacitor 14 and charges battery B during the step-down operation. More specifically, a period in which only switching element E1 is turned on and a period in which both switching elements E1, E2 are turned off are alternately provided in accordance with switching signals S1, S2 from control device 20, and the step-down ratio is This corresponds to the duty ratio of the ON period.

  When the DC voltage VH is supplied from the smoothing capacitor 14 when the torque command τ * of the motor M is positive (τ *> 0), the inverter 16 responds to the switching signals S3 to S8 from the control device 20. The motor M is driven so that the DC voltage VH is converted into an AC voltage by the on / off operation of the switching elements E3 to E8 and a positive torque is output. Further, when the torque command τ * of the motor M is zero (τ * = 0), the inverter 16 is turned on / off by the switching elements E3 to E8 according to the switching signals S3 to S8 from the control device 20. The motor M is driven so that the DC voltage VH is converted into an AC voltage and the torque becomes zero.

  Further, during regeneration of the vehicle on which the motor drive control device 10 is mounted, the torque command τ * of the AC motor M is set negative (τ * <0). In this case, the inverter 16 converts the AC voltage generated by the motor M into a DC voltage by ON / OFF operation of the switching elements E3 to E8 according to the switching signals S3 to S8, and the converted DC voltage is converted into the smoothing capacitor 14. To the converter 12 via Here, “regeneration” includes not only the case where the brake operation is performed by the driver of the vehicle but also the case where the acceleration of the vehicle is stopped or the vehicle is decelerated due to the release of the accelerator operation.

  The current sensor 28 detects the three-phase currents iu, iv, iw flowing through the motor M and outputs them to the control device 20. The rotation angle sensor 30 detects the rotor rotation angle θ of the motor M and outputs it to the control device 20. Instead of detecting the phase currents iu, iv, and iw with the three current sensors 28, there is a relationship of iu + iv + iw = 0, so that the two phase currents are detected and one remaining phase current is calculated. You may ask for it.

  Subsequently, a control method of the motor M by the control device 20 will be described. In general, three control methods are known as a control method for an AC motor: sine wave PWM control, overmodulation PWM control, and rectangular wave control.

  The sine wave PWM control is used as a general PWM control, and controls on / off of the switching element in each phase arm according to a voltage comparison between a sine wave voltage command value and a carrier wave (generally a triangular wave). . As a result, the motor input voltage becomes a sine wave within one control cycle of the inverter 16 for a set of a high level period corresponding to the on period of the upper arm element and a low level period corresponding to the on period of the lower arm element. Thus, the duty ratio is controlled. In the sine wave PWM control method, smooth rotation can be obtained even in a relatively low rotation range, but the modulation rate (or voltage utilization rate) that is the ratio of the effective value of the motor input voltage to the system voltage VH that is the inverter input voltage. ) Can only be increased to a maximum of 0.61.

  On the other hand, in the rectangular wave control method, one pulse of a rectangular wave whose ratio between the high level period and the low level period is 1: 1 is applied to the AC motor within the one control cycle. Thereby, the modulation factor can be increased to 0.78, and the output in a relatively high rotation range can be improved. Moreover, since the field weakening current can be reduced, the occurrence of copper loss in the AC motor M can be suppressed and the energy efficiency can be improved. Furthermore, since the number of times of switching in the inverter 16 can be reduced, there is also an advantage that switching loss can be suppressed.

  The overmodulation PWM control method is an intermediate PWM control method between sine wave PWM control and rectangular wave control, and is distorted so as to reduce the amplitude of the carrier wave and is similar to the above sine wave PWM control method. By performing this PWM control, it is possible to generate a motor input voltage distorted in a substantially sinusoidal shape shifted in the direction of increasing voltage, thereby increasing the modulation factor in the range of 0.61 to 0.78. .

  In the AC motor M, as the rotational speed and output torque increase, the induced voltage increases, and the required voltage increases accordingly. The boosted voltage by the converter, that is, the system voltage VH needs to be set higher than the required motor voltage. On the other hand, the voltage value that can be boosted by the converter 12 has an upper limit (system voltage maximum value).

  Therefore, in a region where the required motor voltage is lower than the maximum value of the system voltage VH, for example, 650 V, the maximum torque control by the sine wave PWM control method or the overmodulation PWM control method is applied and output by the motor current control according to the vector control. The torque is controlled to match the torque command τ *.

  On the other hand, when the necessary motor voltage exceeds the system voltage maximum value, the rectangular wave control method is applied according to the weakening magnetic field control while maintaining the system voltage VH at the maximum value. In this case, since the amplitude of the motor input voltage is fixed, torque control is performed by voltage phase control of a rectangular wave pulse based on the deviation between the estimated torque value and the torque command value.

  FIG. 2 shows an example of a functional block for executing the sine wave PWM control. The control device 20 includes a current command generator 52, a voltage command generator 54, a two-phase three-phase converter 56, a switching signal generator 58, a three-phase two-phase converter 60, a rotation speed calculator 62, and a carrier frequency controller. 64.

  The current command generator 52 receives a torque command τ * input from the external ECU to the control device 20, and receives a torque command τ * and a d-axis current command corresponding to the motor rotational speed Nm from a preset map or table. Id * and q-axis current command Iq * are calculated and output to voltage command generation unit 54. Here, the motor rotation speed Nm is calculated by the rotation speed calculator 62 based on the detection value θ detected by the rotation angle sensor 30.

  The voltage command generator 54 is configured to match the d-axis actual current id and the q-axis actual current iq with the d-axis current command Id * and the q-axis current command Iq *, respectively. Is calculated by the PI calculation of the following equation 1 and output to the two-phase / three-phase converter 56. The d-axis actual current id and the q-axis actual current iq here are based on the three-phase currents iu, iv, iw detected by the current sensor 28 in the three-phase / two-phase converter 60 based on the motor rotation angle θ. Use the converted version.

(Equation 1)
Vd * = Gpd (Id * -id) + Gid (Id * -id) dt
Vq * = Gpq (Iq * -iq) + Giq (Iq * -iq) dt
Here, Gpd and Gpq are proportional gains for d-axis and q-axis current control, and Gid and Giq are integral gains for d-axis and q-axis current control.

  The two-phase / three-phase converter 56 converts the d-axis voltage command Vd * and the q-axis voltage command Vq * into the three-phase voltage commands Vu, Vv, Vw based on the rotation angle θ of the AC motor M and performs switching. The signal is output to the signal generator 58. The system voltage VH is also reflected in the conversion from the d-axis voltage command Vd * and the q-axis voltage command Vq * to the three-phase voltage commands Vu, Vv, and Vw.

  The switching signal generator 58 generates switching signals S3 to S8 composed of PWM pulses and outputs them to the inverter 16 based on the comparison between the three-phase voltage command values Vu, Vv, Vw and the triangular wave carriers. . As will be described later, the carrier frequency used at this time is controlled to be switched between at least two types, for example, a low frequency f1 and a high frequency f2 in response to a signal from the carrier frequency control unit 64. Yes. Here, the high frequency f2 is preferably n times the low frequency f1 (n is a natural number of 2 or more).

  In response to the switching signals S3 to S8 from the switching signal generator 58, the switching elements E3 to E8 of the inverter 16 are subjected to switching control, so that torque corresponding to the torque command τ * is output to the motor M. An alternating voltage is applied. As described above, at the time of overmodulation PWM control, the carrier wave used in the switching signal generator 58 is switched from a general carrier wave at the time of sine wave PWM control to one distorted so as to reduce the amplitude.

  FIG. 3 shows a functional block example of the carrier frequency control unit 64. Carrier frequency control unit 64 includes a duty ratio calculation unit 66, a duty ratio determination unit 68, and a carrier frequency switching unit 70.

  The duty ratio calculation unit 66 receives switching signals S3, S5, and S7 corresponding to the switching elements E3, E5, and E7 arranged on the upper arms of the phase arms 38, 40, and 42 from the switching signal generation unit 58. In response, the duty ratios Du, Dv, Dw for the switching signals S3, S5, S7 are calculated and output to the duty ratio determination unit 68. The duty ratio here is the ratio of the ON time of the switching signal, which is a PWM pulse, to one carrier cycle, and the magnitude of the duty ratio corresponds to the magnitude of the voltage command value. On the other hand, when the voltage command value is large, the duty ratio is also large. Further, the carrier period here is that for the carrier of the low frequency f1.

  In response to the input from the duty ratio calculation unit 66, the duty ratio determination unit 68 determines whether or not each duty ratio Du, Dv, Dw is equal to or greater than a predetermined threshold value. As a threshold here, 0.6 is used suitably and 0.7 is used more suitably. Then, the determination result as to whether or not it is equal to or greater than the threshold is output to carrier frequency switching unit 70 as YES or NO.

  When the determination result in the duty ratio determination unit 68 is YES, the carrier frequency switching unit 70 switches the switching signal Sfu, Sfv, Sfw to the switching signal generation unit 58 so as to switch the carrier frequency from the low frequency f1 to the high frequency f2. Is output. On the other hand, since the processing in the carrier frequency control unit 64 is executed in a routine, the switching signal is switched so that the carrier frequency is switched from the high frequency f2 to the low frequency f1 when the determination result in the duty ratio determination unit 68 is NO thereafter. The switching signals Sfu, Sfv, and Sfw are output to the generation unit 58.

  FIG. 4 shows a current ripple occurrence state of the motor current when the carrier frequency is controlled to be switched by the carrier frequency control unit 64 as described above. This figure is the same as FIGS. 10A and 10B shown in the background art. The voltage command value (same for each phase) 1 in one control cycle (here, electrical angle 360 degrees) of the inverter 16, carrier 2 , Shows a motor current (actual current waveform) 4 including a PWM pulse 3 and a current ripple 5. As shown in FIG. 4, in the control device 20 of this embodiment, the absolute value of the voltage command value 1 is a relatively small region (electrical angle: 0 to about 45 degrees, about 135 to about 225 degrees, about 315 to 360 degrees. In the region where the absolute value of the voltage command value 1 is relatively large (electrical angle: about 45 to about 135 degrees, about 225 to about 315 degrees), the frequency is doubled. It is set to f2. Thus, by increasing the carrier frequency in the region where the voltage command value 1 is large, the current ripple 5 can be reduced as is clear from the comparison with FIG. 10A, and in the region where the voltage command value 1 is small, the carrier is The switching loss in the inverter 16 is not unnecessarily increased by lowering the frequency.

  In the above, control is performed so that the carrier frequency is switched between two types of frequencies f1 and f2 within one control cycle of the inverter 16, but the present invention is not limited to this. Switching control may be performed. For example, the carrier frequency f1 is set in the region where the duty ratio is up to the first threshold 0.6, the carrier frequency f2 is set in the region where the duty ratio is greater than or equal to the first threshold 0.6 and less than the second threshold 0.7, In a region where the duty ratio is equal to or higher than the second threshold value 0.7, control may be performed such that the carrier frequency f3 higher than f2 (for example, a frequency three times f1) is set.

  Next, with reference to FIG. 5 to FIG. 8B in which data actually measured using the control device 20 of the present embodiment are graphed, the considerations on which the present invention is derived will be described. FIG. 5 shows terminal voltages Eu-v, U-phase current Iu, and V-phase current Iv of the U-phase coil and V-phase coil of the motor M. Each of these U-phase current Iu and V-phase current Iv is shown in FIG. FIG. 6 shows current ripples ΔIu and ΔIv within one control cycle obtained from the waveform. As is apparent from FIG. 6, the magnitude of the current ripple fluctuates even within one control cycle.

  7A and 7B show the U-phase voltage command value Eu, the V-phase voltage command value Ev obtained from the current ripple execution value (left vertical axis) shown in FIG. 6 and the PWM waveform Eu-v in FIG. The UV phase voltage command value Eu-v and the current ripple execution values ΔIurms and ΔIvrms (unit: A) are shown. Here, the voltage command value is indicated by a duty ratio (right vertical axis) of maximum value = 1, and “rms” is an abbreviation for root mean square. 8A and 8B show the correlation between the voltage command value shown in FIGS. 7A and 7B and the magnitude of the current ripple.

  As shown in FIGS. 8A and 8B, it can be seen that there is a correlation that the larger the voltage command value, that is, the duty ratio, the larger the current ripple. In particular, it is shown that the current ripple tends to increase from the point indicated by reference numeral 90, that is, the point corresponding to the duty ratio of 0.6, and that the tendency becomes more prominent from the point indicated by reference numeral 92. It has been broken. Therefore, it can be said that the threshold value of the duty ratio used in the carrier frequency switching determination is preferably 0.6, and more preferably 0.7.

  8A and 8B, as schematically shown in FIG. 9, the increase rate of the current ripple is dI / dt = V / L (where I: current, V: voltage, L: inductance). For this reason, it is in good agreement with the theory that the current ripple increases as the applied voltage time to the motor increases, that is, as the duty ratio of the PWM pulse increases.

  From such considerations, increasing the carrier frequency only in the region where the current ripple is large, that is, the region where the voltage command value is large within one control cycle is effective in reducing the current ripple while suppressing the increase in inverter loss. Is derived.

  In the above embodiment, the magnitude of the voltage command value is determined using the duty ratio of the PWM pulse. However, the present invention is not limited to this, and the voltage command value is equivalent to the actual motor input voltage. Alternatively, since the motor input voltage is detected by providing a voltage sensor or the motor input voltage is calculated from the motor current detected by the current sensor 28, the duty ratio is 0.6 (or A case where the magnitude of the motor input voltage is determined using the voltage value corresponding to 0.7) as a threshold should be considered to be included in the technical scope of the present invention.

  In addition, the AC voltage output from the motor via the neutral point of the AC motor is converted to DC voltage by the inverter by inserting the plug at the tip of the electric cord extending from the AC power supply outside the vehicle into the inlet provided on the vehicle body. Thus, the present invention can also be applied to a so-called plug-in type electric vehicle that charges a power source.

It is a schematic block diagram of the motor drive control apparatus containing the control apparatus of the inverter for alternating current motors which is one Embodiment of this invention. It is a figure which shows the functional block of an inverter control apparatus. It is a figure which shows the functional block of a carrier frequency control part. It is a figure which shows the voltage command value in this embodiment, a carrier, a PWM pulse, and an actual current waveform. It is a graph which shows the waveform of an actual motor voltage and a motor current. It is a graph which shows the magnitude | size of an actual current ripple. It is a graph which shows the waveform of a U-phase voltage command value and a current ripple. It is a graph which shows the waveform of V phase voltage command value and a current ripple. It is a graph which shows the correlation with a U-phase voltage command value and a current ripple. It is a graph which shows the correlation with V-phase voltage command value and a current ripple. It is a figure which shows typically the relationship between the duty ratio of a PWM pulse, and a current ripple. It is a figure which shows the voltage command value in a prior art example, a carrier, a PWM pulse, and an actual current waveform. FIG. 11 is a diagram similar to FIG. 10 but showing a state where the carrier frequency is doubled.

Explanation of symbols

  1 voltage command value, 2 carrier, 3 PWM pulse, 4 motor current, 5 current ripple, 10 motor drive control device, 12 converter, 14 smoothing capacitor, 16 inverter, 20 control device, 22, 26 voltage sensor, 23, 28 current Sensor, 24 Temperature sensor, 30 Rotation angle sensor, 32, 36 Power line, 33 Smoothing capacitor, 34 Ground line, 35 Connection line, 37 Reactor, 38 U phase arm, 40 V phase arm, 42 W phase arm, 52 Current command Generator, 54 voltage command generator, 56 two-phase three-phase converter, 58 switching signal generator, 60 three-phase two-phase converter, 62 rotation speed calculator, 64 carrier frequency controller, 66 duty ratio calculator, 68 Duty ratio determination unit, 70 carrier frequency switching unit, B battery, D1, 2, D3, D4, D5, D6, D7, D8 diode, Du, Dv, Dw duty ratio, E1, E2, E3, E4, E5, E6, E7, E8 switching element, M AC motor, N neutral point, SR1, SR2 System relay.

Claims (3)

A control device for an inverter for an AC motor capable of converting a DC voltage supplied from a power source for application to an AC motor into an AC voltage by switching control of a plurality of switching elements,
A switching signal generation unit that generates a switching signal of the switching element based on a comparison between a carrier having a predetermined waveform and a voltage command value input from the outside, and outputs the switching signal to the inverter; in the inverter according to the voltage command value A carrier frequency control unit for controlling the frequency of the carrier,
The carrier frequency control unit includes a carrier frequency switching unit that sets a carrier frequency high in a region where the voltage command value is large and sets a carrier frequency low in a region where the voltage command value is small within one control cycle of the inverter. A control apparatus for an inverter for an AC motor, comprising: In the control apparatus of the inverter for alternating current motors of Claim 1,
The carrier frequency control unit compares the duty ratio input from the duty ratio calculation unit with a predetermined threshold and a duty ratio calculation unit that calculates a duty ratio for each switching signal generated by the switching signal generation unit. A duty ratio determination unit that determines whether to perform frequency switching,
The carrier frequency switching unit receives a determination result by the duty ratio determination unit and outputs a carrier frequency switching signal to the switching signal generation unit.   An inverter controlled by the AC motor inverter control device according to claim 1 or 2, and a DC power supplied from a power source is converted into an AC voltage by the inverter and applied for traveling power. An electric vehicle comprising an AC motor.

Images