JP2009278691A - Controller for permanent magnet type synchronous motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a controller for reducing a measurement time by improving the accuracy of measurement of inductance even if the electric constant of a motor is not recognized. <P>SOLUTION: A voltage compensation value calculator 33 calculates a voltage drop from a high-frequency current command value, a reactance estimation value and an armature resistance estimation value, and the voltage drop is used to feedforward-compensate the voltage command value. A parameter estimating means 36 estimates conductance and reactance so that an effective current/ineffective current detection value calculated from a high-frequency current/high-frequency voltage detection value may coincide with an effective current/ineffective current estimation value, and an impedance calculator 37 calculates a reactance estimation value and an armature resistance estimation value from the above estimation values. Also, an inductance calculator 38 calculates inductance of a motor 80 using the reactance estimation value. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a control device for a permanent magnet type synchronous motor, and more particularly to a technique for measuring the inductance of the motor with high accuracy when calculating the magnetic pole position of the permanent magnet type synchronous motor.

  In order to reduce the cost of a control device for a permanent magnet type synchronous motor (hereinafter also referred to as PMSM), a so-called sensorless control technique that operates without using a magnetic pole position detector has been put into practical use. Various methods have been proposed as this type of sensorless control technology, but many methods are used to calculate the magnetic pole position using the induced voltage induced between the motor terminals by the permanent magnet of the rotor. .

Here, in order to calculate the magnetic pole position using the induced voltage, an accurate inductance value of the electric motor is required.
The inductance of the electric motor can be measured from the relationship between the current when the alternating alternating current is passed through the electric motor and the terminal voltage. For example, there is a technique described in Patent Document 1.

In Patent Document 1, as Embodiment 5, a sine wave alternating alternating current command value is given, and a sine wave component and a cosine wave component are detected from the current detection value at that time. Are calculated, and these deviations are amplified by an integral regulator (integral compensator) to control the sine wave component and the cosine wave component of the voltage command value.
Thereby, the current of the motor can be controlled to the command value, and the inductance of the motor can be calculated from the detected current value, the terminal voltage, and the angular frequency of the alternating current.

In Patent Document 1, as Embodiment 6, a sinusoidal alternating AC voltage is applied to the motor, and the inductance is calculated from the cosine wave component of the current detection value, the amplitude of the AC voltage, and the angular frequency. The amplitude of the AC voltage is increased according to a predetermined rate of change, and the amplitude of the AC voltage is made constant when the magnitude of the AC current reaches a predetermined value, and the inductance is measured in this state.
Since this method does not perform the current feedback control as in the fifth embodiment of Patent Document 1 described above, there is a feature that the current control can be stably performed even when the electric constant of the motor is unknown.

Japanese Unexamined Patent Publication No. 2006-262643 (paragraphs [0077] to [0087], [0088] to [0097], FIG. 11, FIG. 12, etc.)

In order to measure the inductance of the electric motor with high accuracy, it is desirable to control with high accuracy so that the alternating current matches the command value. Also, in order to shorten the inductance measurement time, it is necessary to increase the current response.
When Patent Document 1 is studied from such a viewpoint, in the above-described fifth embodiment, in order to make the current response faster, the integral regulator may be optimally adjusted, but the electric constant of the motor is unknown. In this case, it is difficult to realize it. Further, since there is no linearity between the sine wave component and cosine wave component of the alternating current and the sine wave component and cosine wave component of the AC voltage, it is difficult to optimally adjust the integral controller in this respect.

  Furthermore, according to Embodiment 6 of Patent Document 1, current control can be stably realized even when the electric constant of the electric motor is unknown. However, in order to shorten the inductance measurement time, it is necessary to optimally design the rate of change in the amplitude of the AC voltage. If the electrical constant of the motor is unknown, it is considered difficult to realize this.

  Therefore, a problem to be solved by the present invention is to provide a control device for a permanent magnet synchronous motor that can improve the measurement accuracy of the inductance even when the electric constant of the motor is unknown, and further can shorten the measurement time of the inductance. It is in.

In order to solve the above problem, the invention according to claim 1 regards the terminal voltage and current of the permanent magnet type synchronous motor as vectors,
Current control means for controlling the terminal voltage to control a current of the motor to a current command value including a high-frequency current component alternating in a predetermined vector direction;
Inductance measuring means for measuring the inductance of the motor from the detected current value of the motor and the terminal voltage, and
The current control means includes
Means for amplifying a deviation between the current command value and the current detection value to calculate a first voltage command value;
Means for calculating a second voltage command value from the high frequency current component, the estimated reactance value of the electric motor and the estimated armature resistance value;
Means for calculating a third voltage command value from the sum of the first voltage command value and the second voltage command value, and setting the third voltage command value as a command value of the terminal voltage;
Have
The inductance measuring means includes
First detection means for detecting a high-frequency current having the same angular frequency as the high-frequency current component from the current detection value;
Second detection means for detecting a high-frequency voltage having the same angular frequency as the high-frequency current component from the third voltage command value;
Parameter estimation means for estimating conductance and susceptance from the high-frequency current detection value obtained by the first detection means and the high-frequency voltage detection value obtained by the second detection means;
Impedance calculating means for calculating the reactance estimated value and the armature resistance estimated value from the conductance estimated value and the susceptance estimated value obtained by the parameter estimating means;
Means for calculating the inductance of the motor from the reactance estimate and the angular frequency of the high-frequency current component;
It is what has.

The invention according to claim 2 is the control device according to claim 1,
The parameter estimation means includes
Means for detecting an effective current that is a current in phase with the high-frequency voltage detection value from the high-frequency current detection value and the high-frequency voltage detection value;
Effective current estimation means for estimating an effective current from the high-frequency voltage detection value and the conductance estimation value;
Means for amplifying a deviation between the effective current estimated value obtained by the effective current estimating means and the detected value of the active current to calculate the conductance estimated value;
Means for detecting a reactive current which is a current having a phase difference of 90 degrees from the high frequency voltage detection value from the high frequency current detection value and the high frequency voltage detection value;
Reactive current estimation means for estimating reactive current from the high-frequency voltage detection value and the susceptance estimation value;
Means for amplifying a deviation between the reactive current estimated value obtained by the reactive current estimating means and the detected value of the reactive current to calculate the susceptance estimated value;
It is equipped with.

In the present invention, the voltage drop due to the high-frequency current is calculated from the high-frequency current command value, the reactance estimated value, and the armature resistance estimated value, and this voltage drop (voltage compensation value) is used as the second voltage command value. Feed-forward compensation for voltage command value. Further, the conductance of the armature winding is set such that the effective current detection value and the reactive current detection value calculated from the high frequency current detection value and the high frequency voltage detection value based on the third voltage command value match the respective command values. And the susceptance are estimated, and the estimated reactance value and the estimated armature resistance value are calculated using these estimated values.
As a result, the estimated reactance value and the estimated armature resistance finally converge to the true value, so that the voltage drop due to the high-frequency current is accurately calculated, and this voltage drop is compensated according to the third voltage command value. By controlling the electric motor, even when the electric constant of the electric motor is unknown, the high-frequency current can be controlled with high accuracy, and the inductance measurement accuracy can be improved. In addition, since the current can be controlled with high response, the inductance measurement time can be shortened.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
First, PMSM can realize highly accurate torque control by performing current control according to the d-axis of the rotor (the magnetic pole direction of the rotor) and the q-axis advanced 90 degrees from the d-axis. However, if the magnetic pole position detector is not provided, the d and q axes cannot be directly detected. Therefore, γ and δ of the orthogonal rotation coordinate system that rotates at the angular velocity (= speed calculation value) ω ₁ corresponding to the d and q axes. The control calculation is performed by estimating the axis to the control device side.
The definition of the γ and δ axes is shown in FIG. In FIG. 3, ω _r is the rotational angular velocity of the d and q axes, and θ _err is the angular error (position calculation error) between the d and q axes and the γ and δ axes.

  Next, FIG. 1 is a control block diagram showing an embodiment of the present invention corresponding to claim 1. This block diagram is for measuring a d-axis inductance that is an inductance parallel to the magnetic pole of the PMSM.

  First, the configuration of the main circuit in FIG. 1 will be described. Reference numeral 50 denotes a three-phase AC power source, and the rectifier circuit 60 rectifies the three-phase AC voltage of the power source 50 and converts it into a DC voltage. This DC voltage is supplied to a power converter 70 composed of a PWM inverter, and a predetermined signal for driving the permanent magnet type synchronous motor 80 is controlled by controlling an internal semiconductor switching element by a gate signal from a PWM circuit 13 described later. Converted to three-phase AC voltage.

Next, the configuration and operation of the control device are as follows.
The current coordinate converter 14 converts the phase current detection values i _u and i _w detected by the u-phase current detector 11u and the w-phase current detector 11w, respectively, into the γ and δ axes based on the angle θ ₁ of the γ and δ axes. Coordinates are converted to current detection values i _γ and i _δ . The angle θ ₁ is the magnetic pole position θ ₁₀ calculated before starting the inductance measurement. Thereby, the γ and δ axes can be matched with the d and q axes.

Calculation method of the magnetic pole position theta ₁₀ is arbitrary, for example, by applying the technique described in JP-A-2001-190093, retract the rotor current vector to control the amplitude of the current constant The method of obtaining the magnetic pole position from the angle of the current vector at this time, or as described in Japanese Patent No. 3321472, the applied vector from the parallel component or the orthogonal component of the alternating voltage vector or alternating current vector applied to the motor A method of detecting the phase difference angle with the magnetic flux axis and detecting the magnetic pole position directly or indirectly from the phase difference angle may be used.

In FIG. 1 again, the integrator 30 integrates the angular frequency ω _h of the high frequency current command value to calculate the phase θ _h of the high frequency current command value. The high-frequency current command calculator 31 calculates the γ-axis high-frequency current command value i _γh ^* from the γ-axis high-frequency current amplitude command value I _γh ^* and the phase θ _h using Equation 1.

The adder 32 adds the direct current command value (γ-axis fundamental wave current amplitude command value) I _γf ^* and the γ-axis high-frequency current command value i _γh ^* to calculate the γ-axis current command value i _γ ^* . The direct current command value I _γf ^* is set to zero or a constant positive value. When the DC current command value I _γf ^* is set to a positive constant value, the rotor can be prevented from rotating due to an external force.

The subtractor 19 obtains a deviation between the γ-axis current command value i _γ ^* and the detected γ-axis current value i _γ, and amplifies the deviation by a γ-axis current regulator 20 including a proportional-plus-integral regulator. The γ-axis voltage command value v _γACR ^* is calculated.
Further, the voltage compensation value calculator 33 uses the γ-axis high-frequency current amplitude command value I _γh ^* , the phase θ _h , the armature resistance estimated value R _{γhest of} the motor 80 and the reactance estimated value X _γhest to cause a voltage drop ( Voltage compensation value) is obtained, and this is set as the second γ-axis voltage command value v _γhFF ^* . Specifically, the calculation of Formula 2 is performed.

In addition, the calculation method of the armature resistance estimated value R _γhest and the reactance estimated value X _γhest in Formula 2 will be described later.
The adder 21 adds the first γ-axis voltage command value v _γACR ^* and the second γ-axis voltage command value v _γhFF ^* to perform feedforward compensation, and the third γ-axis voltage command value v _γ ^*. Is calculated. On the other hand, the δ-axis voltage command value v _δ ^* is controlled to zero.
These γ and δ-axis voltage command values v _γ ^* and v _δ ^* are converted into phase voltage command values v _u ^* , v _v ^* and v _w ^* by the voltage coordinate converter 15 based on the angle θ ₁ (= θ ₁₀ ). Converted.
The PWM circuit 13 phase-converts the output voltage of the power converter 70 from the phase voltage command values v _u ^* , v _v ^* , v _w ^* and the DC input voltage E _dc of the power converter 70 detected by the voltage detector 12. A gate signal for controlling the voltage command values v _u ^* , v _v ^* , and v _w ^* is generated and supplied to the power converter 70.
The γ-axis current i _γ can be controlled to the γ-axis current command value i _γ ^* by the arithmetic processing of the control device described above.

Next, a calculation method of the armature resistance estimated value R _γhest and the reactance estimated value X _γhest will be described.
Fourier series calculator 34 as the first detection means, gamma-axis current i _gamma and high-frequency current command value of the phase θ from is _h, the angular frequency of the gamma-axis current i _gamma omega _h cosine wave component I _Ganmaha and sinusoidal components I _γhb is detected. On the other hand, Fourier series calculator 35 as the second detection means, gamma-axis voltage value v _gamma ^* and from the phase theta _h of the high-frequency current command value, gamma-axis voltage cosine of command value v _gamma ^* angular frequency omega _h A wave component V _γha and a sine wave component V _γhb are detected.

The parameter estimation means 36 calculates the _{conductance from} the cosine wave component I _γha and sine wave component I _γhb of the γ-axis current i _γ and the cosine wave component V _γha and sine wave component V _{γhb of} the γ-axis voltage command value v _γ ^*. An estimated value G _γhest and a susceptance estimated value B _γhest are calculated. Details of the parameter estimation means 36 will be described later.

The impedance calculator 37 calculates an armature resistance estimated value R _γhest and a reactance estimated value X _γhest from the conductance estimated value G _γhest and the susceptance estimated value B _{γhest according} to the equations 3 and 4, respectively.

The inductance calculator 38 calculates the measured value L _dest of the d-axis inductance using Equation 5 from the reactance estimated value X _γhest and the angular frequency ω _h .

Next, details of the parameter estimation means 36 will be described. FIG. 2 is a block diagram showing a configuration of the parameter estimation means 36 corresponding to claim 2. In FIG.
In the parameter estimation means shown in FIG. 2, the product of the high-frequency voltage amplitude and the conductance of the armature winding is equal to the magnitude of the current in phase with the high-frequency voltage (hereinafter referred to as an effective current). Utilizing the fact that the product of the susceptance of the child winding is equal to the magnitude of the current that is 90 degrees out of phase with the high-frequency voltage (hereinafter referred to as reactive current), the conductance and susceptance are converged to true values. It is.

2, the amplitude calculator 101, the amplitude V _Ganmahc of the high-frequency voltage included in gamma-axis voltage value v _gamma ^*, and gamma-axis voltage value v _gamma ^* of the cosine wave component V _Ganmaha the sine wave component V _Ganmahb From the above, the calculation is performed using Equation 6.

The effective current / reactive current detector 102 calculates the effective current detection value I _γpowdet and the reactive current detection value I _{γvardet according} to Equations 7 and 8, respectively, using the relationship between the high frequency voltage and the high frequency current.

On the other hand, as shown in Equations 9 and 10, the products of the high-frequency voltage amplitude V _γhc and the conductance estimated value G _γhest and the products of the high-frequency voltage amplitude V _γhc and the susceptance estimated value B _γhest are multiplied by the multipliers 103a and 103a, respectively. The effective current estimated value I _γpost and the reactive current estimated value I _γvarest are calculated by _103b .

The subtractor 104a calculates the deviation I _γpowerr between the effective current estimated value I _γpower and the effective current detected value I _γpower, and the parameter estimator 105a amplifies it to calculate the conductance estimated value G _γhest . Specifically, Formula 11 is used.

In Equation 11 and Equation 12 below, Γ is a parameter estimation integral gain, and ρ is a normalization coefficient.
Similarly, the subtractor 104b calculates a deviation I _γbarerr between the reactive current estimated value I _γvarest and the reactive current detected value I _γvardet, and a parameter estimator 105b amplifies it to calculate a susceptance estimated value B _γhest . Specifically, Formula 12 is used.

Through the above arithmetic processing, the conductance estimated value G _γhest and the susceptance estimated value B _γhest can be converged to true values. Thereby, the armature resistance estimated value R _γhest and reactance estimated value X _γhest calculated by the impedance calculator 37 in FIG. 1 quickly converge to true values.
Therefore, the voltage compensation value calculator 33 accurately calculates the voltage drop due to the high-frequency current based on Equation 2 described above, and outputs the second γ-axis voltage command value v _γhFF ^* , and this γ-axis voltage command value v By controlling the motor 80 via the power converter 70 using the third γ-axis voltage command value v _γ ^* compensated by _γhFF ^{*, the} current can be made highly responsive even when the electric constant of the motor 80 is unknown. Can be controlled.

The embodiment shown in FIG. 1 is for measuring the d-axis inductance, but when measuring the q-axis inductance that is the inductance in the direction orthogonal to the magnetic pole of the rotor, the δ-axis is similarly used. An alternating current may be passed in the direction, and the q-axis inductance may be measured from the δ-axis current i _δ and the δ-axis voltage command value v _δ ^{* at} this time. Detailed description is omitted.

It is a control block diagram which shows embodiment of this invention. It is a block diagram which shows the structure of the parameter estimation means in FIG. It is a figure which shows the definition of (gamma) and (delta) axis | shaft.

Explanation of symbols

50 Three-phase AC power supply 60 Rectifier circuit 70 Power converter 80 Permanent magnet synchronous motor (PMSM)
11u u-phase current detector 11w w-phase current detector 12 voltage detector 13 PWM circuit 14 current coordinate converter 15 voltage coordinate converter 19 subtractor 20 γ-axis current regulator 21 adder 30 integrator 31 high-frequency current command calculator 32 Adder 33 Voltage Compensation Value Calculator 34 Fourier Series Calculator 35 Fourier Series Calculator 36 Parameter Estimator 37, 38 Impedance Calculator 101 Amplitude Calculator 102 Active Current / Reactive Current Detector 103a, 103b Multiplier 104a, 104b Subtraction 105a, 105b Parameter estimator

Claims (2)

Taking the terminal voltage and current of a permanent magnet synchronous motor as vectors,
Current control means for controlling the terminal voltage to control a current of the motor to a current command value including a high-frequency current component alternating in a predetermined vector direction;
Inductance measuring means for measuring the inductance of the motor from the detected current value of the motor and the terminal voltage;
With
The current control means includes
Means for amplifying a deviation between the current command value and the current detection value to calculate a first voltage command value;
Means for calculating a second voltage command value from the high frequency current component, the estimated reactance value of the electric motor and the estimated armature resistance value;
Means for calculating a third voltage command value from the sum of the first voltage command value and the second voltage command value, and setting the third voltage command value as a command value of the terminal voltage;
Have
The inductance measuring means includes
First detection means for detecting a high-frequency current having the same angular frequency as the high-frequency current component from the current detection value;
Second detection means for detecting a high-frequency voltage having the same angular frequency as the high-frequency current component from the third voltage command value;
Parameter estimation means for estimating conductance and susceptance from the high-frequency current detection value obtained by the first detection means and the high-frequency voltage detection value obtained by the second detection means;
Impedance calculating means for calculating the reactance estimated value and the armature resistance estimated value from the conductance estimated value and the susceptance estimated value obtained by the parameter estimating means;
Means for calculating the inductance of the motor from the reactance estimate and the angular frequency of the high-frequency current component;
A control device for a permanent magnet type synchronous motor. The control device according to claim 1,
The parameter estimation means includes
Means for detecting an effective current that is a current in phase with the high-frequency voltage detection value from the high-frequency current detection value and the high-frequency voltage detection value;
Effective current estimation means for estimating an effective current from the high-frequency voltage detection value and the conductance estimation value;
Means for amplifying a deviation between the effective current estimated value obtained by the effective current estimating means and the detected value of the active current to calculate the conductance estimated value;
Means for detecting a reactive current which is a current having a phase difference of 90 degrees from the high frequency voltage detection value from the high frequency current detection value and the high frequency voltage detection value;
Reactive current estimation means for estimating reactive current from the high-frequency voltage detection value and the susceptance estimation value;
Means for amplifying a deviation between the reactive current estimated value obtained by the reactive current estimating means and the detected value of the reactive current to calculate the susceptance estimated value;
A control device for a permanent magnet type synchronous motor.

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