JP2008109710A - Communication system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a communication system which minimizes the effect of noise on a transmission line, and a sender and a receiver used in the communication system. <P>SOLUTION: In the communication system connecting the sender and the receiver via the transmission line, an input signal is transmitted by (orthogonal) carriers which are so arranged as not to interfere with each other on the axis of a frequency. In addition, it is further effective to transmit an input signal by carrier signals which are so arranged as not interfere with each other also on the axis of a time. When a carrier signal influenced by noise during transmission is removed at a receiving side or the percentage in synthesis or distribution is reduced, noise on the transmission line can be eliminated. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a communication system, and more particularly to a communication system via a high-noise line such as a lamp line, and a transmitter and receiver thereof.

  Conventionally, in a communication system in which the influence of narrow band noise such as distortion and impulse noise in a transmission line cannot be ignored, data is spread on the transmission side and then transmitted through the transmission line, and the reception side receives the spread data received. A spread data transmission method of despreading is used as a useful means.

  The data transmission system using this conventional spread data transmission system will be described below by taking as an example a data transmission system using a power line carrier that uses an existing power line (AC100V, 50/60 Hz lamp line) as a transmission line. To do.

  FIG. 15 is a block diagram showing an example of a data transmission system using a power line carrier using a conventional direct spreading method. In FIG. 15, a transmitter 100 and a receiver 200 are connected via a transmission line 300.

  The transmitter 100 includes a mixer 110, a pseudo noise generator 111, a carrier wave oscillator 112, and a balanced modulator 113. The receiver 200 includes a mixer 210, a pseudo noise generator 211, a carrier wave oscillator 212, a balanced modulator 213, and an IF-BPF 214.

  While the input data is input to the mixer 110 on the transmitter side, a spread code (described later) from the pseudo noise generator 111 is also input to the mixer 110. Here, both signals are multiplied and balanced modulation of the next stage is performed. Is input to the device 113. The carrier wave oscillator 112 generates a carrier wave and inputs the carrier wave to the balanced modulator 113, whereby the balanced modulator 113 modulates the carrier wave by a signal from the mixer 110 (a signal obtained by spreading the input signal) and modulates the carrier wave. The carrier wave is transmitted to the receiver via the transmission line 300 except for the carrier wave.

  In the receiver 200, the carrier wave oscillator 212 generates a carrier wave having the same frequency as the carrier wave generated by the carrier wave oscillator 112 on the transmitter 100 side and inputs the carrier wave to the balanced modulator 213. On the other hand, the pseudo noise generator 211 generates a despread code having a phase opposite to that of the spread code generated by the transmitter 100 and inputs the despread code to the balanced modulator 213, whereby the balanced modulator 213 is a carrier wave oscillator. The carrier wave output from 212 is modulated using the despread code output from the pseudo noise generator 211 and output to the mixer 210. The mixer 210 multiplies the modulation signal input via the transmission line 300 by the modulation signal input from the balanced modulator 213 and outputs the result to the IF-BPF 214. The IF-BPF 214 is a BPF (band pass filter) that passes a frequency band of IF (intermediate frequency).

  Now, in the transmitter 100, it is assumed that the data signal input to the mixer 110 is a data signal having a spectrum as shown in FIG. The mixer 110 performs spectrum spreading by multiplying the input data signal using the spreading code provided from the pseudo noise generator 111. The spectrum waveform of the input data signal after this spreading is shown in FIG. The spread data signal is modulated on the carrier wave output from the carrier wave oscillator 112 by the balanced modulator 113 and output onto the transmission line 300. Here, the spread code refers to, for example, “1111100011011101010000100101100” (for example, “0000011100100010101111011010011” for input “0”) which is multi-bit (for example, 31 bits) for input “1”.

  Here, impulse noise (portion indicated by hatching in FIG. 16C) occurs during transmission of the data signal through the transmission line 300, and the receiver 200 receives the signal shown in FIG. 16C. Let us consider the case of receiving a signal.

  As described above, the carrier wave output from the carrier wave generator 212 of the receiver 200 is modulated by the despreading code provided from the pseudo noise generator 211 in the balanced modulator 213. Further, the mixer 210 performs spectrum spreading by multiplying the modulated signal and the spread data signal obtained via the transmission path 300. Here, the despread code is an absolute OR with the above spread code, and all bits of the spread code are “1” with respect to the input “1” (in contrast, all bits of the spread code with respect to the input “0” are “ 0 "), that is, a code obtained by inverting the above spreading code.

  In the multiplication process performed by the mixer 210, the data signal spread by the transmitter 100 is despread, and normal spreading is performed for impulse noise. Accordingly, the spectrum waveform of the data signal after performing the multiplication process (that is, the despreading process) is restored to the state shown in FIG. As a result, the impulse signal generated in step (2) is diffused and the level of the data signal is remarkably lowered, and the influence of the impulse signal on the data signal can be reduced.

  However, in order for the despreading to be performed accurately, it is of course necessary to accurately synchronize the signal input from the transmission path to the mixer 210 and the modulation signal input from the balanced modulator 213. .

  As described above, in the conventional data transmission system based on the direct spreading method, the influence of narrow band noise such as impulse noise and the distortion of the transmission line due to the use state of the equipment connected to the transmission line (for example, the power line (that is, household use) The line noise at the time of starting the compressor of a refrigerator / freezer connected to a 100V electrical outlet) is reduced by performing spectrum spreading and despreading processing as shown in FIGS. 16 (c) and 16 (d). ing.

These spread spectrum processing techniques are described in detail in Non-Patent Document 1.
"Spread spectrum communication system" JATEC Publishing, P9-P28

  The data transmission method using the conventional direct spreading method having the above-described configuration can reduce the influence of narrow band noise and transmission path distortion to some extent as described above. However, conversely, there is a drawback that it is not possible to completely eliminate the influence of narrowband noise and transmission path distortion in the entire frequency band to be used. For this reason, when the noise and distortion generated during transmission is very large compared to the level of the input data signal, the effect of sufficiently reducing the noise and distortion can be achieved even if the diffusion process using the conventional method is performed. It may not be obtained.

  In addition, since the above spread code has a wide frequency range, the dedicated frequency band of the modulation signal is widened, and many side lobes are generated over the wide band in addition to the main lobe. Therefore, there is a drawback that the energy loss at the side lobe increases and the transmission efficiency is lowered. Further, as described above, the despreading process needs to synchronize the signal obtained from the transmission line and the signal obtained from the balanced modulator, and this synchronization process is complicated when executed by a fairly complicated circuit or program. In addition to increasing the cost due to the procedure, there is a case where data detection becomes impossible because the synchronization accuracy is insufficient at present.

  The present invention has been proposed in view of the above-described conventional circumstances, and is a communication that performs high-speed and high-quality data transmission by actively using a frequency band that is not affected by narrow-band noise or transmission path distortion. The purpose is to provide a system. It is another object of the present invention to provide a communication system capable of improving transmission efficiency and improving transmission accuracy by using a plurality of carriers whose frequencies take values at predetermined intervals.

  The present invention employs the following means in order to achieve the above object. First, it is assumed that the present invention is applied to a communication system in which a transmitter 1 and a receiver 2 are connected via a transmission path.

  The transmission signal generating means 10 is provided that converts the input signal into a plurality of carrier signals that take values at predetermined intervals that do not interfere with each other on the frequency axis and outputs the carrier signals. As a result, even if noise that interferes with any of the plurality of frequencies occurs in the transmission path, a good communication result can be obtained by removing only the carrier of the frequency.

  The transmission signal generation means 10 includes a carrier signal generation means 12 that generates a plurality of carrier signals whose frequencies take values at predetermined intervals, and a multiplication means 11 that multiplies each of the carrier signals by an input signal and sends it to the transmission line. With a configuration.

  Whether or not noise is generated in the transmission path is determined by providing the receiver 2 with the transmission path characteristic measuring means 20, detecting the characteristics of the transmission path, receiving the measurement result by the transmission path characteristic measuring means 20, and transmitting the transmitter 1 Alternatively, the determination is made by the selection control means 40 provided in the receiver 2.

  The selection control means 40 reflects the result on the transmission signal from the transmitter 1 or reflects it on the reception signal input to the receiver 2 via the transmission path. That is, the selection control unit 40 controls the generation of the carrier signal in the carrier signal generation unit 12 of the transmitter 1 and does not transmit the carrier signal to the transmission line with poor characteristics or to the transmission line with poor characteristics. To reduce the carrier signal strength ratio. Alternatively, the selection control means 40 does not set a carrier signal from a transmission path with poor characteristics to a signal to be received by the receiver 2, or a ratio of the combined strength of carrier signals from a transmission path with poor characteristics. Is to lower.

  The transmission line characteristic measuring means 20 measures the characteristic of the transmission line based on the absolute value of the intensity of the received signal or the phase difference from the reference phase, and inputs the result to the selection control means 40.

  The present invention is more effective when a carrier signal that does not interfere with other signals on the frequency axis (called orthogonal) on the frequency axis as described above and that forms a carrier signal that does not interfere with other signals on the time axis is formed. be able to.

  That is, the transmitter 1 generates a carrier signal through the plurality of filters 52 that satisfy the orthogonal condition on the frequency axis and the time axis (overlapping orthogonality) in the transmitter 1. On the other hand, the receiver 2 uses a plurality of filters 62 that form a signal that is the same as the transmission signal but only delayed in time. As a result, a transmission signal having a narrow band not only on the frequency axis but also on the time axis can be formed, and a transmission signal that is not easily affected by noise generated on the line can be formed.

  Also in this case, it is preferable to remove carriers flowing on a line with poor transmission characteristics or to lower the distribution rate. When there are a plurality of types of input signals, basically a plurality of sets of transmission signal forming means 10 are required. However, since the encoder is used when the input signal is divided into a number corresponding to a plurality of filters when the above overlap orthogonality is used, the input A is connected to the filters a to c and the input B It is preferable to provide an assignment function such as filters d to f.

  By adopting the above configuration, the present invention improves the transmission efficiency and the transmission accuracy by using a plurality of carriers orthogonal to each other on the frequency axis, and has the effect of narrowband noise and transmission path distortion. High-quality data transmission can be performed at high speed by actively using no frequency band.

  Hereinafter, embodiments of the present invention will be described with reference to FIGS.

(First embodiment)
FIG. 1 is a block diagram showing a communication system according to the first embodiment of the present invention. In FIG. 1, in the communication system according to the first embodiment of the present invention, a transmitter 1 and a receiver 2 are connected via a transmission line 3.

  The transmitter 1 includes transmission signal generation means 10 including a multiplication unit 11 and a carrier signal generation unit 12. The receiver 2 includes four DFT (discrete Fourier transform) processing units 21 to 24 and a relative phase detection circuit 25 constituting the transmission line characteristic measuring means 20, and the measurement result of the transmission line characteristic measuring means 20. And a selection control means 40 for determining the carrier signal synthesis distribution in the reception synthesis means 30 in the next stage, and the reception synthesis means 30.

  In the transmitter 1, the carrier signal generation unit 12 generates a plurality of carrier signals whose frequencies take values at predetermined intervals, and inputs the carrier signals to the multiplication unit 11. The multiplication unit 11 multiplies the modulation data (data after the input data is modulated by a modulator (not shown)) and the plurality of carrier signals supplied from the carrier signal generation unit 12 as described above, thereby transmitting a transmission signal. Output as. This transmission signal is transmitted to the receiver 2 through the transmission path 3.

In the receiver 2, the transmission signal transmitted through the transmission path 3 is input to the four DFT processing units 21 to 24, respectively. Each of the four DFT processing units 21 to 24 has a predetermined signal band for processing, and according to this, an absolute value signal a _{1 to} a ₄ and an angle signal b _{1 to} b ₄ to be described later are subjected to Fourier transform processing. To detect. The absolute value signals a _{1 to} a ₄ detected by the DFT processing units 21 to 24 are input to the selection control means 40, while the angle signals b _{1 to} b ₄ are input to the relative phase detection circuit 25. Is input.

The relative phase detection circuit 25 detects the relative phases of the input angle signals b _{1 to} b ₄ and the reference phase, and outputs them to the selection control means 40 as relative phase signals f _{1 to} f ₄ . Here, the reference phase may be determined in advance or may be given from the transmitter 1 by transmission.

The selection control means 40 is based on the absolute value signals a _{1 to} a ₄ input from the DFT processing units 21 to 24 and the relative phase signals f _{1 to} f ₄ input from the relative phase detection circuit 25. A selection signal Ss for selecting which of the carrier signals is input to the reception synthesis means 30 in the next stage. The reception synthesizing unit 30 uses the selection signal Ss to synthesize the selected carrier signal using one or both of the absolute value signals a _{1 to} a _{4 and the} angle signals b _{1 to} b _4. It has become. The received combined data is demodulated by a demodulator (not shown) to finally obtain output data. The type of signal required in the reception synthesis process is determined by the modulation method and other conditions.

  Hereinafter, in the present embodiment, a case where the modulation data (input signal) input to the multiplication unit 11 of the transmitter 1 is PSK modulation data subjected to PSK (phase shift keying) modulation will be described as an example. To do. It is assumed that the spectrum waveform of this modulation data is as shown in FIG. The carrier signal generator 12 outputs a carrier signal corresponding to the impulse response of the transfer function H (ω) given by the following equation (1). The impulse waveform of the transfer function H (ω) is shown in FIG. It shall be shown in 2 (b).

  From the above equation (1), the carrier signal output from the carrier signal generator 12 is composed of carrier signals (frequency: ω0, ω0 + ωc, ω0 + 2ωc, and ω0 + 3ωc) having four values with equal frequency (ωc). . When the modulation signal and the plurality of carrier signals are input to the multiplication unit 11 as described above, the multiplication unit 11 converts each carrier signal shown in FIG. 2B to the carrier as shown in FIG. A transmission signal having the spectrum waveform shown in FIG.

Although the number of carrier signals is four here, the number is not limited to this, and the number of carrier signals can be arbitrarily set as necessary. In the receiver 2, the DFT processing units 21 to 24 extract DFT processing for each carrier signal, that is, extract absolute value signals a _{1 to} a ₄ and angle signals b _{1 to} b ₄ for each carrier signal from the transmission signal. Here, the DFT processing unit 21 corresponds to a carrier signal of frequency ω0, the DFT processing unit 22 corresponds to a carrier signal of frequency ω0 + ωc, the DFT processing unit 23 corresponds to a carrier signal of frequency ω0 + 2ωc, and the DFT processing unit 24 DFT processing is performed corresponding to the carrier signal of frequency ω0 + 3ωc, and if the number of carrier signals is set to a number other than the above four, it is necessary to change the number of DFT processing units accordingly. Of course there is.

The angle signals b _{1 to} b ₄ detected by each of the DFT processing units 21 to 24 are input to the relative phase detection circuit 25, and the relative phase detection circuit 25 receives the input angle signals b _{1 to} b ₄ and The relative phase with respect to the reference phase in each carrier signal is detected. For example, when a PSK modulation signal as shown in the first embodiment is transmitted, the relative phase is detected by determining a reference phase in advance and calculating a phase difference between the angle of the carrier signal and the reference phase. In addition, if a modulated signal according to the DPSK (differential phase shift keying) modulation method is transmitted, the phase difference between the previous signal and the current signal can be obtained. .

The absolute value signals a _{1 to} a ₄ of the carrier signals detected by the DFT processing units 21 to 24 and the relative phase signals f _{1 to} f ₄ output from the relative phase detection circuit 25 are sent to the selection control means 40. Entered. The selection control means 40 estimates the characteristics of the transmission path as follows based on the intensity of the input absolute value signals a _{1 to} a ₄ and the values of the relative phase signals f _{1 to} f ₄ , and appropriately according to the estimation. The selection signal Ss is formed, and the selection signal Ss is input to the reception synthesis unit 30.

Hereinafter, a method for estimating transmission path characteristics performed by the selection control unit 40 will be described in order with reference to FIGS. The estimation methods include a method using absolute value signals a _{1 to} a ₄ as parameters (FIG. 3), a method using relative phase signals f _{1 to} f ₄ as parameters (FIG. 4), and absolute value signals a ₁ to a. and a method (FIG. 5) for both a ₄ and the relative phase signal f ₁ ~f ₄ as a parameter.

First, in the method using the absolute value signals a _{1 to} a ₄ as parameters, it is determined whether or not there is any distortion depending on the intensity levels of the absolute value signals a _{1 to} a ₄ . This determination is made based on whether or not the intensity level of the absolute value signals a _{1 to} a ₄ exceeds the threshold value by providing a threshold value of an arbitrary intensity level in advance.

For example, the intensity of the absolute value signals a _{1 to} a ₄ of the carrier signals detected by the DFT processing units 21 to 24 is shown in FIG. As shown, when individually attenuated, it can be determined that a carrier signal having an intensity level exceeding a predetermined threshold value α is not affected by transmission path distortion at all. Therefore, in the case of FIG. 3A, the carrier signal of the frequency ω0 + 2ωc that does not exceed the threshold value α is excluded in the selection control means 40, and the carrier signals of the frequencies ω0, ω0 + ωc, ω0 + 3ωc are selected. The three carrier signals are synthesized by the reception synthesis means 30 and output as detection data.

  In addition, as shown in FIG. 3B, in the case of transmission line characteristics in which the intensity level of the absolute value in some carrier signals is increased due to narrow band noise, the threshold value is reversed, as opposed to the above case. It can be determined that the carrier signal that outputs an intensity level equal to or lower than β is not affected by narrowband noise (that is, the signal level that the receiver 2 originally receives does not exceed the transmission level of the transmitter 1). Therefore, it can be determined that a signal having a level exceeding that level includes some noise or the like). Accordingly, in this case, the selection control means 40 excludes carrier signals having the frequency ω0 + ωc exceeding the threshold β, while selecting the carrier signals having the frequencies ω0, ω0 + 2ωc, ω0 + 3ωc, and the three carrier signals. The received signal is combined by the reception combining means 30 and output as detection data.

  Further, as shown in FIG. 3 (a), if the predetermined value is set as the lower limit threshold value α and another predetermined value is set as the upper limit threshold value β as shown in FIG. 3 (b). Any of the above cases can be dealt with.

In the description of the method using the absolute value signals a _{1 to} a ₄ as parameters, the distribution rate at the time of combining the selected carrier signals is uniformly performed. However, the distribution rate is changed according to the strength level. It may be. For example, in FIG. 3A, by selecting a combined distribution ratio of carrier signals of frequencies ω0, ω0 + ωc, and ω0 + 3ωc to 2: 1: 3, it is possible to select a highly reliable received signal according to the characteristics of the transmission path. And can be synthesized. Further, it may be a method of performing synthesis at an equal ratio to all received signals without providing a threshold value. Alternatively, in FIG. 3A, the frequency diversity effect can be obtained by setting the combined distribution ratio of the carrier signals of frequencies ω0, ω0 + ωc, ω0 + 2ωc, and ω0 + 3ωc to 3: 2: 1: 4.

Next, when the relative phase signals f _{1 to} f ₄ are used as parameters, it is determined whether or not any noise or distortion is received by the relative phase signals f _{1 to} f ₄ . In this determination, a threshold value (relative phase range) of an arbitrary relative phase is provided in advance, and whether or not the relative phase signals f _{1 to} f ₄ are within the threshold value (shaded portion in FIG. 4). Do it. For example, the transmission signal is subjected to transmission path distortion in the transmission path 3, and the relative phase signals f _{1 to} f ₄ (indicated by ● in FIG. 4) of the respective carrier signals detected by the DFT processing units 21 to 24. 4, when there is an individual phase fluctuation, the relative phase signals f _{1 to} f ₄ exceeding the threshold value γ ₁ -γ ₂ in the predetermined phase range are output. With respect to the carrier signal, it can be determined that there is no problem with the influence of the transmission path distortion. Therefore, in the case of FIG. 4, in the selection control means 40, the carrier signals ω0 + 2ωc exceeding the threshold value γ ₁ −γ ₂ are excluded and the carrier signals ω0, ω0 + ωc and ω0 + 3ωc are selected, and these three carrier signals are selected. The received signals are synthesized by the receiving synthesis means 30 and output as detection data.

Even when the relative phase signals f _{1 to} f ₄ are used as parameters, as described in the description of the method using the absolute value signals a _{1 to} a ₄ as parameters, the synthesis ratio of the carrier signals is not uniform. You may change and synthesize | combine.

Further, when both the absolute value signals a _{1 to} a ₄ and the relative phase signals f _{1 to} f ₄ are used as parameters, some noise is caused by both the absolute value signals a _{1 to} a ₄ and the relative phase signals f _{1 to} f _4. It is judged whether or not it is subjected to distortion. That is, both the upper and lower threshold values α and β of the intensity levels of the arbitrary absolute value signals a _{1 to} a ₄ and the threshold values γ _{1 to} γ ₂ of the relative phase range described above are provided in advance. In addition, it is determined whether both the intensity level and the relative phase signals f _{1 to} f ₄ are within the threshold value (the hatched portion in FIG. 5), and only carrier signals that satisfy this condition are subject to synthesis. To do.

Therefore, in the case of FIG. 5, the selection control means 40 excludes the carrier signal having the frequency ω0 + 3ωc by the threshold value (in FIG. 5, the intensity level is represented by the distance from the central intersection to the mark ●. ), The carrier signal of the frequency ω0 + 2ωc is excluded by the threshold values of the relative phase signals f _{1 to} f ₄ , the frequency ω0 and the carrier signal of ω0 + ωc are selected, and the received signals of these two carrier signals are received and synthesized. The data is synthesized by means 30 and output as output data.

  As described above, in the communication system according to the first embodiment of the present invention, the received signal in the band where the signal power is attenuated by transmission distortion and the SNR (signal to noise ratio) is deteriorated is combined. The overall SNR can be improved by excluding in stages or by reducing the synthesis rate in the synthesis process. Further, by combining the data transmitted by a plurality of carrier signals by the selection control means 40, a frequency diversity effect can be obtained and the influence due to narrow band noise can be reduced.

In the communication system according to the first embodiment of the present invention, the DFT processing units 21 to 24 are used to detect the absolute value signals a _{1 to} a ₄ and the angle of each carrier signal. The detection can be similarly performed even when a narrow band BPF is used instead of the processing units 21 to 24. In the description of the communication system according to the first embodiment of the present invention, the PSK-modulated data is used. However, the modulation method is not limited to this, and ASK (amplitude shift keying) modulation or DPSK modulation is performed. Even if it is made data, it can be similarly implemented.

  Furthermore, in the transmitter 1 of the communication system according to the first embodiment of the present invention, a plurality of multiplication units 11 and carrier signal generation units 12 are provided, and means for combining all the outputs of the plurality of multiplication units 11 is provided. Thus, it is possible to configure multiplex transmission for a plurality of input data.

(Second Embodiment)
FIG. 6 is a block diagram showing a communication system according to the second embodiment of the present invention.

  The selection control means 40 shown in FIG. 1 can be provided in the transmitter. That is, in FIG. 6, each output of the DFT processing units 21 to 24 is fed back to the selection control means 40 provided on the transmitter 1 side via the transmission path 3.

The selection signal Ss of the selection control means 40 sets a carrier signal to be generated by the carrier signal generator 12. If the transmitter side only needs to select the carrier signal to be transmitted using the absolute value signals a _{1 to} a ₄ in this way, and there is sufficient reliability, the receiver side needs to perform further selection processing. There is no. However, if sufficient reliability cannot be obtained, a carrier signal can be selected using the relative phase signals f _{1 to} f ₄ on the receiver side (see the broken line in FIG. 6).

  In addition, since the structure of those other than this in the 2nd Embodiment is the same as that of the said 1st Embodiment, it attaches | subjects the same reference number about the said structure below, and abbreviate | omits the description.

  In the second embodiment, the modulation data input to the multiplication unit 11 is the PSK modulation data shown in FIG. Further, the carrier signal generator 12 outputs a plurality of carrier signals corresponding to the impulse response of the transfer function H (ω) given by the following equation (2) reflecting the control contents of the selection controller 40 as carrier signals. .

Here, the parameter in the formula (2) (Al to A4) is a value based on the absolute value signal a ₁ ~a ₄ corresponding to each carrier signal fed back from the receiver 2, thus, the parameters ( A1 to A4) reflect the state of the transmission path.

  In the second embodiment, A1 = A2r = A3 = A4 = 1 is given as an initial parameter. Therefore, in this initial state, the impulse waveform of the transfer function H (ω) is the same as the state shown in FIG. 2B, and the transmission signal output from the multiplier 11 is shown in FIG. It has the same spectrum waveform as the state.

In the following, the description will be focused on the part in which the second embodiment performs processing different from that of the first embodiment. The DFT processing units 21 to 24 on the receiver side detect the absolute value signals a _{1 to} a ₄ and the angle signals b _{1 to} b ₄ corresponding to the carrier signals, as in the case of the first embodiment. Then, the absolute value signals a _{1 to} a ₄ are fed back to the selection control means 40 on the transmitter side via the transmission line 3, and the angle signals b _{1 to} b ₄ are input to the relative phase detection circuit 25.

The selection control means 40 receives the absolute value signals a _{1 to} a ₄ corresponding to the carrier signals, determines the intensity level, and generates parameters (A _{1 to A 4} ) based on the determination result.

Specifically, for example, the selection control means 40 causes the intensity levels of the absolute value signals a _{1 to} a ₄ of each carrier signal fed back from the receiver to be as shown in FIG. , The parameters A2 and A3 corresponding to the carrier signals of the frequencies ω0 + ωc and ω0 + 2ωc below the threshold α are set to “0”, and the parameters A1 and A4 corresponding to the carrier signals of the frequencies ω0 and ω0 + 3ωc are set. The carrier signal generator 12 is controlled so as to “1”. By this control, the impulse response of the transfer function H (ω) output from the carrier signal generator 12 is as shown in FIG. 7B, and therefore the spectrum waveform of the signal transmitted from the multiplier 11 after the control is As shown in FIG. As a result, data can be transmitted while avoiding the use of carrier signals with transmission path distortion.

As described above, in the communication system according to the second embodiment of the present invention, the absolute value of each carrier signal reflecting the characteristics of the transmission path is fed back from the receiver 2 to the transmitter 1, and the absolute value signal a ₁ using ~a _4, since the signal power by the transmission channel distortion transmitter 1 is not performed data transmission in the band to deteriorate the SNR attenuated, thereby improving the SNR of the entire transmission signal.

  In the above description, only the case where the lower threshold value α is used has been described. However, when the upper threshold value β is used as shown in the first embodiment, or both the upper and lower threshold values are set. It can also be applied when used.

In the second embodiment of the present invention, the absolute value signals a _{1 to} a ₄ corresponding to each carrier signal are used as signals for feedback to the selection control means 40 to control the carrier signal. Even when the relative phase signals f _{1 to} f ₄ corresponding to the respective carrier signals are used, the same effect can be obtained, and furthermore, both the absolute value signals a _{1 to} a ₄ and the relative phase signals f _{1 to} f ₄ are used. Of course, the same effect can be obtained.

In the parameter control performed by the selection control means 40, “0” is set in the parameter corresponding to the absolute value signals a _{1 to} a ₄ which are below the threshold value. For example, A 1 = 1.2, A value corresponding to the values of the absolute value signals a _{1 to} a ₄ may be set such as A 2 = 0.5, A 3 = 0.5, and A 2 = 1.2.

Further, in the second embodiment of the present invention, as described above, the receiver side is also provided with the selection control means 40 ′, and the selection control means 40 on the transmitter side uses the absolute value signals a _{1 to} a _4. If so, the selection control means 40 ′ on the receiver side may perform the carrier signal selection control process using the relative phase signals f _{1 to} f ₄ .

(Third embodiment)
FIG. 8 is a block diagram showing a communication system according to the third embodiment of the present invention.

  As can be seen from a comparison between FIG. 6 and FIG. 8, in the third embodiment, the selection control means 40 provided in the transmitter 1 of the second embodiment is provided in the receiver 2. Is different. In addition, since the structure of those other than this in the 3rd Embodiment is the same as the structure in the said 2nd Embodiment, the description is abbreviate | omitted.

  As described above, in the communication system according to the third embodiment of the present invention, the selection control means 40 is configured on the receiver side, and the (parameter) signal for controlling the carrier signal generator 12 is fed back to the transmitter side. I am doing so. For this reason, in addition to the effect which the communication system which concerns on the said 2nd Embodiment show | plays, the structure of the feedback transmission path 3 becomes easy.

(Fourth embodiment)
FIG. 9 is a block diagram showing a communication system according to the fourth embodiment of the present invention.

  In FIG. 9, the transmitter 1 and the receiver 2 are connected via two transmission paths, a transmission path 3 from the transmitter 1 to the receiver 2 and a transmission path 6 from the receiver 2 to the transmitter 1. Has been.

  The transmitter 1 includes two multiplication units 11a and 11b, two carrier signal generation units 12a and 12b, two selection control units 40a and 40b, and a transmission synthesis unit 14. The receiver 2 includes four DFT processing units 21 to 24, a relative phase detection circuit 25, and a reception synthesis unit 30.

  Here, as can be seen by comparing FIG. 6 and FIG. 9, the transmitter 1 of the fourth embodiment includes two systems of the configuration of the transmitter 1 of the second embodiment, and further includes a transmission combining unit. 14 is different. The receiver 2 of the fourth embodiment is basically the same as the receiver 2 of the second embodiment, but the reception combining unit 30 of the fourth embodiment is a receiver. The detection data is output by the number of modulation data (input data of the transmitter 1) input to 7. The other configurations in the fourth embodiment are the same as the configurations in the first and second embodiments. Therefore, the same reference numerals are assigned to the configurations and the description thereof is omitted. .

Absolute value signals a _{1 to} a ₄ corresponding to carrier signals obtained by the DFT processing units 21 to 24 in the receiver 2 are input to the selection control means 40 a and 40 b via the transmission path 3. Then, the selection control unit 40a and 40b, respectively controls the carrier signal generating unit 12a and 12b based on the absolute value signal a ₁ ~a _4. Carrier signal generators 12a and 12b generate carrier signals based on the above control, and input the carrier signals to multipliers 11a and 11b.

  The multiplier 11a multiplies each carrier signal input from the carrier signal generator 12a and the first modulated data and outputs the result, and the multiplier 11b outputs each carrier signal output from the carrier signal generator 12b. The carrier signal and the second modulated data are multiplied and output. Further, the transmission combining unit 14 combines the outputs of the multiplying units 11 a and 11 b and outputs the combined result to the transmission path 3.

  As described above, the fourth embodiment has two systems of the receiver 4 of the second embodiment, so that a plurality of carrier signals are assigned to two different modulation signals (input signals), The carrier signal is modulated with the two input data and transmitted simultaneously. However, in the following description, four carrier signals are assumed to be configured with the same circuit scale as in the first and second embodiments, but the transmission quality is equivalent to that in each of the above embodiments. In order to achieve this, it is preferable to allocate four carrier signals for each of the two different modulation data.

  Also, the number of modulation data to be input is not limited to two. If a number of multiplication units, carrier signal generation units, and selection control means corresponding to the number of modulation data to be input are provided, even more inputs can be provided. Can respond.

  In the fourth embodiment, the first modulation data input to the multiplication unit 11a is the PSK modulation data shown in FIG. 10A, and the second modulation data input to the multiplication unit 11b is The PSK modulation data shown in FIG. The carrier signal generator 12a outputs a plurality of carrier signals corresponding to the impulse response of the transfer function Ha (ω) given by the following equation (3) reflecting the control content of the selection control means 40a as the carrier signal. The carrier signal generator 12b outputs a plurality of carrier signals corresponding to the impulse response of the transfer function Hb (ω) given by the following equation (4) reflecting the control content of the selection control means 40b as the carrier signal.

Here, Al to A4 and B1~B4 are parameters obtained based on the absolute value signal a ₁ ~a ₄ fed back from the receiver 2. In the fourth embodiment, the parameters set in the carrier signal generator 12a are A1 = A2 = 1 and A3 = A4 = 0, and the parameters set in the carrier signal generator 12b are B3 = B4. = 1 and B1 = B2 = 0.

  Therefore, in this initial state, the impulse waveforms (carrier signals) of the transfer functions Ha (ω) and Hb (ω) are in the states shown in FIGS. 10 (c) and 10 (d), respectively. The output transmission signal exhibits the spectrum waveform shown in FIG.

In the following, description will be made centering on the part in which the fourth embodiment performs processing different from the first and second embodiments. Each DFT processing unit 21 to 24 detects the absolute value signals a _{1 to} a ₄ and the angle signals b _{1 to} b ₄ corresponding to the carrier signals as in the case of the second embodiment, and the absolute value The signals a _{1 to} a ₄ are fed back to the selection control means 40 a and 40 b of the transmitter 1 through the transmission line 3, and the angle signals b _{1 to} b ₄ are input to the relative phase detection circuit 25.

Absolute value signal a ₁ ~a feedback the received selection control means 40a of ₄ as described above, to determine the intensity level for the first modulated data undergoing the absolute value signal a ₁ ~a _4, the determination result The parameters (A1 to A4) are controlled based on the above. The selection control unit 40b determines the intensity level for the second modulated data undergoing the absolute value signal a ₁ ~a _4, controlling parameters (B1 to B4) based on the determination result.

Basically, the selection control unit 40a and 40b, and selects the same process relating to the absolute value signal a ₁ ~a ₄ in the selection control means 40 of the first embodiment. That is, for example, when the intensity levels of the absolute value signals a _{1 to} a ₄ corresponding to the carrier signals are individually attenuated as shown in FIG. The means 40a controls the carrier signal generator 12a so that the parameter A2 for the carrier signal of the frequency ω0 + ωc below the threshold is set to “0” together with A3 and A4, and the selection control means 40b is the frequency below the threshold. The carrier signal generator 12b is controlled so that the parameter B3 for the carrier signal of ω0 + 2ωc is set to “0” together with B1 and B2 (in FIG. 11, the signal intensity level relating to the first modulation data is indicated by the white arrow. The signal intensity level for the second modulation data is indicated by hatched arrows). By this control, the spectrum of the signal transmitted from the transmission synthesizer 14 becomes as shown in FIG. 11B, and two information signals can be transmitted simultaneously while avoiding a carrier signal having a transmission path distortion. it can.

  In the above case, if the parameters A1 and A4 are set to "1" and the parameters A2, A3 and B1 to B4 are set to "0" in the first carrier signal generator 12a, the carrier signal ω0 + 3ωc is Instead, the first modulation data is superimposed, as shown in FIG. In this case, only the information of the first modulation data can be transmitted, but the influence of the narrowband noise on the first modulation data can be reduced by the frequency diversity effect.

  As described above, the communication system according to the fourth embodiment of the present invention has a spectrum when performing multiple transmissions for transmitting a plurality of independent data or performing high-speed data transmission by dividing data into a plurality of carrier signals. If there is a band in which the signal power is attenuated due to transmission path distortion and the SNR deteriorates by providing the above-described configuration for spreading for each data and feeding back the transmission path characteristics from the receiver 2 to the transmitter 1. By reducing the number of multiplexed data or reducing the data rate corresponding to the degraded band, it is possible to improve the SNR and realize high quality data transmission.

In the fourth embodiment of the present invention, as in the case of the second embodiment, the absolute value signals a _{1 to} a ₄ are signals that are fed back to the selection control means 40a and 40b to control the carrier signal. However, the same effect can be obtained even when the relative phase signals f _{1 to} f ₄ are used instead or when both are used.

  In the above description, the parameter corresponding to the carrier signal that falls below the set threshold is set to “0” so that an arbitrary carrier signal is not used. However, the usage distribution of each carrier signal is changed. The parameters may be controlled.

In the above description, only the lower limit threshold value is used. However, it goes without saying that an upper limit threshold value or an upper and lower limit threshold value may be used. Of course, the phase signals f _{1 to} f ₄ shown may be used together.

  Furthermore, the selection control means 40 ′ is also provided on the receiver side, and further selection processing is performed in parallel using the output of the relative phase detection circuit 25, thereby enabling higher quality data transmission.

(Fifth embodiment)
FIG. 12 is a block diagram showing a communication system according to the fifth embodiment of the present invention. As can be seen from a comparison between FIG. 9 and FIG. 12, in the fifth embodiment, the selection control means 40a and 40b that were configured in the transmitter 1 of the fourth embodiment are configured in the receiver 2. The point is different. In addition, since the structure of those other than this in the 5th Embodiment is the same as that of the said 4th Embodiment, the description is abbreviate | omitted.

  As described above, in the communication system according to the fifth embodiment of the present invention, the selection control means 40a and 40b are configured in the receiver 2, and the (parameter) signal for controlling the carrier signal generators 12a and 12b is transmitted to the transmitter. 2 is fed back. For this reason, in addition to the effect which the communication system which concerns on the said 4th Embodiment show | plays, the structure of the feedback transmission path 3 becomes easy.

(Sixth embodiment)
Although the case where the input signal is multiplied by the carrier signal has been described above, the carrier signal can be directly extracted from the input signal using a filter. In this case, in addition to extracting signals that do not interfere with each other (hereinafter referred to as orthogonal) on the frequency axis with a plurality of filters having center frequencies at predetermined intervals on the frequency axis, the input signal is also extracted on the time axis. It is more effective if there is no interference (also referred to as orthogonal) between the preceding signal and the succeeding signal.

  FIGS. 13 and 14 are basic configuration diagrams of a communication system using overlapping orthogonality that is orthogonal on both the frequency axis and the time axis. In the transmitter 1, the input signal is input to the encoder 50, where the number of input divided signals corresponding to the filters 52a to 52d in the next stage is generated. Here, in order to make the explanation easy to understand, it is assumed that the input signal is a digital “1”, but it is needless to say that a modulation signal as shown in the first to fifth embodiments may be used.

  The signals generated as described above are further upsampled into a plurality of pieces within the same bit rate by the upsampling means 51a to 51d. Specifically, a plurality (M−1) of “0” s are inserted after the input “1”. The number of “0” s to be inserted is not particularly limited. For example, when 3 (and therefore M = 4) “0” s are inserted, the time taken for 1 bit rate of “1” corresponding to the input signal is ¼. It becomes. Therefore, the larger the number of inserted “0” s, the shorter the time taken for 1 bit rate of “1”. In order to reduce the influence of noise as will be described later, the number of inserted “0” s Is more preferable.

The input signal up-sampled in this way is input to a plurality (here, 4) of filters 52a to 52d whose center frequency takes a value at a predetermined interval. Here, the impulse response for each sample of the filter of this transmitter is f _i (n), {i = 1, 2, .. M} (10)
The design conditions of the filter can be expressed by

L: Number of filter taps i
i ₂ : suffix indicating an arbitrary carrier signal l: overlap coefficient M; number of samples per bit of data A; integer of A> 0 δ: delta function is orthogonal on both time axis and frequency axis as described above A filter that satisfies the conditions can be obtained. The above equation (11) is either when not overlap with l = 0 i.e. other samples, i ₁ = i ₂ That will have a value when that does not overlap with the carrier signal of another frequency, the other case of "0" As a result, a carrier signal having an infinitesimal time band and frequency band is formed.

  The outputs of the filters 52a to 52d designed in this way are combined by the combining means 53 and transmitted via the transmission path, whereby the transmission signal is a carrier signal having a predetermined interval value on the frequency. It becomes.

The transmission signals transmitted in this way are the same number of filters as the filters 52a to 52d of the transmitter 1 provided in the receiver 1, and the center frequencies of the filters correspond to the filters 52a to 52d, respectively. Are input to the filters 62a to 62d. Here, the impulse response to each sample of each filter of the receiver 2 is h _i (n), {i = 1, 2, .. M} (21)
The filter design conditions are expressed by the following 22 equations.

H _i (z) = z- ^(l-1) F _i (z ^-1 ) (22)
That is, if the outputs from the filters 62a to 62d are designed to be the same signal delayed on the time axis as the inputs of the filters 52a to 52d, the filters 62a to 62d satisfy the following orthogonal condition. Become.

  Thus, the output is not affected unless the noise is generated at the center frequency of each of the filters 62a to 62d and at the same time (in the above example, the first quarter time of one bit rate). Become.

  The outputs of the filters 62a to 62d of the receiver 2 are inputted to the down-sampling means 61a to 61d, and the data corresponding to the samples added by the up-sampling means 51a to 51d at the time of transmission is removed, and a decoder which is a reception synthesis means A signal necessary for synthesis (an angle signal, an absolute value signal, or an actual signal output from the down-sampling means 61a to 61d) is input to 60.

On the other hand, the outputs of the down-sampling means 61a to 61d are led to a conversion means 70 for detecting absolute value signals a _{1 to} a ₄ and angle signals b _{1 to} b _{4, and} are shown in the _first to fifth embodiments. Similarly to the DTF processing units 21 to 24, the absolute value and angle of each frequency component of the transmission signal are detected, and the angle signals b _{1 to} b ₄ are further input to the relative phase detection means 25 to detect the relative phase. The Here, the absolute value signals a _{1 to} a ₄ and / or the relative phase signals f _{1 to} f ₄ thus detected are input to the selection control means 40 provided in the transmitter 1 or the receiver 2. As in the first to fifth embodiments, it is used for transmission signal control or reception signal control.

Here, in the example shown in FIG. 13, both of the absolute value signals a _{1 to} a ₄ and the relative phase signals f _{1 to} f ₄ are used, and the signal selection means 40 selects a carrier or assigns a distribution ratio. As a result, the decoder performs a decoding process based on the result.

Further, in the example shown in FIG. 14, the absolute value signals a _{1 to} a ₄ from the converting means 70 are fed back to the encoder 50 of the transmitter 1, and the carrier is selected by the encoder 50 or the distribution ratio is given. It will be. Also in this case, it goes without saying that the selection control process may be further performed on the receiver side using both of the relative phase signals f _{1 to} f ₄ from the relative phase detection means 25.

  In the above, when there are a plurality of input signals to the transmitter 1, the encoder 50 determines which of the filters 52a to 52d is assigned to the plurality of input signals. This eliminates the need to have a number of processing circuits corresponding to the number of input signals as shown in FIG.

  As described above, according to the present embodiment, transmission signals that do not interfere with each other on the time axis and the frequency axis can be formed. Therefore, as long as noise that matches both time (sample time) and frequency is not generated in the transmission path, the filter 62a. The influence of noise does not appear in the output of ˜62d, and extremely high transmission efficiency can be obtained. Furthermore, even if the influence of noise appears in the outputs of the filters 62a to 62d, the selection control means 40 can remove the signal of the affected frequency.

  The present invention improves transmission efficiency and transmission accuracy by using a plurality of carriers orthogonal to each other on the frequency axis, and actively uses a frequency band that is not affected by narrowband noise or transmission path distortion. Therefore, high-speed and high-quality data transmission can be performed, which is useful for a communication system through a high-noise line such as an electric light line.

The block diagram which shows the structure of the communication system which concerns on the 1st Embodiment of this invention. The figure which shows an example of the transfer function of a carrier signal and the spectrum waveform of data in the receiver of FIG. The figure which shows an example of the selection process of the carrier signal by the absolute value signal performed by the reception synthetic | combination means of FIG. The figure which shows an example of the selection process of the carrier signal by the relative phase signal performed with the selection synthetic | combination circuit of FIG. The figure which shows an example of the selection process of the carrier signal by both the absolute value signal and relative phase signal which are performed by the selection synthetic | combination circuit of FIG. The block diagram which shows the structure of the communication system which concerns on the 2nd Embodiment of this invention. The figure which shows an example of the selection process of the carrier signal by the absolute value signal performed by the selection control means and carrier signal generation part of FIG. The block diagram which shows the structure of the communication system which concerns on the 3rd Embodiment of this invention. The block diagram which shows the structure of the communication system which concerns on the 4th Embodiment of this invention. The figure which shows an example of the transfer function of a carrier signal and the spectrum waveform of data in the receiver of FIG. The figure which shows an example of the selection process of the carrier signal by the absolute value signal performed in the selection control means and carrier signal generation part of FIG. The block diagram which shows the structure of the communication system which concerns on the 5th Embodiment of this invention. The block diagram which shows the structure of the communication system which concerns on the 6th Embodiment of this invention. The block diagram which shows another structure of the communication system which concerns on the 6th Embodiment of this invention. Block diagram showing an example of the configuration of a conventional data transmission system The figure which shows an example of the spectrum waveform in the conventional transmitter and receiver

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Transmitter 2 ... Receiver 3 ... Transmission path 11, 11a, 11b ... Multiplication part 12, 12a, 12b ... Carrier signal generation part 14 ... Transmission composition part 21-24 ... DFT process part 25 ... Relative phase detection circuit 30 ... Selection synthesis circuit 40a, 40b ... selection control unit

Claims (5)

A transmitter that performs communication using all or part of a plurality of carriers orthogonal to each other on a frequency axis,
A control signal receiving unit that receives a control signal including information indicating a carrier that is not used for communication among the plurality of carriers transmitted by the receiving device;
A transmission apparatus comprising: a transmission unit that generates a transmission signal without performing modulation by data for a carrier that is not used for communication specified by the control signal, and transmits the transmission signal to the reception apparatus. A receiver that performs communication using all or part of a plurality of carriers orthogonal to each other on a frequency axis,
A control signal transmission unit that transmits a control signal including information indicating a carrier not used for communication among the plurality of carriers to a transmission device;
A receiving device comprising: a receiving unit that receives a signal that is not used for communication specified by the control signal and is not modulated by data. A transmission method in which a transmission device communicates with a reception device using all or part of a plurality of carriers orthogonal to each other on a frequency axis,
A control signal transmitted from a receiving device and including information indicating a carrier not used for communication among the plurality of carriers;
A transmission method for generating a transmission signal for a carrier not used for communication specified by the control signal without performing modulation by data, and transmitting the transmission signal to the receiving apparatus. A receiving method in which a receiving apparatus communicates with a transmitting apparatus using all or a part of a plurality of carriers orthogonal to each other on a frequency axis,
A control signal including information indicating a carrier not used for communication among the plurality of carriers is transmitted to the transmission device;
A receiving method for receiving a signal generated without modulation by data for a carrier not used for communication specified by the control signal. A signal generation device used for a transmitter that performs communication using all or part of a plurality of carriers orthogonal to each other on a frequency axis,
Based on information indicating a carrier that is not used for communication among the plurality of carriers included in the control signal transmitted from the receiver, a carrier that is not used for communication specified by the control signal is not modulated by data. A signal generation device including a signal generation unit that generates a signal to be transmitted to the receiver.

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