JP2002238257A - Control method for resonance dc-dc converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a control method for resonance DC-DC converter capable of maintaining zero-voltage switching of a switching device even if an auxiliary resonance circuit is not connected during low load current of light load and attaining high efficiency because no connection of the auxiliary resonance circuit eliminates loss resulting from the auxiliary resonance circuit. SOLUTION: In this control method for resonance DC-DC converter consisting of a resonance full-bridge inverter, a high-frequency transformer 6, and a rectifying circuit; the switching interval of IGBT3a to 3d is changed so as to complete charging and discharging of snubber capacitors 5a to 5d at the time of switching IGBT3a to 3d of the full-bridge inverter.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance circuit in which a full-bridge inverter is connected to a primary side of a transformer and a rectifier circuit for outputting a required DC voltage to a load is connected to a secondary side of the transformer. BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method of controlling a resonant DC-DC converter capable of suppressing switching loss of a switching element and noise generated by switching.

[0002]

2. Description of the Related Art In recent years, power electronics have increased harmonics in power systems. In an industrial DC power supply device, research and development of a high power factor converter and a PWM converter are being performed to suppress this. Further, in order to suppress electromagnetic induction noise generated from a power supply device, research has been conducted on noise reduction by applying a soft switching technology for performing zero voltage switching (ZVS) or zero current switching (ZCS) of a switch element.

[0003] In the field of DC-DC converters as well, practical research has been conducted to improve characteristics by applying soft switching technology. By the soft switching, not only the noise but also the switching loss of the switching element can be significantly reduced, so that the efficiency of the converter can be improved.

As a typical DC-DC converter circuit system, there is a primary-side phase shift PWM control system in which a DC output voltage is controlled on the primary side of a transformer. FIG. 9 shows a conventional circuit of a primary-side phase shift PWM control DC-DC converter.

In FIG. 9, a primary-side full-bridge inverter has an IGBT for each arm to which a DC power supply 1 is input.
3a to 3d, reverse conducting diodes 4a to 4f, snubber capacitors 5a to 5d, a resonance reactor 7a, and resonance IGBTs 13a to 13b. The output of the primary side full-bridge inverter is A rectifier circuit including a rectifier diode 8a to 8d, a reactor 10 and a capacitor 11 is connected to a primary side and a rectifier diode 8a to 8d is connected to a secondary side. An output of the rectifier circuit is connected to a load resistor 12.

The detailed circuit operation is described in “An Improved Full-B
ridge Zero-Voltage-Transition PWM DC / DC Converter
with Zero-Voltage / Zero-Current Switching of the Au
xiliary Switches ”: IEEE IAS Vol36, No.2, March / April
2000, pp558-566, etc. Resonant type D shown in FIG.
In the C-DC converter, the ON widths of all the IGBTs 3a to 3d of the primary side full-bridge inverter are the same, and the IGBT switching interval of each arm is the same between the two arms in a short period equivalent to the dead time. Between two pairs of IGBTs between the two arms (between the IGBT 3a and the IGBT 3d,
By changing the ON overlap width of the IGBT 3b and the IGBT 3c), the voltage value of the DC output, which is the output of the full-bridge inverter, can be controlled.

FIG. 10 shows a state of commutation between the IGBT 3a and the IGBT 3b. Since the load current cannot be obtained when the load resistance 12 is lightly loaded, the snubber capacitors 5a and 5b are connected to the IGBTs 3a to 3b during commutation as shown in FIG.
b becomes impossible to discharge or charge for zero voltage switching within the dead time period. Therefore, as shown in FIG. 10B, the IGBTs 3a and 3b are turned on with the voltage remaining in the snubber capacitors 5a and 5b, and this turn-on is hard switching.

On the other hand, considering the resonance type DC-DC converter shown in FIG. 10 in terms of efficiency, when the load resistor 12 has a light load, a loss occurs due to hard switching of the IGBTs 3a to 3d, and the efficiency at the time of light load is reduced. become.

As another circuit system, there is a secondary side phase shift PWM control system in which output voltage control is performed on the secondary side of the transformer.

FIG. 11 shows a secondary phase shift PWM control DC.
1 shows a conventional circuit of a DC converter. The secondary phase shift PWM control DC-DC converter shown in FIG. 11 includes DC voltage dividing capacitors 2a to 2d,
GBTs 3a to 3d, reverse conducting diodes 4a to 4d, snubber capacitors 5a to 5d, and resonance reactors 7a to 7
b, a high-frequency transformer 6 receiving the output of the full-bridge inverter on the primary side,
Rectifier diodes 8a to 8d, IGBT for secondary phase control
It comprises a rectifier circuit 9a to 9b, a reactor 10 and a capacitor 11 for giving a DC output to a load resistor 12.

The detailed circuit operation is described in Michihira et al .: “New high-frequency translink soft switching PWM DC-D
C Converter and Open Loop Characteristics ", IEEJ Transactions on Digital Science, Vol. 116, No. 5, pp. 546-555, 1996.

In the resonance type DC-DC converter shown in FIG. 11, the primary-side full-bridge inverter has no phase between arms and is symmetrically driven at 50% duty including dead time. The resonance reactors 7a to 7b are IGB
A current is supplied from the DC power supply 1 to ensure zero voltage switching of T3a to 3d.

When the resonance reactors 7a to 7b are not connected, the load current cannot be obtained when the load resistor 12 is lightly loaded, so that the above-mentioned primary side phase shift PWM control D
As in the case of the C-DC converter, the snubber capacitors 5a to 5d cannot discharge or charge for zero voltage switching in the snubber capacitors 5a to 5d during the dead time during the commutation. Therefore, the IGBTs 3a to 3d are turned on with the voltage remaining in the snubber capacitors 5a to 5d, and the turn-on is hard switching.

Therefore, resonance reactors 7a to 7d are supplied so that a constant current is supplied when IGBTs 3a to 3d are turned off.
7b is connected. Thereby, even at the time of light load, the snubber capacitors 5a to 5d are charged or discharged for a certain period of time, and commutation by zero voltage switching is secured.

On the other hand, considering the resonance type DC-DC converter shown in FIG. 11 from the viewpoint of efficiency, a resonance current always flows through a DC voltage dividing capacitor → resonance reactor → IGBT or a loop of a reverse conduction diode resonance circuit. For,
The loss due to this resonance circuit lowers the efficiency.

Further, when a sufficient load current can be obtained as in the case of a heavy load, the resonance current of the element current of the load current plus the resonance current in the IGBTs 3a to 3d becomes an excess current. For this reason, conduction loss of the IGBT is increased, and the efficiency is reduced.

[0017]

SUMMARY OF THE INVENTION It is an object of the present invention to provide a resonance type DC-DC converter which can maintain zero voltage switching of a switching element without connecting an auxiliary resonance circuit at light load and low load current. It is still another object of the present invention to provide a control method capable of achieving high efficiency because the loss due to the auxiliary resonance circuit is eliminated because the auxiliary resonance circuit is not connected.

[0018]

In order to achieve the above object, the present invention comprises a resonance circuit for connecting a soft switching capacitor in parallel to a semiconductor switching element to which a DC power supply is input, and for charging and discharging the capacitor. A resonant full-bridge inverter for converting the DC power into an AC output;
A method of controlling a resonance type DC-DC converter, comprising: a transformer received on a secondary side; and a rectifier circuit for rectifying a secondary side output of the transformer to DC and outputting a predetermined DC voltage to a load. When switching the switching element, the switching interval of the semiconductor switching element is changed so that the charging and discharging of the capacitor is completed.

In the method of controlling the resonant DC-DC converter according to the present invention, the charging and discharging time of the capacitor can be secured even at a low current under a light load, so that zero voltage switching of the semiconductor switching element is achieved. it can. In particular, the efficiency at light load is improved.

Further, in order to achieve the above object, the present invention provides:
A resonant full-bridge inverter for connecting a soft switching capacitor in parallel to a semiconductor switching element having a direct-current power supply as input and for converting the direct-current power to an alternating-current output by providing a resonant circuit for charging and discharging the capacitor; A transformer receiving the AC output of the transformer on the primary side, a rectifier circuit for rectifying the secondary side output of the transformer to DC and outputting a predetermined DC voltage to the load, A resonance type DC-DC converter including a semiconductor switching element inserted in a direction path, wherein when the semiconductor switching element of the full-bridge inverter on the primary side of the transformer is switched, charging and discharging of the capacitor is performed. The semiconductor switching element of the full bridge inverter on the primary side of the transformer to be completed And wherein the changing the switching interval.

In the control method of the resonance type DC-DC converter according to the present invention, the capacitor connected in parallel to the semiconductor switching element in the primary full-bridge inverter of the transformer even at a low current under a light load. Of the semiconductor switching element can be achieved. Since the auxiliary resonance reactor for charging and discharging the capacitor is not required, the loss due to the auxiliary resonance circuit is eliminated, and the efficiency is improved.

[0022]

FIG. 1 is a control block diagram of a resonance type DC-DC converter according to a first embodiment of the present invention.

The resonance type DC-DC converter of this embodiment comprises a full-bridge inverter, a high-frequency transformer 6, and a rectifier circuit. As shown in FIG. 1, the output of a full-bridge inverter including IGBTs 3a to 3d, reverse conducting diodes 4a to 4d, and snubber capacitors 5a to 5d for each arm to which the DC power supply 1 is input is supplied to the primary side of the high-frequency transformer 6. Receiving, rectifying diodes 8a to 8d on the secondary side,
A rectifier circuit including a reactor 10 and a capacitor 11 is connected, and an output of the rectifier circuit is connected to a load resistor 12.

In the switching of the IGBTs 3a to 3d, first, as step 100, the duty of the switching frequency
The reference gate pulses of the IGBTs 3a to 3d are generated from 50% of the reference pulses.

Next, in step 101, the overlap width of the left arm and the right arm is determined by DC output voltage control, and the reference gate pulse is shifted so that the left arm and the right arm have this overlap width. Thus, the reference gate pulse of each of the IGBTs 3a to 3d is determined.

When the switching of the IGBTs 3a to 3d is performed in each arm, the snubber capacitors of the IGBTs turned off by the zero voltage switching are charged and the snubber capacitors of the other IGBTs in the same arm are discharged in steps 102A and 102B. At this time, the voltage of each snubber capacitor is observed, and it is confirmed that the voltage of the snubber capacitor of the IGBT to be turned on becomes 0 V or almost 0 V, and the discharge of the snubber capacitor is completed.

If the load current is sufficiently large and the charging / discharging is completed within the dead time as in the case of a heavy load, the reference gate pulse is directly outputted to steps 104A and 104B and step 1
05A and 105B are output to the gate circuit.

When the load current is small as in the case of a light load, the charge / discharge time is extended.
B and steps 104A and 104B, step 1
03A and 103B to extend the switching interval,
Delay turning on the GBT. During this time, the switching state of the IGBT of another arm is extended by the switching interval in order to secure the overlap width after switching.

FIG. 2 shows this state. At light load,
Since it takes time to discharge the snubber capacitor 5a of the IGBT 3a, the switching interval is extended. When the voltage of snubber capacitor 5a becomes 0V, IGBT 3a
Turn on. At this time, the IGBT turned on by another arm
The ON width of 3c is extended by the switching interval. IGBT
Interval and IG of switching between 3c and IGBT 3d
This is because the overlap width between the BT 3a and the IGBT 3d is considered. However, even if the DC output voltage is the same, since the switching interval changes, the overlap width differs between a heavy load and a light load.

When the load current at the time of light load is very small, the charging and discharging of the snubber capacitor is completed with the voltage of the snubber capacitor balanced up and down by half of the DC power supply voltage. In this case, no matter how long the switching interval is extended, the snubber capacitor voltage of the IGBT that is turned on does not become 0 V, so that the IGBT cannot be turned on.

Therefore, a limiter is provided to set an upper limit for the switching interval, and when the switching interval is extended to a predetermined interval, the interval is not further extended. When the switching interval reaches the upper limit, the IGBT is turned on regardless of the snubber capacitor voltage value at that time. When the voltage remains in the snubber capacitor, the IGBT is turned on by hard switching.

As described above, in the control method according to the present embodiment, the charging and discharging time of the snubber capacitor can be secured even at a low current at a light load, so that zero voltage switching of the IGBT can be achieved, and particularly at a light load. Efficiency can be improved.

FIG. 3A is a control configuration diagram of a DC-DC converter according to a second embodiment of the present invention, and FIG. 3B is a diagram illustrating ON / OFF timing of the IGBT.

In FIG. 3A, reference numerals 3a to 3d denote IGBTs receiving the DC power supply 1, 4a to 4d denote reverse conducting diodes, 5a to 5d denote snubber capacitors, 6 denotes a high-frequency transformer, and 8a to 8d denote rectifying diodes. Reference numeral 10 denotes a reactor, 11 denotes a capacitor, 12 denotes a load resistance, and 14 denotes a current detector.

In this embodiment, step 1 in FIG.
In place of 01, as shown in FIG. 3A, a step 105 for determining the on-pulse width is provided. Then, the load current is detected by the current detector 14, and particularly when the load current becomes a predetermined current value or less at a light load, the DC voltage is not controlled by the overlap width between the arms, but the load between the arms is controlled. Are switched at the same time, and as shown in FIG. 3B, DC voltage control is performed with this ON width. This facilitates the control of the gate pulse when the switching interval of the IGBT changes.

As a result, in the control method of the present embodiment, zero voltage switching of the IGBT can be achieved at light load, and the efficiency at light load can be particularly improved.

FIG. 4 is a diagram showing a third embodiment of the present invention.
It is a control block diagram of a C-DC converter. 3a to 3d are IGBTs that receive the DC power supply 1, 4a to 4d are reverse conducting diodes, 5a to 5d are snubber capacitors, 6 is a high-frequency transformer, 8a to 8d are rectifying diodes, 10 is a reactor, 11 is a capacitor, and 12 is a capacitor. A load resistor, 14 is a current detector, 15 is a switching assisting reactor, and 16 is an open / close switch.

In this embodiment, the load current is detected by the current detector 14, and particularly when the load current becomes a predetermined current value or less at light load, the switching assist reactor 15 is switched by the switching operation of the open / close switch 16. An inductor is inserted between the primary terminals of the transformer 6 in parallel, and an inductance is inserted between the primary terminals of the transformer 6 in parallel. A current corresponding to the overlap width of the IGBT flows through this inductance regardless of the load current. With the current of this inductance, the charging and discharging of the snubber capacitors of each arm can be reliably completed even when the load current is small.

As described above, in the control method of the present embodiment, the charging and discharging time of the snubber capacitor can be secured even at a low current under a light load, so that zero voltage switching of the IGBT can be achieved. It is possible to improve efficiency.

FIG. 5 shows a fourth embodiment of the present invention.
It is a control block diagram of a C-DC converter. 3a to 3d, 1 is an IGBT receiving the DC power supply 1, 4a to 4d are reverse conducting diodes, 5a to 5d are snubber capacitors, 6 is a high-frequency transformer, 8a to 8d are rectifying diodes, 9a to 9b.
Is an IGBT for secondary phase control, 10 is a reactor, 11 is a capacitor, and 12 is a load resistance.

In the switching of the IGBT, first, as step 100, a reference gate pulse of the IGBT is generated from a reference pulse having a duty of 50% of the switching frequency. The primary-side full-bridge inverter simultaneously switches IGBT2 elements that form a pair between the two arms. The on-pulse width is the same as the duty width of the reference pulse. IG
When the BT is switched, the snubber capacitor of the IGBT turned off by the zero voltage switching is charged, and the snubber capacitors of the other IGBTs in the same arm are discharged. At this time, the voltage of each snubber capacitor is observed as steps 102A and 102B, and it is confirmed that the voltage of the snubber capacitor of the IGBT to be turned on becomes 0 V or almost 0 V, and the discharge of the snubber capacitor is completed. When charging and discharging are completed within the dead time when the load current is sufficiently large as in the case of heavy load, the reference gate pulse is output to the gate circuit as it is. When the load current is small as in the case of a light load, the charge / discharge time is extended.
The switching interval is extended by 103B, and the ON of the IGBT is delayed.

FIG. 6 shows this state. At light load,
Since it takes time to discharge the snubber capacitors of the IGBTs 3a and 3d, the switching interval is extended. When the snubber capacitor voltage becomes 0 V, IGB
Turn on T3a and IGBT3d.

When the load current at the time of light load is very small, the charging and discharging of the snubber capacitor is completed with the voltage of the snubber capacitor balanced up and down by half of the DC power supply voltage. In this case, no matter how long the switching interval is extended, the snubber capacitor voltage of the IGBT that is turned on does not become 0 V, so that it cannot be turned on. Therefore, step 1
For 03A, 103B to 105A, 105B, li
By providing a miter, an upper limit is set for the switching interval,
After extending to a predetermined interval, do not extend the interval any further. If the switching interval reaches the upper limit, IGB
T is turned on at that time regardless of the snubber capacitor voltage value. When the voltage remains in the snubber capacitor, the IGBT is turned on by hard switching.

Thus, in the control method of the present embodiment, it is possible to achieve zero voltage switching of the IGBT at light load, and particularly to improve the efficiency at light load.

FIG. 7A is a control configuration diagram of a DC-DC converter according to a fifth embodiment of the present invention. 3a
3d are IGBTs that receive the DC power supply 1, 4a to 4d are reverse conducting diodes, 5a to 5d are snubber capacitors, and these are configured as primary full bridges, and are configured as rectifier circuits via the high frequency transformer 6. Rectifier diode 8a
To 8d, connected to the secondary phase control IGBTs 9a and 9b, the reactor 10, the capacitor 11, and the current detector 14, and a DC output is given to the load resistor 12.

In the present embodiment, the load current is detected by the current detector 14, and especially when the load current becomes a predetermined current value or less at a light load, the secondary phase control IGBT 9a, The two elements 9b are turned on without switching. Instead of controlling DC voltage on the secondary side,
At the same time, DC voltage control is performed by changing the ON width of a pair of IGBTs between the arms to be switched. In the case of DC voltage control on the secondary side, the IGBT in the full-full bridge on the primary side
When the switching interval of 3a to 3d changes greatly, it becomes very difficult to control the secondary phase control IGBTs 9a and 9b in the secondary rectifier circuit. Therefore, in this case, the control is facilitated by turning on the secondary phase control IGBTs 9a and 9b and performing DC voltage control by the IGBTs 3a to 3d in the primary full bridge as shown in FIG. 7B. .

Thus, in the control method of the present embodiment, it is possible to achieve zero voltage switching of the IGBT at light load, and particularly to improve the efficiency at light load.

FIG. 8 shows a sixth embodiment of the present invention.
It is a control block diagram of a C-DC converter. Reference numerals 3a to 3d denote IGBTs which receive the DC power supply 1, 4a to 4d denote reverse conducting diodes, 5a to 5d denote snubber capacitors, 6 denotes a high-frequency transformer, 8a to 8d denote rectifying diodes, and 9a to 9b denote 2 diodes.
The next phase control IGBT, 10 is a reactor, 11 is a capacitor, 12 is a load resistor, 14 is a current detector, 15 is a switching assisting reactor, and 16 is an on / off switch.

In the present embodiment, the load current is detected by the current detector 14, and particularly when the load current becomes equal to or less than a predetermined current value under a light load, the switching assist reactor 15 is switched by the switching operation of the open / close switch 16. An inductor is inserted between the primary terminals of the transformer 6 in parallel, and an inductance is inserted between the primary terminals of the transformer 6 in parallel. A current corresponding to the ON width of the IGBT flows through the inductance regardless of the load current. By the current of this inductance, the charging and discharging of the snubber capacitor can be surely completed even when the load current is small.

Thus, in the control method of the present embodiment, it is possible to achieve zero voltage switching of the IGBT at light load, and it is possible to improve the efficiency especially at light load.

[0051]

As described above, according to the present invention, in a resonance type DC-DC converter, charging of a capacitor for zero voltage switching connected in parallel to a main circuit IGBT at a low load current at a light load. Since the discharge time can be ensured, it is possible to provide a control method capable of achieving zero voltage switching of the semiconductor switching element and improving the efficiency particularly under light load.

[Brief description of the drawings]

FIG. 1 shows a resonance type DC-DC according to a first embodiment of the present invention.
FIG. 2 is a circuit configuration diagram showing a converter.

FIG. 2 is an explanatory diagram showing a circuit operation according to the first embodiment of the present invention.

FIG. 3 shows a resonance type DC-DC according to a second embodiment of the present invention.
FIG. 2 is a circuit configuration diagram showing a converter.

FIG. 4 shows a resonance type DC-DC according to a third embodiment of the present invention.
FIG. 2 is a circuit configuration diagram showing a converter.

FIG. 5 shows a resonance type DC-DC according to a fourth embodiment of the present invention.
FIG. 2 is a circuit configuration diagram showing a converter.

FIG. 6 is an explanatory diagram showing a circuit operation according to a fourth embodiment of the present invention.

FIG. 7 shows a resonance type DC-DC according to a fifth embodiment of the present invention.
FIG. 2 is a circuit configuration diagram showing a converter.

FIG. 8 shows a resonance type DC-DC according to a sixth embodiment of the present invention.
FIG. 2 is a circuit configuration diagram showing a converter.

FIG. 9 is a circuit diagram showing a first example of a conventional resonance type DC-DC converter.

FIG. 10 is an explanatory diagram showing a circuit operation of the resonance type DC-DC converter.

FIG. 11 is a circuit diagram showing a second example of a conventional resonance type DC-DC converter.

[Explanation of symbols]

1: DC power supply, 2a to 2d: DC voltage dividing capacitor,
3a to 3d: IGBT, 4a to 4f: reverse conducting diode, 5a to 5d: snubber capacitor, 6: high frequency transformer, 7a to 7b: resonance reactor, 8a to 8d: rectifier diode, 9a to 9b: secondary phase control IGBT, 1
0: reactor, 11: capacitor, 12: resistor, 13
a to 13b: IGBT for resonance, 14: Current detector, 15
... reactor for switching assistance, 16 ... open / close switch.

Continued on the front page F term (reference) 5H730 AA02 AA14 AS23 BB27 BB57 BB76 DD02 DD41 EE05 EE08 EE13 EE18 FD31 FG05 FG07 FG22

Claims (10)

[Claims] 1. A resonance type full-bridge inverter for connecting a soft switching capacitor in parallel to a semiconductor switching element having a DC power supply as an input and having a resonance circuit for charging / discharging the capacitor and converting the DC power to an AC output. A resonant DC having a transformer receiving the AC output of the full bridge inverter on the primary side, and a rectifying circuit for rectifying the secondary side output of the transformer to DC and outputting a predetermined DC voltage to a load. -A control method for a DC-DC converter, comprising: changing a switching interval of the semiconductor switching element so that charging and discharging of the capacitor are completed when the semiconductor switching element is switched. . 2. The control method for a resonance type DC-DC converter according to claim 1, wherein an upper limit is set for the switching interval. 3. The control method for a resonance type DC-DC converter according to claim 1, wherein an ON width of said semiconductor switching element is variable. 4. A semiconductor device according to claim 1, wherein when the load current becomes equal to or less than a predetermined current value, the two sets of semiconductor switching elements forming a pair between the two arms of the full-bridge inverter are simultaneously switched. 4. The control method for a resonance type DC-DC converter according to any one of claims 1 to 3. 5. The method according to claim 1, wherein an inductance is inserted in parallel with the AC output of the full-bridge inverter when the load current becomes equal to or less than a predetermined current value. A method for controlling a resonance type DC-DC converter. 6. A resonance type full-bridge inverter for connecting a soft switching capacitor in parallel to a semiconductor switching element having a DC power supply as an input and having a resonance circuit for charging and discharging the capacitor and converting the DC power to an AC output. A transformer that receives the AC output of the full-bridge inverter on the primary side, a rectifier circuit that rectifies the secondary-side output of the transformer into DC and outputs a predetermined DC voltage to a load, Resonant DC comprising a semiconductor switching element inserted in each direction path for bidirectional
-In the control method of the DC converter, the full bridge on the primary side of the transformer is so set that the charging and discharging of the capacitor are completed when the semiconductor switching element of the full bridge inverter on the primary side of the transformer is switched. Resonant type DC- characterized by changing a switching interval of a semiconductor switching element of an inverter.
Control method of DC converter. 7. The control method for a resonance type DC-DC converter according to claim 6, wherein an upper limit is set for the switching interval. 8. When the switching interval is longer than a predetermined time, all the semiconductor switching elements of the rectifier circuit are turned on, and the semiconductor switching elements are kept on until the switching interval becomes shorter than a predetermined time. The resonance type DC-DC according to claim 6 or 7, wherein
How to control the converter. 9. The control method for a resonance type DC-DC converter according to claim 8, wherein an ON width of said semiconductor switching element of said full bridge inverter is variable. 10. The method according to claim 6, wherein an inductance is inserted in parallel with the AC output of the full-bridge inverter when the load current falls below a predetermined current value.
10. The control method for a resonance type DC-DC converter according to any one of claims 9 to 9.