JP2002262567A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce a design margin and ensure the breakdown voltage of switching a element at short-circuiting of a load. SOLUTION: In a composite-resonance converter, a self-exciting type is used for a primary-side voltage-resonance converter, and a control voltage variably controlled according to the level of a secondary-side direct-current output voltage is applied to a gate electrode of a MOSFET for a constant-voltage control. According to the state of conduction of the MOSFET, the capacitance to a self-oscillation circuit is varied, and a switching frequency is variably controlled. Thus, the switching element is switch-controlled by a combined control method wherein the conduction angle and the switching frequency of the switching elements are simultaneously varied. In a load short-circuited state, or when power is applied, the switching frequency is controlled so as to be maximized within a control range or to exceed the maximum, and an excessive voltage is thereby prevented from being applied to the switching element.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 11 is a circuit diagram showing an example of a prior art switching power supply circuit which can be constructed based on the invention proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in FIG. 1, a rectified and smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC is generated from a commercial AC power supply (AC input voltage VAC) by a bridge rectifier circuit Di and a smoothing capacitor Ci. .

[0005] A single-ended single-end system is adopted as a voltage resonance type converter that receives and inputs the rectified smoothed voltage Ei (DC input voltage) and is intermittent. The drive system employs a self-excited configuration. In this case, the switching element Q1 forming the voltage resonance type converter includes:
A high breakdown voltage bipolar transistor (BJT; junction transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A series circuit of a clamp diode DD and a resistor RD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr
Forms a primary-side parallel resonance circuit together with the leakage inductance L1 obtained in the primary winding N1 of the insulating converter transformer PIT, whereby an operation as a voltage resonance type converter can be obtained. The drive winding NB, the resonance capacitor CB, and the base current limiting resistor RB are connected to the base of the switching element Q1.
Is connected. The switching element Q1 is switched by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation driving circuit. In addition, at the time of startup, it is started by a startup current flowing from the line of the rectified smoothed voltage Ei to the base via the startup resistor Rs.

The orthogonal control transformer PRT is configured by winding a control winding Nc so that the winding direction of the drive winding NB and the current detection winding ND is orthogonal to the winding direction. It is provided for controlling the switching frequency of the primary side voltage resonance type converter as described later.

Although the structure of the orthogonal control transformer PRT is not shown, the three-dimensional core is formed by joining the ends of two double U-shaped cores having four magnetic legs to each other. To form Then, a resonance current detecting winding ND and a driving winding NB are wound around predetermined two magnetic legs of the three-dimensional core in the same winding direction.
Are wound in a direction orthogonal to the resonance current detection winding ND and the drive winding NB.

[0008] The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. This insulated converter transformer PIT has a primary winding N with respect to an EE type core.
1 and the secondary winding N2 are divided and wound, and a gap is formed with respect to the center magnetic leg so that a loose coupling state with a required coupling coefficient can be obtained and a saturated state can be obtained. I try to make it difficult.

The primary winding N1 of the insulating converter transformer PIT is connected between the line of the DC input voltage (rectified smoothed voltage Ei) and the collector of the switching element Q1. The switching element Q1 performs switching with respect to the DC input voltage. As a result, the switching output of the switching element Q1 is supplied to the primary winding N1, and an alternating voltage having a period corresponding to the switching frequency is generated. I do.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

An isolated converter transformer PIT in this case
First, the anode of the rectifier diode DO1 is connected to the winding end of the secondary winding N2, and the cathode is connected to the positive terminal of the smoothing capacitor CO1, thereby forming a half-wave rectifier circuit. Has formed. With this half-wave rectifier circuit, a secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1. In this case, a tap is provided for the secondary winding N2, and a half-wave rectifier circuit composed of a rectifier diode DO2 and a smoothing capacitor CO2 is formed as shown in FIG. Then, depending on the half-wave rectifier circuit, the secondary-side DC output voltage E O which is lower than the secondary-side DC output voltage EO1 is provided.
O2 is obtained. Note that, specifically, the secondary side DC output voltage EO1 = 135V and the secondary side DC output voltage EO2 = 15V.

These secondary-side DC output voltages EO1, EO2 are:
Each is supplied to a required load circuit. The secondary DC output voltage EO1 is branched and output as a detection voltage of the control circuit 1.

In the control circuit 1, resistors R3 and R4 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q3 is connected to this connection point (voltage division point). . The anode of the shunt regulator Q3 is grounded, and the cathode is connected to the line of the secondary side DC output voltage EO2 via the control winding NC of the orthogonal control transformer PRT. Here, the cathode of the shunt regulator Q3 is the capacitor C1
It is connected to the connection point of the resistors R3 and R4 via 1. A series connection circuit of a capacitor C3 and a resistor R5 is connected in parallel to the resistor R4.

The control circuit 1 formed by the above connection forms a function as an error amplifier having the DC output voltage EO1 as a detection input. That is, the DC output voltage EO1 is changed to the resistance R
3. The voltage divided by R4 is input to the control terminal of the shunt regulator Q3 as a control voltage. Therefore, in the shunt regulator Q3, a current having a level corresponding to the DC output voltage EO1 is caused to flow as the control current Ic to the control winding NC. That is,
The control current level flowing through the control winding NC is variably controlled. By varying the level of the control current flowing through the control winding Nc, the orthogonal control transformer PRT is controlled so as to vary the inductance LB of the drive winding NB. As a result, the resonance frequency of the resonance circuit including the drive winding NB and the resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the switching element Q1 is variably controlled. By changing the switching frequency of the switching element Q1 in this way, the secondary DC output voltage is controlled to be constant. That is, the power supply is stabilized. Here, in changing the switching frequency, the period during which the main switching element Q1 is turned off is kept constant, and then the period during which the main switching element Q1 is turned on is variably controlled. That is, the conduction angle control for the ON period and the switching frequency control are performed. In the present specification, such complex control is referred to as a “complex control method”.

FIG. 12 shows an operation waveform of each part under heavy load as an operation of a main part of the power supply circuit having the configuration shown in FIG. Here, the operation on the primary side is mainly shown. A series resonance circuit (N
In (B, CB), the resonance operation is performed by the alternating voltage obtained in the drive winding NB, thereby obtaining a sinusoidal series resonance current I2 as shown in FIG. When the series resonance current I2 passes through the base current limiting resistor RB, a base current (drive current) IB flows through the base of the switching element Q1, as shown in FIG. With the driving current IB, the switching element Q1 performs a switching operation.

At this time, the collector current IQ1 flowing through the collector of the switching element Q1 has a waveform shown in FIG. Further, both ends of the parallel connection circuit of the switching element Q1 // parallel resonance capacitor Cr are connected to both ends of FIG.
As shown in (a), a parallel resonance voltage V1 is generated by the operation of the parallel resonance circuit. This parallel resonance voltage V1
As shown in the figure, a period TON in which the switching element Q1 is on is 0 level, and a waveform of a sine wave pulse is obtained in the period TOFF in which the switching element Q1 is off, which corresponds to the operation as a voltage resonance type.

The switching operation of the switching element Q1 at the above timing causes the winding current I1 flowing through the primary winding N1 to have an alternating waveform corresponding to the switching cycle as shown in FIG. Become.

Here, during the period TON during which the switching element Q1 is turned on, the series resonance current I2 shown in FIG.
The region of positive polarity corresponds to the region of the forward bias current of the drive current IB in FIG. Also, the same period TON
In this case, a region where the series resonance current I2 has a negative polarity is a reverse bias current of the drive current IB. The area of the reverse bias current of the drive current IB during this period TON is the accumulation time of the switching element Q1.

A low-speed damper diode D having a long reverse recovery time is provided between the base and the emitter of the switching element Q1.
A series circuit of D and a resistor RD is connected. In the period TOFF during which the switching element Q1 is turned off, the negative series resonance current I2 flows through the resistor RD → the clamp diode DD → the base current limiting resistor RB → the resonance capacitor CB → the drive winding NB. (G) Damper current I
D1 is a waveform obtained in the period TOFF. Then, when the period TON starts, the parallel resonance capacitor Cr
Charge / discharge energy flows through the clamp diode DD → the base → the collector of the switching element Q1, and this becomes a negative damper current (ID) at the start of the period TON (turn-on). When this period ends, the damper diode DD suddenly rises in the direction of the positive polarity in the region of the reverse recovery time, and thereafter gradually becomes 0 level at the end of the period TON as shown in the figure. A pulsating waveform is obtained.

In response to the drive current IB and the damper current ID1 flowing as described above, the base-emitter voltage VBE of the switching element Q1 becomes negative during the period TOFF as shown in FIG. In the period TON, the damper period at the start of the period has a sharply negative negative peak, and when this period ends, the waveform has a fixed positive level and an offset with respect to the zero level. is there. This offset level is determined, for example, by the resistance value of the resistor RD.

FIG. 13 shows the control characteristics of the power supply circuit of FIG. 11 which operates as described above. As the load current Io of the secondary side DC output voltage EO1 changes in the range of 0 to 1.5 mA, the control current Ic changes as shown in the figure. In other words, the load current increases, resulting in a heavy load condition,
Control is performed such that the control current level decreases as the secondary side DC output voltage EO1 decreases. As a result, the switching frequency fs is controlled so as to decrease as the load becomes heavy. In addition, the case where the AC input voltage VAC is 120 V and VAC = 90 V is shown as corresponding to the fluctuation of the AC input voltage VAC.
AC input voltage VAC = 90 when AC = 120V
V, the switching frequency f
As for s, the condition when the AC input voltage VAC = 120 V is higher than the condition when the AC input voltage VAC = 90 V. This is because the control current Ic is controlled to increase when the level of the AC input voltage VAC increases and the secondary side DC output voltage EO1 increases, and the switching frequency fs also increases accordingly. This indicates that the vehicle is controlled to be raised.

[0023]

The above-mentioned FIG.
As can be seen from the control characteristics shown in FIG. 13, the power supply circuit shown in FIG. 13 operates to control the switching frequency lower as the load becomes heavy in stabilizing the power supply. Therefore, the control function for the switching frequency fs does not operate when the load is short-circuited, and the switching frequency fs is out of the steady-state control range.
Lower than 80 KHz.

In such a state, as shown in FIG. 14 (a), the period TON greatly increases as the parallel resonance voltage V1 generated at both ends of the primary side parallel resonance capacitor Cr. The peak level of the resonance pulse generated during the period TOFF increases. At this time, the collector current flowing into the collector of the main switching element Q1 becomes a saw-tooth wave as shown in FIG.

For this reason, for example, the AC input voltage VAC = 10
In the case of the 0 V system, 90
It is necessary to select a withstand voltage product of 0 V or more so that the switching element Q1 is not destroyed even if the load short-circuit state as described above is reached. However, as the switching element (BJT) has a higher breakdown voltage, the fall time increases. For this reason, the switching characteristics are reduced and the power loss is increased, and as a result, the AC / DC power conversion efficiency is reduced. Therefore, when the switching element Q1 is selected to have a withstand voltage lower than 900 V, for example, 800 V, and the reduction of the AC / DC power conversion efficiency is to be eliminated, for example, the secondary winding of the insulated converter transformer PIT is used. After connecting an overcurrent detection resistor in series with the line N2, or connecting an overcurrent detection resistor in series with the emitter of the switching element Q1,
It is necessary to provide an overload protection circuit. However, this configuration involves a power loss of the overcurrent detection resistor and also increases the number of components constituting the circuit.

Further, in the circuit shown in FIG. 11, during the transition time from when the AC switch is turned on and the commercial AC power is turned on to when the rated load power is reached,
The switching frequency fs increases from a low frequency outside the control range to an appropriate control frequency. Therefore, also at this time, the peak values of the primary side parallel resonance voltage V1 and the collector current IQ1 may rise and be destroyed.
For this reason, for example, it is necessary to provide a soft start circuit. However, also in this case, since a circuit portion such as a soft start circuit is added, the number of components in the entire power supply circuit is greatly increased.

In a power supply circuit having a switching converter for performing switching frequency control in a self-excited manner as shown in FIG.
Will be provided. However, in this orthogonal control transformer PRT, the core gap is made as small as about 10 μm in order to reduce the amount of control current flowing through the control winding. For this reason, an error in the accuracy of the gap is inevitable at the time of manufacturing, but this causes variation in the inductance value of the drive winding NB wound on the orthogonal control transformer PRT. If the inductance value of the drive winding NB varies, an error occurs in the resonance frequency of the self-excited oscillation drive circuit formed with the drive winding NB. Therefore, if the commercial AC power supply is a 100 V system, the AC input voltage VAC = 8
It is necessary to configure the circuit with a large margin so that the stabilization control can be performed from 0 V or more, which is not easy as a design.

[0028]

In view of the above-mentioned problems, the present invention has the following configuration as a switching power supply circuit. That is, a switching means formed of a single switching element for switching the DC input voltage, an insulating converter transformer for transmitting the output of the switching means obtained on the primary winding to the secondary winding, and an insulating converter transformer A primary-side parallel resonance circuit formed by a primary winding and a primary-side parallel resonance capacitor and provided to make the operation of the switching means a voltage resonance type, and a secondary winding wound around the insulating converter transformer. And a secondary parallel resonance circuit formed by connecting the secondary parallel resonance capacitor in parallel, and a DC output voltage by inputting an alternating voltage obtained in the secondary parallel resonance circuit and performing a rectification operation. And a DC output voltage generating means configured to obtain A switching drive unit configured as a series resonance circuit including a live winding, an inductor, a capacitor, and a resistor to apply a switching drive signal to the switching element to perform a switching operation; and connected in parallel to the capacitor of the series resonance circuit. A series circuit of a capacitor and a MOS-FET, and a control voltage that is varied according to the level of the DC output voltage obtained by the DC output voltage generating means.
Constant voltage control means for controlling the switching frequency supplied to the switching element by applying the voltage to the gate of the OS-FET to perform constant voltage control on the DC output voltage.

According to the above configuration, the switching element provided on the primary side of the composite resonance type converter is driven in a self-excited manner. Further, for constant voltage control, the gate voltage of the MOS-FET is controlled according to the DC output voltage on the secondary side, and the switching frequency of the switching element is controlled according to the gate voltage. With such a configuration for a constant voltage, for example, it is possible to obtain a control operation in which the switching frequency is controlled to increase as the load becomes heavy. With such a configuration of constant voltage control, for example, in the case of the self-excited type, it is possible to omit the orthogonal control transformer used for the switching frequency variable control.

[0030]

FIG. 1 shows a configuration of a power supply circuit according to a first embodiment of the present invention. The power supply circuit shown in FIG. 1 employs a configuration as a composite resonance type switching converter including a voltage resonance type converter on a primary side and a parallel resonance circuit on a secondary side. In the power supply circuit shown in this figure, first, as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. Is provided, and the AC input voltage V
A rectified and smoothed voltage Ei corresponding to a level of one time of AC is generated.

As the switching converter that receives the rectified smoothed voltage Ei (DC input voltage) and is intermittent,
A voltage resonance type converter including a stone switching element Q1 and performing a so-called single-ended switching operation is provided. The voltage resonance type converter here has a self-excited type configuration, and a high breakdown voltage bipolar transistor (BJT: junction type transistor) is used as the switching element Q1. Switching element Q
The collector of 1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the emitter is connected to the primary side ground.

The collector of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the emitters. The capacitance of the parallel resonance capacitor Cr and the primary winding N of the insulating converter transformer PIT
The primary side parallel resonance circuit is formed by the leakage inductance obtained in (1). And
The resonance operation by the parallel resonance circuit is obtained according to the switching operation of the switching element Q1, so that the switching operation of the switching element Q1 is a voltage resonance type.

A clamp diode DD1 and a resistor RD are inserted in series between the base and the emitter of the switching element Q1. In this case, the cathode of the clamp diode DD1 is connected to the base of the switching element Q1, and the anode of the clamp diode DD1 is connected to the resistor R1.
Connected to the primary side ground via D. Further, a damper diode D2 is connected in parallel between the collector and the emitter of the switching element Q1. Thereby, the damper current flowing through the base-collector electrode of the switching element is prevented from flowing.

The base of the switching element Q1 is connected to the line of the rectified and smoothed voltage Ei via the starting resistor Rs.
It is configured to start up when a base current obtained through s flows.

As shown, LC of [resonant capacitor CB1-drive winding NB-base current limiting resistor RB-inductor LB] is connected to the base of switching element Q1.
An R series connection circuit is connected. This series connection circuit is a self-excited oscillation drive circuit for driving the switching element Q1 in a self-excited manner. In this case, the drive winding NB in the self-excited oscillation drive circuit is wound around the primary side of the insulating converter transformer PIT, and is excited by the switching output voltage obtained in the primary winding N1. And, as the self-excited oscillation drive circuit, the resonance capacitor CB1-drive winding NB
A series resonance circuit is formed by the inductor LB; The resonance frequency of this series resonance circuit is
And the inductance of the drive winding NB and the resonance capacitor CB
And a capacitance of 1. However, in this example, a series circuit of a capacitance varying capacitor CB2 and a MOS-FET (Q2) is connected in parallel with the resonance capacitor CB1. This function will be described later.

In the self-excited oscillation drive circuit, an alternating voltage as a drive voltage is generated in the drive winding NB excited by the primary winding N1 of the insulating converter transformer PIT. This drive voltage is applied via a base current limiting resistor RB and a series resonance circuit (CB1-NB-LB).
The drive current is output to the base of the switching element Q1. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit. Then, the switching output obtained at the collector is transmitted to the primary winding N1 of the insulating converter transformer PIT.

In the case of the circuit shown in this figure, a tertiary winding N3 is wound around the primary side of the insulating converter transformer PIT. A half-wave rectifier circuit including a diode D1 and a capacitor C1 is connected to the tertiary winding N3. A low-level DC voltage of a predetermined level is obtained at both ends of the capacitor C1. And
The positive terminal of the capacitor C1 is connected to the phototransistor of the photocoupler PC. In the phototransistor of the photocoupler PC, a current corresponding to a voltage obtained at the positive terminal of the capacitor C1 and a current corresponding to a current of a photodiode disposed on the secondary side of the insulating converter transformer PIT described later flow. The voltage divided by the resistors R1 and R2 is applied to the gate of the MOS-FET (Q2). In addition,

The drain of the MOS-FET (Q2) is connected to the capacitance varying capacitor CB2, and the source is connected to the primary side ground. The clamp diode DD2 is connected in parallel to the drain-source of the MOS-FET (Q2) in the direction shown. In this case, the clamp diode DD2 includes a MOS-FET
A so-called body diode built in (Q2) can be used. Since the series circuit of the capacitance varying capacitor CB2 and the MOS-FET (Q2) is connected in parallel with the resonance capacitor CB1 as described above, the conduction state of the MOS-FET (Q2) causes the series resonance circuit (CB1). -NB-LB). That is, the switching frequency applied to the base of the switching element Q1 is variably controlled.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. The insulated converter transformer PIT includes an EE-type core in which two sets of E-type cores made of, for example, a ferrite material are combined so that their magnetic legs face each other. Utilizing primary winding N
1 and the secondary winding N2 are wound separately. Then, a gap is formed with respect to the center magnetic leg. As a result, loose coupling with a required coupling coefficient can be obtained. The gap is two sets of E
Each center magnetic leg of the mold core can be formed by making it shorter than two outer magnetic legs. In addition, a loosely coupled state, for example, k ≒ 0.85 is obtained as the coupling coefficient k, so that a saturated state is hardly obtained.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, this power supply circuit has a "composite resonance circuit" in which a primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and a secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. It is referred to as a “resonant switching converter”.

With respect to the secondary side of the power supply circuit formed as described above, a half of the secondary side rectifying diode DO1 and the smoothing capacitor CO1 connected to the winding start end of the secondary winding N2. A wave rectifier circuit is provided to obtain a secondary side DC output voltage EO1 corresponding to an approximately equal level of the alternating voltage induced in the secondary winding N2. Also,
Here, a tap output is provided for the secondary winding N2,
As shown, a secondary rectifier diode D02 and a smoothing capacitor CO2 are provided between the tap output and the secondary ground.
By connecting a half-wave rectifier circuit composed of the following, a low-voltage secondary-side DC output voltage EO2 is obtained. In this case, the secondary side DC output voltage EO1 is input to the control circuit 1 as a detection voltage for constant voltage control. The secondary side DC output voltage EO2 is also used as an operation power supply of the control circuit 1.

In the control circuit 1, resistors R11 and R12 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q3 is connected to this connection point (voltage division point). . The anode of the photodiode of the photocoupler PC is connected to the DC output voltage EO2 via a series circuit of resistors R13 and R14.
The cathode of the photodiode is connected to the secondary side ground. The cathode of the shunt regulator Q3 is connected to the connection point of the resistors R13 and R14. The anode of the shunt regulator Q3 is grounded.
Here, the cathode of the shunt regulator Q3 is connected to the connection point of the resistors R11 and R12 via the capacitor C12. A series connection circuit of a capacitor C11 and a resistor R13 is connected in parallel to the resistor R11.

The control circuit 1 formed by the above-described connection form functions as an error amplifier having the DC output voltage EO1 as a detection input, and its operation is as follows. In the photodiode of the photocoupler PC, a current flows from the line of the secondary side DC output voltage EO2 through the series connection of the resistors R13 and R14, and the shunt regulator Q3 receives the current through the resistor R13. The current obtained from the line of the DC output voltage EO2 is divided. Further, a voltage obtained by dividing the DC output voltage EO1 by the resistors R11 and R12 is input to the control terminal of the shunt regulator Q3 as a control voltage. As a result, in the shunt regulator Q3, the secondary side DC output voltage EO
The current level flowing from 2 through the resistor R13 can be varied according to the DC output voltage EO1.

Here, it is assumed that the load power of the secondary side DC output voltage EO1 has changed so that the level of the secondary side DC output voltage EO1 becomes low under the condition of light load. In this case,
Since the current level shunted from the line of the secondary side DC output voltage EO2 to the shunt regulator Q3 also changes, the resistance R
The current level flowing through the photodiode via the series connection of 13-R14 will increase accordingly. As a result, the conducting current level in the phototransistor of the photocoupler PC on the primary side increases,
Accordingly, the MOS-FE is controlled by the resistors R1 and R2.
The control voltage Vc obtained at the gate of T (Q2) is increased by being applied only at a level corresponding to the current conduction level in the phototransistor.

In this way, the conduction state of the MOS-FET (Q2) is controlled by the control voltage Vc whose level changes, whereby the capacitance-variable capacitor CB2 connected in series with the MOS-FET (Q2) is controlled. The capacitance is controlled, that is, the series resonance circuit (CB1-
The function of varying the capacitance of (NB−LB) is obtained. Therefore, the switching frequency applied to the base of the switching element Q1 is variably controlled.

The self-excited oscillation circuit formed with the series resonance circuit (CB1-NB-LB) uses the voltage VNB induced in the drive winding NB as a voltage source during the ON period TON of the switching element Q1. The base current IB flows through the switching element Q1 due to the current Io, and during the OFF period TOFF of the switching element Q1, the resonance current Io has the opposite polarity, and the base element IB has the negative polarity, so that the switching element Q1 is turned off. The switching frequency fs is

(Equation 1) And the capacitance CB12 in (Equation 1)
Is

(Equation 2) Therefore, if the capacitances of the capacitors CB1 and CB2 are selected so that CB1 << CB2, the MOS-FET
Switching frequency fs by gate voltage of (Q2)
Is variable.

When the switching frequency is varied, the operation is performed by the above-described complex control method. The period during which the switching element Q1 is off is constant, and the conduction angle control for controlling the period during which the switching element Q1 is on is simultaneously performed. . And
By such an operation, when the level of the secondary-side DC output voltage EO2 is lowered, an operation of increasing the level is obtained, and the voltage is made constant.

FIG. 2 is a waveform chart showing the operation of the main part in the power supply circuit having the configuration shown in FIG. This figure shows the operation under the condition that the AC input voltage VAC is 100 V and the load is heavy. Here, a drive voltage VNB is generated in the drive winding NB forming the self-excited oscillation drive circuit of the switching element Q1, as shown in FIG. 2 (h). A resonance current Io flows through the series resonance circuit (NB, CB1, LB) as a circuit as shown in FIG. This resonance current I
o indicates that a positive level is obtained in the period TON and the period TOFF
Is a negative polarity level. Then, when the resonance current Io becomes a positive level in the period TON, FIG.
As shown in (d), a base current IB according to the illustrated waveform flows to the base of the switching element Q1 during the period TON. As a result, the switching element Q1 is turned on. On the other hand, in the period TOFF, the base current IB becomes 0 level because the resonance current Io has the negative polarity level, and the switching element Q1 is turned off. Thus, the switching element Q1 is driven by switching. In addition, the negative-polarity resonance current Io in the period TOFF flows as the current ID through the clamp diode DD1 as shown in FIG.

Then, with the operation of the drive circuit system described above, the base-emitter voltage VB of the switching element Q1
As shown in FIG. 2 (f), E has a negative polarity in the period TOFF, and maintains a constant level given a predetermined offset with respect to the 0 level in the period TON.

When the switching element Q1 performs the switching operation, the resonance voltage V1 obtained at both ends of the primary side parallel resonance capacitor Cr is changed during the period TON during which the switching element Q1 is turned on, as shown in FIG. Then 0
At the level, a waveform that is a sine wave pulse is obtained in the OFF period TOFF, which indicates that the primary-side switching converter operates in a voltage resonance type.

The collector current IQ1 flowing to the collector of the switching element Q1 is at the 0 level during the period TOFF as shown in FIG. 2B. Assuming that the damper diode D2 does not exist in the period TON, the clamp diode DD → Q1 at the start of the period TON.
Although the damper current flows in the positive direction of the negative electrode via the base → Q1 collector, the damper diode D2 is arranged in parallel with the switching element Q1. It will be slight. After that, a waveform flows in the positive direction from the collector to the emitter. The primary winding current I1 flowing through the primary winding N1 is inverted from the positive polarity to the negative polarity in the period TOFF at a timing according to the switching cycle of the switching element Q1, as shown in FIG. In the period TON, a waveform that reverses from negative polarity to positive polarity is obtained.

FIG. 4 shows constant voltage control characteristics of the power supply circuit shown in FIG. In this figure, the control voltage V with respect to the load current Io of the secondary side DC output voltage EO1 is shown.
The relationship between c and the switching frequency fs is shown. Here, for the AC input voltage VAC, VAC = 120
V and 90 V are shown. In the present embodiment, as the load current Io increases in the range of 0 mA to 1.5 mA, that is, as the secondary DC output voltage decreases under heavy load conditions, the control voltage Vc becomes Is variably controlled so as to rise as shown in FIG. Then, control is performed such that the switching frequency fs decreases as the control voltage Vc increases.

In the present embodiment, by obtaining such control characteristics, for example, the secondary side DC output voltages EO1, EO2
When the load is short-circuited, the control voltage Vc is controlled so as to be equal to 0, whereby the switching frequency fs rapidly rises to 210 KHz or more in the circuit configuration of FIG. The operation at this time includes the primary-side parallel resonance voltage V1 in FIG.
As shown as the collector current IQ1 in (b),
It can be seen that the period TON is shortened and the switching frequency is controlled to be higher. Accordingly, the peak level of the sine wave pulse in the period TOFF is suppressed, and the primary side parallel resonance voltage V1 can be set to a lower level than in the normal operation. Also, the peak level of the collector current IQ1 during the period TON is suppressed,
This level is also lower than in the normal operation. Further, with the control characteristics shown in FIG. 4, the switching frequency is controlled to be high even in the transition period until the secondary DC output voltage reaches the rated load power when the power is turned on. . Therefore, also at this time, the operation described with reference to FIG. 3 can be obtained.

This means that no excessive voltage is applied to the switching element Q1 even when the load is short-circuited or the power is turned on. This eliminates the need for providing a circuit for overload protection and a soft-start circuit as the power supply circuit of the present embodiment, and the number of components constituting the circuit is reduced accordingly. It is possible to promote weight reduction and cost reduction.

Further, as for the switching element Q1,
It is possible to select a device with a lower withstand voltage. For example, in the case of the power supply circuit of the present embodiment, it is possible to select a product with a withstand voltage of 800 V or less. Then, since a low breakdown voltage product can be selected for the switching element Q1, switching characteristics are not reduced. Therefore, the power conversion efficiency is improved, and in the case of the circuit of the present embodiment, it is possible to maintain the AC / DC power conversion efficiency of 92% or more.

The power supply circuit according to this embodiment includes:
The primary-side voltage resonance type converter is self-excited and the switching frequency is controlled by the composite control method. However, the circuit configuration described with reference to FIG. In this case, the orthogonal control transformer PRT is omitted. Thereby, in the present embodiment, the drive winding NB caused by the gap variation at the time of manufacturing the orthogonal control transformer PRT
Will be solved. Therefore, it is possible to set a small margin for the range of the AC input voltage VAC.
The circuit can be easily designed.

Further, the provision of the damper diode D2 in parallel with the switching element Q1 prevents a negative damper current from flowing through the clamp diode DD1, the base of the Q1, and the collector of the Q1. Is stabilized. MOS-FET
As for (Q2), a low-voltage small-capacity product having a withstand voltage of 30 V and a rated current of 1 A or less may be used.

FIG. 5 shows a configuration example of a power supply circuit according to the second embodiment. In this figure, the same parts as those in FIG. In the power supply circuit shown in this figure, [rectifier diodes Di1, Di2, smoothing capacitors Ci1, Ci2] are shown as rectifying and smoothing circuits for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. A voltage doubler rectifier circuit is formed by connecting the connection types. In this voltage doubler rectifier circuit, a smoothing capacitor Ci1 connected in series is connected.
A rectified smoothed voltage Ei corresponding to twice the AC input voltage VAC is generated at both ends of -Ci2 and supplied to the primary-side voltage resonance type converter.

In this case, the self-excited oscillation driving circuit for the switching element Q1 is connected in series from the base side of the switching element Q1, in the order of the base current limiting resistor RB, the capacitor CB1, the inductor L1, and the driving winding NB. Connected.

Also in this example, the resonance capacitor C
Capacitance variable capacitors CB2 and MO in parallel with B1
A series circuit of S-FET (Q2) is connected. Then, a half-wave rectifier circuit including a diode D1 and a capacitor C1 is connected to the tertiary winding N3 on the primary side of the insulating converter transformer PIT, and the positive side of the capacitor C1 is connected to the phototransistor of the photocoupler PC; In the phototransistor of the photocoupler PC, a current corresponding to a voltage obtained at the positive terminal of the capacitor C1 and a current of a photodiode disposed on the secondary side of the insulating converter transformer PIT described later flows. As a voltage divided by R1 and R2,
The voltage applied to the gate of the MOS-FET (Q2) is the same as in the power supply circuit of FIG.

On the secondary side of the insulating converter transformer PIT, a center tap is provided for the secondary winding N2, and rectifier diodes DO1 and DO2 and a smoothing capacitor CO1 are connected as shown in the figure. ,
A full-wave rectifier circuit is formed so that a secondary-side DC output voltage EO1 corresponding to the same voltage as the voltage obtained in the secondary winding N2 is generated at both ends of the smoothing capacitor CO1. .

The DC output voltage EO1 is also supplied to the control circuit 1. The control circuit 1 has, for example, the configuration shown in FIG.
The current according to the level of the DC output voltage EO1 is caused to flow through the photodiode of the photocoupler 2.

Therefore, in the case of the circuit example of FIG.
The control voltage Vc applied to the gate of the FET (Q2) is variably controlled in accordance with the level of the DC output voltage EO1;
The conduction state of the S-FET (Q2) is controlled. Thus, an action of varying the capacitance of the series resonance circuit (CB1-NB-L1) is obtained. That is, the switching frequency and the conduction angle applied to the base of the switching element Q1 are compositely controlled, and the DC output voltage E01 is stabilized.

Also, in the second embodiment,
FIG. 5 shows a rectifier circuit system provided on the secondary side of the power supply circuit.
6 to 1 without being limited to the configuration shown in FIG.
It is also conceivable to adopt the configuration shown in FIG. FIG.
In the above, the secondary-side DC output voltage EO1 is connected to the secondary-side parallel resonance circuit (N2 // C2) by connecting a full-wave rectifier circuit composed of a bridge rectifier circuit DBR and a smoothing capacitor CO1 in the illustrated connection form. An arrangement is shown for obtaining this.

In FIG. 7, a rectifier diode DO1, a rectifier diode DO2, and a smoothing capacitor CO are connected to the secondary side parallel resonance circuit (N2 // C2) as shown in FIG.
By connecting A and COB, a voltage doubler rectifier circuit using a full-wave rectification method is formed. In this case, a secondary side DC output voltage EO1 corresponding to twice the alternating voltage level generated in the secondary winding N2 is obtained at both ends of the series connection circuit of the smoothing capacitors COA-COB. .

The configuration of the secondary side shown in FIG.
A secondary-side series resonance capacitor Cs is connected in series to the winding end of the secondary winding N2. Thereby, on the secondary side of the insulating converter transformer PIT, a secondary side series resonance circuit is formed by the leakage inductance of the secondary winding N2 and the capacitance of the secondary side series resonance capacitor Cs. Therefore, in this case,
Primary parallel resonance circuit (N1, Cr) provided on the primary side
And a secondary-side series resonance circuit (N2, C
s) constitutes a composite resonance type converter. Then, by connecting a bridge rectifier circuit DBR and a smoothing capacitor CO1 to the secondary-side series resonance circuit as shown in FIG. 8, the secondary-side series resonance circuit (N2,
A full-wave rectifier circuit for full-wave rectification of the secondary side series resonance voltage due to the resonance action of Cs) is formed. Then, at both ends of the smoothing capacitor CO1, a secondary side DC output voltage EO1 corresponding to an equal multiple of the alternating voltage level generated in the secondary winding N2 is obtained.

In FIG. 9, a rectifier diode DO1, a rectifier diode DO2, and smoothing capacitors COA and COB are connected to the secondary side series resonance circuit (N2, Cs) as shown in FIG. A voltage rectifier circuit is formed.

In FIG. 10, two sets of secondary side series resonance capacitors Cs1 and Cs2 are connected to the secondary winding N2 as shown, and four rectifier diodes DO1, DO2 and D03 are further connected. , DO4 by a connection form shown in the figure to form a secondary-side rectifier circuit. A quadruple voltage rectifier circuit is formed as the secondary rectifier circuit configured in this manner. In describing the operation of this quadruple voltage rectifier circuit, the description of [series resonance capacitor Cs1, rectifier diode DO]
1, DO2, smoothing capacitor COA]. First, during the period when the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on,
The rectified current rectified by the rectifier diode DO2 is converted to the series resonance capacitor Cs by the series resonance effect of the leakage inductance of the secondary winding N2 and the series resonance capacitor Cs1.
The operation of charging 1 is obtained. Then, during a period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectifying operation is performed, a series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the secondary winding N2. And smoothing capacitor COA
Is charged. By performing the rectifying operation as described above, a DC voltage (rectified smoothed voltage) corresponding to almost twice the induced voltage of the secondary winding N2 is obtained at both ends of the smoothing capacitor COA. The same operation is also performed by a rectifier circuit including a set of [series resonance capacitor Cs2, rectifier diodes DO3, DO4, and smoothing capacitor COB], so that both ends of the smoothing capacitor COB have almost the induced voltage of the secondary winding N2. A DC voltage corresponding to twice is obtained.

As a result of the voltage doubler rectifying operation performed by the rectifier circuits of the respective stages as described above, the voltage induced by the secondary winding N2 is applied across the smoothing capacitor COA and the smoothing capacitor COB connected in series. The secondary side DC output voltage EO1 corresponding to almost four times is obtained.

In the power supply circuit according to the second embodiment which can adopt such a configuration, for example, the same operation as that described with reference to the waveform diagrams of FIGS. 2 and 3 is obtained.
A constant voltage control characteristic similar to that shown in FIG. 4 is obtained. Also, the orthogonal control transformer PRT is omitted. Therefore, also in the second embodiment, the same functions and effects as those of the first embodiment shown in FIG. 1 can be obtained.

Although the embodiments have been described above, the present invention is not limited to the configurations shown in the drawings as the above embodiments. For example, in each of the above-described embodiments, the case of a single-end system including one set of switching elements is described. However, a so-called push-pull system including two sets of switching elements is used.
It may be a self-excited voltage resonance type converter. Also, the secondary side may be provided with a rectifier circuit having a circuit configuration other than that shown in each drawing.

[0073]

As described above, according to the present invention, the primary-side voltage resonance type converter of the composite resonance type converter is of the self-excited type, and the MOS-FE is used for the constant voltage control.
A control voltage variably controlled according to the level of the secondary DC output voltage is applied to the T gate electrode. Then, the capacitance of the self-excited oscillation circuit changes depending on the conduction state of the MOS-FET, and the switching frequency is variably controlled. Thereby, the switching element
Switching control is performed by a complex control method in which the conduction angle and the switching frequency are simultaneously varied. Or it will be controlled to be more. As a result, an excessive voltage is not applied to the switching element, so that it is not necessary to provide a circuit for overload protection and a soft start circuit.
As a result, for example, the power supply circuit can be reduced in size and weight and cost can be promoted.

Further, by suppressing the voltage level applied to the switching element as described above, it is possible to select a low withstand voltage product for the switching element.
As a result, the deterioration of the switching characteristics can be avoided, and the power conversion efficiency can be improved. Further, the MOS-FET having a low withstand voltage and a small capacity is preferable. Further, the provision of the damper diode in parallel with the switching element prevents a negative damper current from flowing through the base-collector of the switching element, thereby stabilizing the circuit operation.

Further, in the configuration of the constant voltage control of the present invention, it is possible to omit the orthogonal control transformer, so that the variation of the control range of the switching frequency due to the gap at the time of manufacturing the orthogonal control transformer PRT. Will be solved. Therefore, it is possible to set a small margin with respect to the range of the AC input voltage, so that the circuit design becomes easy.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part in the power supply circuit of the embodiment.

FIG. 3 is a waveform chart showing a control operation of a primary-side switching frequency when a load is short-circuited in the power supply circuit according to the embodiment;

FIG. 4 is an explanatory diagram illustrating constant voltage control characteristics of the power supply circuit according to the embodiment;

FIG. 5 is a circuit diagram showing a configuration example of a power supply circuit according to a second embodiment of the present invention.

FIG. 6 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 7 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 8 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 9 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 10 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 11 is a circuit diagram showing a configuration example of a power supply circuit as a prior art.

12 is a waveform chart showing an operation of a main part in the power supply circuit shown in FIG.

13 is an explanatory diagram showing constant voltage control characteristics of the power supply circuit shown in FIG.

FIG. 14 is a waveform diagram showing a control operation of a primary-side switching frequency when a load is short-circuited in the power supply circuit shown in FIG. 11;

[Explanation of symbols]

1 control circuit, Q1 switching element, Q2 MOS
-FET PIT isolated converter transformer, N1 primary winding, N2 secondary winding, N3 tertiary winding, D2 damper diode, Cr primary side parallel resonance capacitor, NB
Drive winding, CB1 resonance capacitor, CB2 capacitance variable capacitor, LB inductor, PC photocoupler

Claims (3)

[Claims] A switching means formed of a single switching element for switching a DC input voltage; an insulating converter transformer for transmitting an output of the switching means obtained on a primary winding to a secondary winding; A primary-side parallel resonance circuit formed by a primary winding of the insulating converter transformer and a primary-side parallel resonance capacitor and provided so that the operation of the switching means is of a voltage resonance type; A secondary-side parallel resonance circuit formed by connecting a secondary-side parallel resonance capacitor in parallel to the secondary winding, and a rectifying operation by inputting an alternating voltage obtained in the secondary-side parallel resonance circuit. DC output voltage generating means configured to obtain a DC output voltage by performing Switching drive means configured to perform a switching operation by applying a switching drive signal to the switching element, the switching drive means being configured as a series resonance circuit including a drive winding wound around the inductor, an inductor, a capacitor, and a resistor. A series circuit of a capacitor and a MOS-FET connected in parallel with the capacitor; and a control voltage that is varied according to the level of the DC output voltage obtained by the DC output voltage generating means.
A constant voltage control unit configured to control a switching frequency supplied to the switching element by applying the voltage to the gate of the ET to perform constant voltage control on the DC output voltage. Characteristic switching power supply circuit. 2. The switching device according to claim 1, wherein said switching element is formed by a bipolar transistor.
3. The switching power supply circuit according to item 1. 3. The switching power supply circuit according to claim 1, wherein a damper diode is connected in parallel with said switching element.