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GB2510393A - An envelope-tracking amplifier with a linear amplifier having an output offset current for improved efficiency - Google Patents

An envelope-tracking amplifier with a linear amplifier having an output offset current for improved efficiency Download PDF

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Publication number
GB2510393A
GB2510393A GB1301852.8A GB201301852A GB2510393A GB 2510393 A GB2510393 A GB 2510393A GB 201301852 A GB201301852 A GB 201301852A GB 2510393 A GB2510393 A GB 2510393A
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GB
United Kingdom
Prior art keywords
current
voltage
offset current
output
power supply
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
GB1301852.8A
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GB201301852D0 (en
Inventor
Gerard Wimpenny
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Nujira Ltd
Original Assignee
Nujira Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Nujira Ltd filed Critical Nujira Ltd
Priority to GB1301852.8A priority Critical patent/GB2510393A/en
Publication of GB201301852D0 publication Critical patent/GB201301852D0/en
Priority to CN201911066474.5A priority patent/CN110708026B/en
Priority to CN201480018288.3A priority patent/CN105103441B/en
Priority to EP19201054.4A priority patent/EP3667905B1/en
Priority to KR1020157023234A priority patent/KR20150117678A/en
Priority to JP2015555733A priority patent/JP6452623B2/en
Priority to PCT/EP2014/051967 priority patent/WO2014118347A1/en
Priority to US14/764,806 priority patent/US9634612B2/en
Priority to EP14702042.4A priority patent/EP2951920B1/en
Publication of GB2510393A publication Critical patent/GB2510393A/en
Withdrawn legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0216—Continuous control
    • H03F1/0222—Continuous control by using a signal derived from the input signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0216—Continuous control
    • H03F1/0222—Continuous control by using a signal derived from the input signal
    • H03F1/0227—Continuous control by using a signal derived from the input signal using supply converters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H03F3/19—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/211—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/217—Class D power amplifiers; Switching amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/30—Single-ended push-pull [SEPP] amplifiers; Phase-splitters therefor
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04W—WIRELESS COMMUNICATION NETWORKS
    • H04W52/00—Power management, e.g. TPC [Transmission Power Control], power saving or power classes
    • H04W52/02—Power saving arrangements
    • H04W52/0209—Power saving arrangements in terminal devices
    • H04W52/0251—Power saving arrangements in terminal devices using monitoring of local events, e.g. events related to user activity
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/102—A non-specified detector of a signal envelope being used in an amplifying circuit
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/135—Indexing scheme relating to amplifiers there being a feedback over one or more internal stages in the global amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/451—Indexing scheme relating to amplifiers the amplifier being a radio frequency amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2203/00—Indexing scheme relating to amplifiers with only discharge tubes or only semiconductor devices as amplifying elements covered by H03F3/00
    • H03F2203/20—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F2203/21—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F2203/211—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H03F2203/21106—An input signal being distributed in parallel over the inputs of a plurality of power amplifiers
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
    • Y02D30/00—Reducing energy consumption in communication networks
    • Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Amplifiers (AREA)
  • Transmitters (AREA)

Abstract

The dissipations in the sourcing 250 and sinking 252 output portions of a linear amplifier in an envelope tracking power supply are made equal, despite the asymmetric nature of the envelope waveform (figure 4), by providing an offset current Ios that makes the average pull-up current less than the average pull-down current, thereby improving the efficiency of the output stage. The offset current may be fed through an inductor 256 from a controlled voltage Vsupply2 provided by a switch mode converter under control of a loop (figure 8) acting to equalise the dissipations in the sourcing and sinking parts. Alternatively the control loop may apply an offset current proportional to the difference between the mean and mid-point voltages of the envelope waveform

Description

I
IMPROVED EFFICIENCY FOR LINEAR AMPLIFIER OF ENVELOPE TRACKING
MODULATOR
BACKGROUND TO ThE INVENTION:
Field of the Invention:
The invention relates to envelope tracking modulated power supplies suitable for radio frequency power amplifier applications. The invention is particularly concerned with such power supplies in which a reference signal is used as an input to a low frequency path and a high frequency path, and in which each path generates separate outputs which are combined to I on a supply voltage.
Description of the Related Art:
Envelope tracking power supplies for radio frequency power amplifiers are well-known in the art. Typically a reference signal is generated based on an envelope of an input signal to be amplified. .An envelope tracking power supply generates a supply voltage for the power amplifier which tracks the reference signal.
Figure 1 shows a prior art envelope tracking lET) modulator architecture in which a frequency splitter 12 is used to divide an incoming envelope reference signal on line 10 into a high frequency (HF) path signal on line 14 and a low frequency (LF) path signal on line 16. The frequency splitter 12 may include a low pass filter 18 in the low frequency path and a high pass filter 20 in the high frequency path. The signal in the LF path on line 16 is amplified by an efficient switched mode amplifier 22, and the signal in the HF path on line 14 is amplified by a wideband linear amplifier 24. A frequency __._ sa aa as.
selective combiner 26 is used to combine the signals in the LF and HF paths after amplification. In Figure 1 the combiner 26 is illustrated as including a low frequency combining element (and high frequency blocking element) 28 in the low frequency path, and a high frequency combining element (and low frequency blocking element) 30 in the high frequency path. A combined signal from the combiner 26 on line 32 provides a feed to a load 34 which for the purposes of example is illustrated as a resistor. In a typical application the load is a power amplifier (PA) * and the reference signal is derived from an input signal to be amplified by the power amplifier.
An example of a power amplifier system Incorporating a supply architecture such as illustrated in Figure 1 can be found in "Hand Separation and Efficiency Optimisation in IS Linear-Assisted Switching Power Amplifiers, Yousefzadeh et al, [IEEE Power Electronics Specialists conference 2005) Figure 2 shows an alternative prior art arrangement in which the frequency selective combiner 26 is an inductor-capacitor (LO) combiner. The low frequency combining element is an inductor 28a, and the high frequenry combining element is a capacitor Boa. In this arrangement a feedback path 35 takes a signal from the combiner (or modulator) output on line 32, to the input of the linear amplifier 24. The signal on the feedback path 35 is subtracted from the signal in the high frequency path on line 14 by subtractor 38, to provide an input to the linear amplifier 24. The inclusion of the feedback path 36 achieves improved tracking accuracy compared to the arrangement of Figure 1.
An example of a power amplifier system incorporating a supply architecture such as illustrated in Figure 2 can be found in "Efficiency Optimisation in Linear-Assisted Switching Power Converters for Envelope Tracking in RF Power Amplifiers", Yousefzadeh et ci, [IEEE Symposium on Circuits and Systems 2005] It is an aim of the invention to provide an envelope tracking modulated power supply which. offers improvements over the prior art, such as the art ngements of Figures 1 and. 2 -
SUMMARY OF TIlE INVENTION
The invention provides an. envelope tracking power supply arranged to generate a modulated supply voltage in dependence on a reference signal, comprising a first Dath for tracking low frequency variations in the reference signal and a second path for tracking high frequency variations in the reference signal, the second path including a linear amplifier, wherein the output stage of the linear amplifier comprises a current source and a current sink connected to the high frequency output, there further being provided a DC offset current at the high frequency output.
The DC offset current may be chosen to minimise the power dissipated in the output stage of the linesr amplifier.
The DC offset current may he derived from a further voltage supply which is lower than the output stage voltage supply.
The DC offset current may be provided via an inductor connected between the further power supply and the high frequency output.
The envelope tracking power supply may further comprise sensing the cower difference in an output, and integrating the sensed power difference to control a switch mode converter to generate a second supply voltage to generate the DC offset current. Sensing the power difference may comprise measuring a suppiy voltage for generating the DC offset current, the output voltage, the source current and the sink current.
A taroet DC offset current may be determined in dependence on the difference between the input volt age waveform and the halved sum of the maximum and minimum voltage levels of the input waveform voltage. An error between the target DC offset current and a measured DC offset current may integrated and used to control a switch mode converter to generate a second supply voltage to generate the DC offset current.
An RE' amplifier may include an envelope tracking power supply.
A mobile device for a mobile communication system may include an envelope tracking power supply.
An infrastructure element for a mobile communications system may include an envelope tracking power supply.
The invention further provides a method for an envelope tracking tower supply arranqed to generate a modulated supply voltage in dependence on a reference siqnal, compri sing provid.inq a first math for tracking low frequency variations in the reference signal and providing a second path for tracking high frequency variations in the reference signal.
the second path including a linear amplifier, wherein the output stage of the linear amplifier comorises a current source and a current sink connected to the high frequency output, the method further comprising providing a DC offset current at the high frequency output.
HPIEF DESCPIPTION OF THE FIGURES: The invention is now described by w. y of example with reference to the accompanying Figures, in which: Figure 1 illustrates a prior art envelope tracking modulated supply with high and low frequency paths; Figure 2 illustrates a prior art envelope tracking modulated supply incorporating feedback in the high frequency path; Figure 3 illustrates a modified implementation an output of a linear amplifier in accordance with the arrangement of Figure 1 or Figure 2; Figure 4 (a) to 4 (c) illustrate current flow in the arrangement of Figure 3; Figure 5 illustrates the implementation of the output of a linear amplifier in the arrangement of Figure 1 or Figure 2 in accordance w.iti. an embodiment of the present invention; Figures 6(a) to 6(c) illustrate current flow in the the arrangement of Figure 5; Figures 7(a) and 7(d) illustrate waveform plots in the arrangement of Figures 3 and 5; Figure 8 illustrates the implementation of the output of a linear amplifier in the arrangement of Figure 1 or Figure 2 in accordance with an exemolary embodiment of the present invention; and Figure 9 illustrates the implementation of the output of a linear amplifier in the arrangement of Figure 1 or Figure 2 in accordance with another exemplary embodiment of the present invent ion.
DESCRIPTION OF THE PREFERRED ErnODIMENTS
In the following description the invention is described with reference to exemplary embodiments and implementations. The invention is not limited to the specific details of any arrangements as set out, which are provided for the purposes of understanding the invention.
Embodiments of the invention may apply to different feedback architectures for the linear amplifier in the high frequency path. The invention and its embodiments are riot limited to a 0 particular feedback arrangement in the high frequency path.
For example in the foregoing illustration of Figure 2 an arrangement is illustrated in which the linear amplifier receives a feedback from the outnut of the combiner. The present invention may also apply, for example, to an arrangement in which the linear amplifier receives a feedback from the output of the linear amplifier at the ii.put to the combiner, and in which the path containing the linear amplifier does not include a high frequency filter such as filter 20 in Figure 2, the linear amplifier path receiving the full--spectrum reference signal.
In general in a hybrid envelope tracking modulator (i.e. an architecture using a switched mode amplifier and a linear amplifier) as illustrated in Figure 2, a significant proportion of the total modulator power dissipation occurs in the output stage of the linear amplifier This can be understood with reference to Figure 3, which illustrates an exemplary implementation of the linear amplifier 24 Class B output stage. As illustrated, a current source 250 is connected between a supply voltage and a common node 254, and a current sink 252 is connected between t:ne common node 254 and electrical ground V. An instantaneous source current rsc flows in the current source element 250, and an instantaneous current I< flows in the current sink element 252. At any given instant current flows in either the source device 250 or the sink device 252, and the current in the inactive device is zero An output voltage V is formed at node 254. The combiner capacitor 30a of Figure 2 is illustrated as cormected between the node 254 and the output of the combiner. A current k flows-S the combiner capacitor BOa, For the purposes of example, the arrangement of Figure 3 shows a feedlac k path 40 which renresents a feedback from the output of the linear amplifier, before the combiner, to the input of the linear amplifier. The feedback is not described in more detail. herein because it does not form part of the present invention. The current flow in the feedback path is assumed to be sufficiently low to be ignored.
No DC current can flow through the combiner capacitor 30a.
Hence in the prior art arrangement of Figure 3 the value of the average source current I from current source 250 must be equal to the value of the average sink current I< from current sink 252.
In general the recuired modulator output voltage provided by the linear amplifier 24 may typically exhibit significant asymmetry, and this in turn results in asymmetry of the output current I. of the linear amplifier 24.
This is illustrated by the waveform of Figure 4(a) which shows a plot of ouput current I against time. The current above the zero level 302 represents output positive currents which flow in the source transistor 250, and the current below the zero level 302 represents the output negative currents which flow in the sink transistor 252. The combined source and sink currents represent the output current I. The values of each of the average source $Rc and sink I currents are equal as shown in Figures 4(b) and 4(c), which shows plots of the source and sink currents against time.
The line 304 in Figure 4(b) shows the average current in source device 250 and the line 306 in Figure 4(c) shows the average current in the sink device 252. The average current in source device 250 is equal to the average current in sink device 252.
However in the example as shown the power dissipated in the upper device (the current source 250) is much greater than the power dissipated in the lower device (current sink 252). This disparity in power dissipation is due to the waveform asymmetry and results in much higher voltages across the upper (source) device.
It can thus be seen that the necessity for the average sink and source currents to be equal for the output topology of Figure 3 is disadvantageous.
In accordance with a preferred embodiment of the invention, an extra voltage supply is used to add a DC (or low frequency) offset current via an inductor to the output node of the linear amplifier 24. There is thus no longer a requirement for the average source and sink currents to be equal.
Figure 5 shows such a modified topology. The arrangement of Figure 3 is modified such that an inductor 256 is included between a second supply voltage vLfl and the node 254. The inductor 256 provides an offset current I<, which flows in the inductor 256 from the voltage supply VnPhn.
The instantarxeous current in the current source 250 is modified to and the instantaneous current irs the current sink 252 is The output current I flows in the output capacitor 30a, and the output voltage V is formed at the node S 254.
The waveform of Figure 6(a) shows the output current I of the linear amplifier 24, which is the same as that shown in Figure 4(a) In accordance with the invention therefore, the output curren.t I of the linear amplifier is unchanged. As illustrated in Figure 6, the portion of the output current above the line 602 is provided by the source transistor 250, The portion of the output current below the line 602 is provided by tile sink transistor 252, Figures 6 (b) and 6(c) show the modified source and sink currents from the current source element 250 and current sink element 252 respectively. In the example shown the mod.i fled source current 1s is decreased by offset current ios and the modified sink current is increased by offset current los. The lines 604 and 606 in the respective Figures 6(b) and 6(c) represent the modified average currents flowing in the respective source and sink transistors.
As illustrated, by comparing Figures 4(b) and 4(c) with Figures 6 (b) and 6(c) , the effect of the additional offset current I supplied via inductor 256 is to reduce the average source current by I from I to I, and. in crease the average sink current by I from SK to SMK' This reduces the power dissinated in the current source 250 and increases the power dissipated in the current sink 252, Figure 7(a) shows the dissipation 702, 704 in the output stage source and sink current devices 250 and 252 I0 respectively, and the total dissipation 706, as a function of the offset current I for the arrangement of Figure 5.
It can he seen that for the particular waveform illustrated the minimum dissipation, in Figure 7(a) is approximately 20% less than the dissipation with no offset current. This difference in dissipation is strongl.y dependant on the waveform asymmetry and is larger for more asymmetric waveforms.
The instantaneous power dissipation in the source and sink output. devices 250 and 252 cannot easily be directly measured.
hut the average current through the source and sink devices 250 and 252 and the average output voltage can all be readily measured. Hence it is possible to calculate the sensedt powers as a proxy for the dissipated powers using these average parameters.
Figure 7 (b) shows the sensed powers for the source. device 710, sink device 708 and the total sense.d power 712.
inductor 250 ideally has zero DC resistance, hence the DC voltage at both terminals of the inductor 250 is the same.
Referring to Figure 5, the sensed source power can be calculated as: avg x avg (ISRC') The sensed sink power can be calculated as: avg (V) x avg (I) wh.ere = the supply voltage applied to the feed inductor; = the output voltage of the stage; avg (Ia) = the average source current; and avg (Ise) = the average sink current., The minimum in total sensed power occurs at the same value of offset current as the minimum dissipated power, as sbowx in Figure 7, hence minimising sensed power maximises the efficiency of the supply modulator.
The additional voltage supply Vppj in Figure s is assumed to be generated using a high efficiency power converter, and the power loss In feed inductor 256 is assumed to be minimal.
Figure 8 shows a direct technique for generating the offset current in which a negative feedback loop may he used to minimise the total sensed power by mi.nimising the difference between the two sensed powers, by integrating the error to make small adjustments to the-output voltage of the second supply VSUPPLY2 Figure 8 shows the output stage of the error amplifier comprising current source 250, current sink 252, combining capacitor 30a and DC current offset feed inductor 256. The supply voltage Vgy is provided by a switch mode converter 810, which is connected to a suppl.y voltage denoted by reference numeral 814.
The input to the switch mode converter 810 is provided by an integrator 816. The input to the integrator 816 is provided by a signal processing block 818, which generates a signal representing the sensed power difference on line 818 to the input of the integrator 816 based on the second supply voltage Vsupply2, the average output voltage Yea, and the average of the source and sink currents $RC' and iNx An indirect method of controlling the offset current exploits the fact that the offset current required depends on the asymmetry of the waveform. If the waveform is symmetrical the mean voltage lies midway between the minimum and maximum values of the waveform. If the mean voltage is less than midway between the minimum and maximum values of the waveform a positive offset current is required to minimise the output stage power dissipation. Similarly if the mean voltage is greater than midway between the minimum and maximum values of the waveform a negative offset current is required to minimise the output stage dissipation.
Figure 9 shows a control loop for implementing this indirect concept.
The control loop includes the current source 250 and current sink 252 of the output stage, the combining capacitor 30a, and the DC current offset feed inductor 256. The inductor 256 is connected to the node 254 via current sense resistor 800.
The supply voltage Vsupply2 is provided by a switch mode converter 802, which is connected to a supply voltage Vp denoted by reference numeral 804.
The input to the switch mode converter is provided by an integrator 806. A first input of the integrator is provided by a subtractor 808, which provides a difference on line 812 between the voltage which is midway between the minimum and maximum values of the input waveform (equal to (Vmax+VThin) /2) and the input waveform yin 810 to give a voltage representative of the offset current target on line 814 at the first input to the integrator 806. The second input to the integrator 806 is provided by a voltage source 816, which measures the current in the resistor 800 and provides a voltage representing the offset current.
The current offset target on line 814 is set as the difference between the mean and median waveform voltages as described above. The error between the target and measured offset current is integrated by integrator 806 and used to control the switch mode converter 802 which generates the second supply voltage VpLi2 which supplies the offset current to the linear amplifier output stage via the inductor 256.
The czeneration of the offset current and the second supply may be achieved in a number of ways, both indirectly and directly, and the invention is not limited to any particular technique As discussed above the present invention may he applied to the output of a linear amplifier in a correction path of a modulated power supply, such as the linear amplifier of Figure I or Figure 2.
Such modulated power supplies may be used to provide the modulated power supply to an HF amplifier, which may comprises the load of. Figure 1 or Figure 2.
HF amplifiers are used in mobile communication systems, in wireless devices and wireless infrastructure.
The. invention arid its embodiments relates to the application of envelope tracking (ET) to radio frequency (HF) power amplifiers and is applicable to a broad range of implementations including cellular handsets, wireless infrastructure, and military power amplifier applications at high frequencies to microwave frequencies.
The invention has been described herein by way of example with reference to embodiments. The invention is not limited to the described embodiments, nor to specific combinations of features in emhodiwents. Modifications may be made to the embodiments within the scope of the invention. The scope of the invention is defined by the appended claims, , 4 tflfl fl4

Claims (18)

  1. CLAIMS: 1. An envelope tracking power supply arranged to generate a modulated supply voltage in dependence on a reference signal, comprising a first path for tracking low frequency variations in the reference signal and a second path for tracking high frequency variations in the reference signal, the second path including a linear amplifier, wherein the output stage of the linear amplifier comprises a current source and a current sink connected to the high frequency output, there further being provided a DC offset current at the high frequency output.
  2. 2. The envelope tracking power supply of claim 1 wherein the DC offset current is chosen to minimise the power dissipated in the output stage of the linear amplifier.
  3. 3. The envelope tracking power supply of claim 1 or claim 2 wherein the DC offset current is derived from a further voltage supply which is lower than the output stage voltage supply.
  4. 4. The envelope tracking power supply of any preceding claim wherein the DC offset current is provided via an inductor connected between the further power supply and the high frequency output.
  5. 5. The envelope tracking power supply of any preceding claim further comprising sensing the power difference in an output, and integrating the sensed power difference to control a switch mode converter to generate a second supply voltage to generate the DC offset current. I5
  6. 6. The envelope tracking power supply of claim S wherein sensing the power difference comprises measuring a supply voltage for generating the DC offset current, the output voltage, the source current and the sink current.
  7. 7. The envelope tracking power supply of any one of claims 1 to 4 wherein a target DC offset current is determined in dependence on the difference between the input voltage waveform and the halved sum of the maximum and minimum voltage levels of the input waveform voltage.
  8. 8. The envelope tracking power supply of claim 7 wherein an error between the target DC offset current and a measured DC offset current is integrated and used to control a switch mode converter to generate a second supply voltage to generate the DC offset current.
  9. 9. Mi RF amplifier including an envelope tracking power supply according to any one of claims 1 to 8.
  10. 10. A mobile device for a mobile communication system including an envelope tracking power supply according to any one of claims 1 to 9.
  11. 11. An infrastructure element for a mobile communications system including an envelope tracking power supply according to any one of claims 1 to 10.
  12. 12. A method for an envelope tracking power supply arranged to generate a modulated supply voltage in dependence on a reference signal, comprising providing a first path for tracking low frequency variations in the reference signal and providing a second path for tracking high frequency variations in the reference signal, the second path inc].uding a linear amplifier, wherein the output stage of the linear amplifier comprises a current source and a current sink connected to the S high frequency output, the method further comprising providing a DC offset current at the high frequency output.
  13. 13. The method of claim 12 wherein the DC offset current is chosen to minimise the power dissipated in the output stage of the linear amplifier.
  14. 14. The method of claim 12 or claim 13 wherein the DC offset current is derived from a further voltage supply which is lower than the output stage voltage supply.
  15. 15. The method of any one of claims 11 to 14 wherein the DC of feet current is provided via an inductor connected between the further power supply and the high frequency output.
  16. 16. The method of any one of claims 11 to 15 further comprising sensing the power difference in an output, and integrating the sensed power difference to control a switch mode converter to generate a second supply voltage to generate the DC offset current.
  17. 17. The method of claim 16 wherein sensing the power difference comprises measuring a supply voltage f or generating the DC offset current, the output voltage, the source current and the sink current.
  18. 18. The method of any one of claims 11 to 15 wherein a target DC offset current is determined in dependence on the difference between the input, voltage waveform and the halved sum of the maxinrum and minimum voltage levels of Lhe input wvef arm voltage.
    19, The method of claim 18 wherein an error between the target DC offset: current and a measured DC offset current is integrated and used to control, a switch mode converter to generate a second supply voltage to generate the DC offset current.
GB1301852.8A 2013-02-01 2013-02-01 An envelope-tracking amplifier with a linear amplifier having an output offset current for improved efficiency Withdrawn GB2510393A (en)

Priority Applications (9)

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GB1301852.8A GB2510393A (en) 2013-02-01 2013-02-01 An envelope-tracking amplifier with a linear amplifier having an output offset current for improved efficiency
EP14702042.4A EP2951920B1 (en) 2013-02-01 2014-01-31 Improved efficiency for linear amplifier of envelope tracking modulator
KR1020157023234A KR20150117678A (en) 2013-02-01 2014-01-31 Improved efficiency for linear amplifier of envelope tracking modulator
CN201480018288.3A CN105103441B (en) 2013-02-01 2014-01-31 Improve the linear amplifier of the envelope-tracking modulator of efficiency
EP19201054.4A EP3667905B1 (en) 2013-02-01 2014-01-31 Improved efficiency for linear amplifier of envelope tracking modulator
CN201911066474.5A CN110708026B (en) 2013-02-01 2014-01-31 Linear amplifier for envelope tracking modulator with improved efficiency
JP2015555733A JP6452623B2 (en) 2013-02-01 2014-01-31 Improved efficiency of the linear amplifier of the envelope tracking modulator
PCT/EP2014/051967 WO2014118347A1 (en) 2013-02-01 2014-01-31 Improved efficiency for linear amplifier of envelope tracking modulator
US14/764,806 US9634612B2 (en) 2013-02-01 2014-01-31 Efficiency for linear amplifier of envelope tracking modulator

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US9634612B2 (en) 2017-04-25
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CN105103441B (en) 2019-11-29
US20160006396A1 (en) 2016-01-07
CN105103441A (en) 2015-11-25
KR20150117678A (en) 2015-10-20
WO2014118347A1 (en) 2014-08-07
GB201301852D0 (en) 2013-03-20
EP2951920B1 (en) 2019-12-11
EP3667905B1 (en) 2021-07-14
JP6452623B2 (en) 2019-01-16
EP3667905A1 (en) 2020-06-17
EP2951920A1 (en) 2015-12-09
JP2016505230A (en) 2016-02-18

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