GB2393865A - A transimpedance amplifier for detecting very small chromatograph or mass spectrometer currents with a greater bandwidth - Google Patents
A transimpedance amplifier for detecting very small chromatograph or mass spectrometer currents with a greater bandwidth Download PDFInfo
- Publication number
- GB2393865A GB2393865A GB0218082A GB0218082A GB2393865A GB 2393865 A GB2393865 A GB 2393865A GB 0218082 A GB0218082 A GB 0218082A GB 0218082 A GB0218082 A GB 0218082A GB 2393865 A GB2393865 A GB 2393865A
- Authority
- GB
- United Kingdom
- Prior art keywords
- amplifier
- output
- operational amplifier
- resistor
- transimpedance amplifier
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/04—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only
- H03F3/08—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only controlled by light
- H03F3/087—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only controlled by light with IC amplifier blocks
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Amplifiers (AREA)
Abstract
A transimpedance amplifier for detecting very small dynamic currents (less than a picoampere) comprises an operational amplifier 204 with negative feedback through a parallel RC network C2,R2. The feedback network is coupled to a lag-lead tap 218 on a series output RC network R1,R3,C1 with a similar time constant. The provision of the small resistor R3 increases the operable bandwidth of the amplifier allowing relatively small currents to be measured with greater accuracy. The transient response exhibits reduced ringing. The introduction of a second operational amplifier (307,figure 4) in the forward path of the loop increases open loop gain, thereby improving gain accuracy. Bandwidth and response time can be further improved by providing the second amplifier with a differentiating (lead) response (C4,figure 5). The improved amplifier may be used in a chromatograph or a mass spectrometer.
Description
A Transimpedance amplifier.
The present invention relates to transimpedance amplifiers, particularly to lo transimpedance amplifiers for measuring small signals having bandwidths of a few hertz to millihertz.
Relatively simple apparatus such as, for example, electrometers, pH meters and vacuum ion gauges, and more complex systems such as, for example, mass spectrometers, optical 5 spectrometry and chromatography, require accurate measurement of very small currents typically in the order of one picoampere (1 x10-2 Amperes) to one femtoampere (lxlO-'5 Amperes). As the source of these small currents is invariably of high impedance, the commonly used to and preferred measurement circuitry is the transimpedance amplifier.
Referring to Figure 1, a known circuit configuration of a transimpedance amplifier 100 comprises a current source 102, an operational amplifier 104 having inverting and non inverting inputs 106 and 108, respectively, and an output 110 having a resistor R1 25 connected in series with a capacitor C1, a sampling network 112, a feedback network 114 having a resistor R2 connected in parallel with a capacitor C2 and a summing node 117.
The current source 102 is connected to the inverting input 106 of the operational amplifier 104. The non- inverting input 108 ofthe operational amplifier 104 is connected 30 to earth 116. At the output 110, a first terminal of the resistor R1 is connected to the output and a second terminal is connected to a tap 1 18. A first terminal of the capacitor
Cl is also connected to the tap. The second terminal ofthe capacitor C1 is connected to earth 1 16. Connected to the tap 1 18 is the sampling network 1 12. The feedback network 114 leads from the sampling network to the inverting input 106 of the operational amplifier 104, that is between the output 110 and the input 106. The point at which the 5 feedback network 114 is connected to the input 106 is the summing node 117 where the output signal sampled by the sampling network 112, and having been fed back via the feedback network 114, is summed with the input signal. C3 represents the capacitance of the signal source, any connecting conductors and of the operational amplifier.
lo The voltage at the inverting input 106 is very nearly the same potential as it is at the non inverting input 108. As the non-inverting input 108 is connected to earth 116, the inverting input 106 is said to be at, so called, virtual earth. Transimpedance amplifiers are commonly used as a current measuring instrument since they are essentially an ammeter with substantially zero voltage across the meter. However, they have undesirable limitations when measuring small currents such as, for example, of between 1 p i c o ampere to 1 femto amp ere.
The circuit is operable under the condition Rlx Cl=R2xC2. In order to detect such small signals it is necessary for the feedback resistor R2 to be large, such as, for example 20 between 100 Giga-ohm to 100 Tera-ohm. The need to use resistors of such high value, in combination with incidental and stray capacitance, results in bandwidths of approximately one Hertz or less. Consequently, this results in relatively slow response time to changes in signal. Under this condition the performance of the circuit, and the components therein, are severely compromised as in use they are operating on the edge of 25 stability, resulting in lightly damped ringing oscillations on the output for changes in the input current, therefore rendering the circuit practically unusable for measuring such small currents.
In applications such as, for example, the elusion of fractions from a chromatograph or the 30 output signals from a static vacuum mass spectrometer, the changes in signals are not controllable in terms of time response and therefore the measured signal can be
substantially misleading and unreliable. For example, in relation to the elusion of fractions from a chromatograph, the peaks of the signal to be measured will have passed before the amplifier can properly respond to them, thereby giving erroneous readings. In another example, a mass spectrometer may be used to study isotopic ratios in rare gas 5 such as, for example, Helium emissions from the environs of volcanoes in an attempt to predict eruptions thereof. The relatively small samples of rare gas available in combination with the need to measure accurately the relatively large ratios between the isotopes, which also decay in intensity in time, means that a short response time and sensitivity are desirable features of the measuring apparatus.
It is an object of the present invention to provide a transimpedance amplifier capable of measuring small currents and having a relatively short response time.
According to the present invention a transimpedance amplifier comprises a current s source, having a capacitance C3, connected to signal amplification means having an output, said output comprising a resistor Rl connected to a capacitor C1 in series between said output and earth, a sampling network, connected to the output via a tap connected between the resistor R1 and the capacitor C1, operable to sample an output signal, a feedback network, comprising a resistor R2 connected in parallel to a capacitor 20 C2 between the sampling network and a summing node, said summing node disposed at an input of the amplification means and operable to sum the sampled signal with an input signal Dom the current source, characterized in that the output further comprises resistor R3 connected in series between the tap and the capacitor C 1.
The amplification means may comprise an operational amplifier, having an inverting input, to which the current source is connected and a noninverting input connected to earth.
At the output of the amplifier a wide band output signal and a narrow band output signal may be measured. If required the wide band output signal and the narrow band output signal may be measured simultaneously.
5 Alternatively, the amplification means may comprise a plurality of operational amplifiers.
At least one operational amplifier may be connected in a non-inverting configuration and at least one operational amplifier may be connected in an inverting configuration.
At least one operational amplifier may be adapted to operate as a differentiator in the lo feedback network.
The amplification means may comprise a first operational amplifier and a second operational amplifier, the first operational amplifier having a non-inverting input, to which the current source is connected, an inverting input connected to earth, and an 5 output connected to the second operational amplifier, the second operational amplifier having an inverting input, to which is connected the output of the first operational amplifier, a non-inverting input, connected to earth, and an output providing the output of the transimpedance amplifier, and between the output of the first operational amplifier and the inverting input of the second operational amplifier, a capacitor C4 connected in 20 series with a resistor R4, across which a resistor R5 is connected in parallel, and a resistor R6 connected between the inverting input and the output of the second operational amplifier. The second operational amplifier is preferably adapted to operate at a wider bandwidth 25 than the first operational amplifier so as to reduce the influence of the second operational amplifier on the overall bandwidth of the transimpedance amplifier. The overall bandwidth of the transimpedance amplifier is preferably determined by the first operational amplifier.
For a transimpedance amplif er comprising a plurality of operational amplifiers, a wide band output signal, an intermediate band output signal and a narrow band output signal is measurable. These outputs may be measured simultaneously.
5 Advantageously, for the condition of (R1+R3) x C1 = R2 x C2, the value of R3 is substantially AoC,2 1 wherein AD is a zero-frequency open-loop gain of the operational amplifier and T is a lo time constant associated with a -3dB corner frequency fC=1/(2T).
The present invention will now be described further with reference to the accompanying drawings, in which: 5 Figure 1 is a circuit diagram of a transimpedance amplifier according to the prior art;
Figure 2 is a circuit diagram of a first embodiment of a transimpedance amplifier according to the present invention; 20 Figure 3 is a frequency response of the transimpedance amplifiers of Figures 1 and 2; Figure 4 is a circuit diagram of a second embodiment of a transimpedance amplifier according to the present invention; 25 Figure 5 is a circuit diagram of a third embodiment of a transimpedance amplifier according to the present invention; Figure 6 is a frequency response of the transimpedance amplifier of Figure 5; 30 Figure 7 is a transient response of the transimpedance amplifier of Figure 5;
Figure 8 is the transient response of Figure 7 at a larger scale; Referring to Figure 2 there is shown a first embodiment of a transimpedance amplifier 5 200 according to the present invention, comprising a current source 202, an operational amplifier 204 having inverting and non-inverting inputs 206 and 208, respectively, and an output 210, having a resistor R1 connected in series with a resistor R3 which in turn is connected in series to a capacitor C 1. The transimpedance amplifier 200 further comprises a sampling network 212, a feedback network 214 having a resistor R2 l o connected in parallel with a capacitor C2, and a summing node 217.
The current source 202 is connected to the inverting input 206 of the operational amplifier 204. The non- inverting input 208 of the operational amplifier 204 is connected to earth 216. A first terminal of the resistor R1 is connected to the output 210 and a 5 second terminal is connected to a tap 218. A first terminal of the resistor R3 is also connected to the tap 218. The second terminal of the resistor R3 is connected to a first terminal of capacitor C 1. The second terminal of the capacitor C 1 is connected to earth 216. Connected to the tap 218 is the sampling network 212. The feedback network 214 leads from the sampling network to the inverting input 206 of the operational amplifier 20 204, that is, between the output 210 and the input 206. The point at which the feedback network 214 is connected to the input 206 is the summing node 217 where the output signal sampled by the sampling network 212 is summed with the input signal, having been fed back via the feedback network 214. C3 represents the capacitance of the signal source, any connecting conductors and of the operational amplifier.
A wide-band response may be measured at the output at a terminal between the operational amplifier 204 and the resistor R1, and a narrow band response may be measured at the output at a terminal between the resistor R3 and the capacitor C 1. The wide-band and narrow-band responses may be measure simultaneously. The narrowband 30 output may be buffered if required.
As previously explained, in measuring small currents the performance of the transimpedance amplifier of the closest known prior art, as shown in Figure 1, is severely
compromised by the system being on the edge of instability resulting in lightly damped ringing oscillations on the output for changes of the input current. The introduction of
5 resistor R3 of the correct value allows the instability to be overcome giving much increased bandwidth and reduced rise-time with no transient overshoot or other undesirable consequences.
Mathematical analysis of the transimpedance amplifier of Figure 2, with the condition lo (R+R3)xC=R2xC2 and knowledge of the zero-frequency openloop gain AD of the operational amplifier and the time constant T associated with the -3dB corner frequency fc =l/(2T) the required value of R3 is given, with small approximation, by: R3 = 2( 2 2 3)1 ohm Figure 3 is a graph showing a comparison of the bandwidths of the known transimpedance amplifier of Figure 1 and that of the transimpedance amplifier of the first embodiment of the present invention of Figure 2. It can be seen that both the wide-band and the narrow-band responses of the present invention are much improved relative to the 20 prior art. Further, the wide-band response also has a steeper cutoff slope than the narrow
band, which is of beneficial in limiting noise in the system.
Referring to Figure 4, there is shown a second embodiment of a transimpedance amplifier 300 according to the present invention, comprising a current source 302, first and second 25 operational amplifiers 304 and 305, respectively, each having an inverting input, 306 and 307, and a non-inverting input 308 and 309, respectively, and an output 310 and 311.The output 311 of the second operational amplifier 305 having a resistor R1 connected in series with a resistor R3 which in turn is connected in series to a capacitor C 1. The transimpedance amplifier 300 further comprising a sampling network 312, a feedback
network 314 having a resistor R2 connected in parallel with a capacitor C2 and a summing node 317.
The current source 302 is connected to the inverting input 306 of the first operational 5 amplifier 304. The non- inverting input 308 of the operational amplifier 304 is connected to earth 316. The output 310 ofthe first operational amplifier 304 is connected to the non-inverting input 309 of the second operational amplifier 305. The inverting input 307 of the second operational amplifier 305 is connected to a first terminal of a resistor R5.
The second terminal of resistor R5 is connected to earth. A resistor R6 is connected lo between the output 311 and the inverting input 307 of the second operational amplifier305. A first terminal of the resistor R1 is connected to the output 311 and a second terminal is connected to a tap 318. A first terminal of the resistor R3 is also connected to the tap 318. The second terminal of the resistor R3 is connected to a first terminal of capacitor C1. The second terminal of the capacitor C1 is connected to earth 5 316. Connected to the tap 318 is the sampling network 312. The feedback network 314 leads from the sampling network 312 to the inverting input 306 of the first operational amplifier 304, that is between the output 311 and the input 306. The point at which the feedback network 314 is connected to the input 306 is the summing node 317 where the output signal, sampled by the sampling network 312, is summed with the input signal, 20 having been fed back via the feedback network 314. C3 represents the capacitance of the signal source, any connecting conductors and of the first operational amplifier.
The introduction of the second operational amplifier 305 increases the open loop gain,
which improves the gain accuracy of the transimpedance amplifier 300.
It will be appreciated that further stages of operational amplifiers may be added for other circuit configurations.
Referring to Figure 5, there is shown a third embodiment of a transimpedance amplifier 30 400 according to the present invention, comprising a current source 402, first and second operational amplifiers 404 and 405, respectively, each having an inverting input, 406 and
407, a non-inverting input 408 and 409, and an output 410 and 411, respectively. The output 411 of the second operational amplifier 405 having a resistor R1 connected in series with a resistor R3 which, in turn, is connected in series to a capacitor C 1. The transimpedance amplifier 400 further comprising a sampling network 412, a feedback 5 network 414 having a resistor R2 connected in parallel with a capacitor C2 and a summing node 417.
The current source 402 is connected to the non-inverting input 408 of the first operational amplifier 404. The inverting input 406 of the first operational amplifier 404 is connected lo to earth 416. The output 410 of the first operational amplifier 404 is connected to the inverting input 407 of the second operational amplifier 405. Between the output 410, of the first operational amplifier 404, and the inverting input 407, of the second operational amplifier 405, a capacitor C4 is connected in series with a resistor R4, across which a resistor R5 is connected in parallel. A resistor R6 is connected between the output 411 S and the inverting input 407, forming a feedback loop to the second operational amplifier 405. The non-inverting input 409, of the second operational amplifier 405, is connected to earth 416.
A first terminal of the resistor R1 is connected to the output 411 and a second terminal is 20 connected to a tap 418. A first terminal of the resistor R3 is also connected to the tap 418.
The second terminal of the resistor R3 is connected to a first terminal of capacitor C 1.
The second terminal of the capacitor C1 is connected to earth 416. Connected to the tap 418 is the sampling network 412. The feedback network 414 leads from the sampling network 412 to the non-inverting input 408 of the first operational amplifier 404, that is, 25 between the output 411 and the input 408. The point at which the feedback network 414 is connected to the input 408 is the summing node 417 where the output signal sampled by the sampling network 412, and having been fed back via the feedback network 414, is summed with the input signal. C3 represents the capacitance of the signal source, any connecting conductors and of the first operational amplifier.
The configuration of the second operational amplifier 405 provides a differentiating function determined by the capacitor C4 and the resistor R6. The differentiating function further extends the bandwidth and reduces the response time. It will be noted that the first operational amplifier 404 is disposed in a non-inverting configuration. However, the 5 signal at the output 411, of the transimpedance amplifier 400, is inverting because of the inverting configuration of the second operational amplifier 405, thereby maintaining negative feedback. The components R4, RS, R6 and C4 are chosen to enhance the gain in the frequency region above the typical cutoff of the transimpedance amplifier according to the closest known prior art. The resistors R5 and R6 set the zero frequency gain of the
lo second operational amplifier 405. The resistor R4 is operable as a damping control on the response of the differentiator feedback around the second operational amplifier 405. It is preferable that the configuration of the second operational amplifier 405 provides a wider bandwidth than the first operational amplifier 404, thereby reducing the influence of the second operational amplifier 405 on the overall bandwidth of the transimpedance is amplifier 400. The configuration of the first operational amplifier 404 is determined by the input bias current of the circuit, which must be significantly less than the currents to be measured.
The transimpedance amplifier 400 provides for measurement of wide-band, intermediate 20 band and narrow band output signals, which, if desired, may be measured simultaneously.
The wideband output signal is measured at the output 411 of the second operational amplifier 405. The intermediate band output signal is measured at the output 410 of the first operational amplifier 404. The narrowband output signal is measured at the junction of the resistor R3 and the capacitor C1.
2s Figure 6 shows the frequency response for the wide-band, intermediate band and narrow band outputs of the transimpedance amplifier 400 and that of the transimpedance amplifier l DO, according to the closest known prior art. It can be observed that the
frequency response of the output of the transimpedance amplifier 100, according to the 30 closest known prior art, is substantially that of the narrow-band response of the output of
the transimpedance amplifier 400. Accordingly, the improvement of bandwidth which the
transimpedance amplifier 400, according to the present invention, provides over the closest known prior art can be observed from the frequency responses of the intermediate
band and wideband outputs.
5 The wideband, intermediate band and narrowband responses of the transimpedance amplifier 400 and the response of the transimpedance amplifier 100, according to the closest known prior art, to an input current step function are shown in Figure 7. It can be
observed that the transimpedance amplifier 400, according to the present invention, provides significant improvement in rise-time relative to the closest known prior art. This
lo is further shown in the graph of Figure 8, in which the scale of the time axis has been increased to illustrate the significant improvements in rise-time, which the present invention provides.
The configuration of the transimpedance amplifiers 200, 300 and 400 also provides 5 significantly improved performance in relation to noise sensitivity relative to the closest known prior art.
Mathematically, for the first embodiment of the transimpedance amplifier 200, of Figure 2, according to the present invention, wherein R3<<R1 and, since the parallel impedance 20 Z2 of R2 and C2 is significantly larger than the effective source resistance at node X, it is assumed that v2 may be determined as a fraction of lo ignoring the loading effect of C2R2. Thus: v = lC,; = v (sC,R3 +1) (A)
C3 represents the capacity of the source connection and the amplifier input capacity.
Since the current source is here assumed ideal its source resistance is infinite as is the input resistance of the amplifier.
5 Thus: i = ii +i2 = 2 +VI5C3 with Z2 = (B) It is also necessary to allow for the open-loop frequency response A of the 0 amplifier. Commonly used electrometer amplifiers have a single-pole response with the corner frequency determined by a time constant T i.e. fC = 1/(2 T). Thus for a zero frequency gain AO: A= Ao (C) l+sT and the relation between vO and v/ is: -v v = -v, A or v1 = 0 (D) 20 Then from Eqn. A: I -Vo VosC3 Fo(l+sCR3) (l+sC2R2) AZ2 A (1 + sC, (R. + R3))R2 (E) = -V 1 + sC3 + (+ ' 3)] since C2R2 = C,(R, +R3)
The transimpedance transfer furction is: I (1 + sC2 R2) + SC3 + (1 + sC R3) AR2 A R2 (F)
- AR2 (l+sC2R2)+sC3R2 +A(l+sCIR3) 5 Since we are concerned with the response at high frequency well abovefc,we can write approximately: A= AT since sT>>1 (G) o Then: H(s)= Y = AoR2/ST SC2 R2 + I + SC3 R2 + O ( 3)
-AoR s2TR2(C2 +C3)+sT+Ao(l+sc'R3) (H) -Ao R2 s2TR2Cp+s(T+AOc,R3)+ Ao where Cp=C2+C3 5 i.e. there are two poles given by: 5 = - (T + Ao C R3) _ [(T + Ao Ci R3) - 4TCpR2 Ao] 2 2TCp R2 (I) Common electrometer amplifiers have a corner frequency 5Hz so T= 0.03, and 20 with typical values of Ao = 106, C/ =lOOnF, R3=lOOQ then AoCR3 10 and we may
neglect T relative to this. For critical damping of the second order system the poles must be equal (and real) i.e. the condition required is: (T + AD Cal R3)2 = 4TCp R2 AD or Ao2 C,2 R32 = 4TCp R2 AD neglecting T t4TR C] 2 s It will be appreciated that the output configuration according to the present invention is applicable to other circuit configurations and applications, thereby providing extensions of bandwidth and reduction in the response time of the outputs thereof.
as
Claims (1)
1. A transimpedance amplifier comprising a current source, having a capacitance C3, connected to signal amplification means having an output, said output comprising a 5 resistor R1 connected to a capacitor C 1 in series between said output and earth, a sampling network, connected to the output via a tap connected between the resistor R1 and the capacitor C1, operable to sample an output signal, a feedback network, comprising a resistor R2 connected in parallel to a capacitor C2 between the sampling network and a summing node, said summing node disposed at an input of the lo amplification means and operable to sum the sampled signal with an input signal from the current source, characterized in that the output further comprises resistor R3 connected in series between the tap and the capacitor C 1.
5 2. A transimpedance amplifier as claimed in Claim 1, wherein the amplification means
comprises an operational amplifier, having an inverting input, to which the current source is connected and a non-inverting input connected to earth.
3. A transimpedance amplifier as claimed in Claim 1 or Claim 2, adapted to separatley 20 measure a wide-band output signal and a narrow-band output signal.
4. A transimpedance amplifier as claimed in Claim 3, adapted to separately measure the wide-band output signal and the narrow-band output signal simultaneously.
25 5. A transimpedance amplifier as claimed in Claim 1, wherein the amplification means comprises a plurality of operational amplifiers.
6. A transimpedance amplifier as claimed in Claim 5, wherein at least one operational amplifier is connected in an inverting configuration and at least one operational 30 amplifier is connected in a non-inverting configuration.
7. A kansimpedance amplifier as claimed in Claim 5 or Claim 6, wherein at least one operational amplifier is adapted to operate as a differentiator.
8. A transimpedance amplifier as claimed in any of Claims 5 to 7, wherein the 5 amplification means comprises first and second operational amplifiers, the first operational amplifier having a non-inverting input, to which the current source is connected, an inverting input connected to earth, and an output connected to the second operational amplifier, the second operational amplifier having an inverting input, to which is connected the output of the first operational amplifier, a non o inverting input, connected to earth, and an output providing the output of the transimpedance amplifier, and between the output of the first operational amplifier and the inverting input of the second operational amplifier, a capacitor C4 is connected in series with a resistor R4, across which a resistor R5 is connected in parallel, and a resistor R6 connected in parallel with the second operational amplifier between the inverting input and the output thereof.
9. A transimpedance amplifier as claimed in Claims 5 to 8, wherein the second operational amplifier is adapted to operate at a wider bandwidth than the first operational amplifier.
10. A tansimpedance amplifier as claimed in claims 5 to 9, wherein a wideband output signal, an intermediate-band output signal and a narrow-band output signal are available for measurement.
25 11. A transimpedance amplifier as claimed in Claim 10, wherein at least two of the wide band output, intermediate band output and the narrow band output are measurable simultaneously. 12. A transimpedance amplifier as claimed in any of the preceding claims, wherein, for 30 the condition of (Rl+R3) x Cl = R2 x C2, the value of R3 is substantially
R3 = [ 2 ( 2 2 3)] ohm wherein AD is the zero-frequency open-loop gain of the operational amplifier and T is the time constant associated with the 3dB corner frequency fc=l/(2T).
5 13. A transimpedance amplifier substantially as herein described with reference to, as illustrated in, Figures 2 to 8.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GB0218082A GB2393865B (en) | 2002-08-03 | 2002-08-03 | Wideband transimpedance amplifier |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GB0218082A GB2393865B (en) | 2002-08-03 | 2002-08-03 | Wideband transimpedance amplifier |
Publications (3)
| Publication Number | Publication Date |
|---|---|
| GB0218082D0 GB0218082D0 (en) | 2002-09-11 |
| GB2393865A true GB2393865A (en) | 2004-04-07 |
| GB2393865B GB2393865B (en) | 2005-10-12 |
Family
ID=9941719
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| GB0218082A Expired - Fee Related GB2393865B (en) | 2002-08-03 | 2002-08-03 | Wideband transimpedance amplifier |
Country Status (1)
| Country | Link |
|---|---|
| GB (1) | GB2393865B (en) |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB2424330A (en) * | 2005-03-17 | 2006-09-20 | Scott Hamilton | A transimpedance amplifier with a shielded feedback resistor |
| US9431976B2 (en) | 2014-06-05 | 2016-08-30 | Thermo Fisher Scientific (Bremen) Gmbh | Transimpedance amplifier |
Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4069459A (en) * | 1976-08-23 | 1978-01-17 | Santa Barbara Research Center | Feedback capacitor divider |
| EP0098662A1 (en) * | 1982-07-09 | 1984-01-18 | Motorola, Inc. | Infra-red receiver front end |
| US4855687A (en) * | 1988-02-29 | 1989-08-08 | Micro Video, Inc | Transimpedance amplifier with noise reduction and bandwidth compensation |
| US6611174B1 (en) * | 2002-05-10 | 2003-08-26 | Adam J. Sherman | Self-compensated transimpedance amplifier |
-
2002
- 2002-08-03 GB GB0218082A patent/GB2393865B/en not_active Expired - Fee Related
Patent Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4069459A (en) * | 1976-08-23 | 1978-01-17 | Santa Barbara Research Center | Feedback capacitor divider |
| EP0098662A1 (en) * | 1982-07-09 | 1984-01-18 | Motorola, Inc. | Infra-red receiver front end |
| US4855687A (en) * | 1988-02-29 | 1989-08-08 | Micro Video, Inc | Transimpedance amplifier with noise reduction and bandwidth compensation |
| US6611174B1 (en) * | 2002-05-10 | 2003-08-26 | Adam J. Sherman | Self-compensated transimpedance amplifier |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB2424330A (en) * | 2005-03-17 | 2006-09-20 | Scott Hamilton | A transimpedance amplifier with a shielded feedback resistor |
| US9431976B2 (en) | 2014-06-05 | 2016-08-30 | Thermo Fisher Scientific (Bremen) Gmbh | Transimpedance amplifier |
Also Published As
| Publication number | Publication date |
|---|---|
| GB2393865B (en) | 2005-10-12 |
| GB0218082D0 (en) | 2002-09-11 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| US5754056A (en) | Charge detector with long integration time | |
| US4663586A (en) | Device with automatic compensation of an ac attenuator | |
| CN102545793A (en) | Weak current amplifier with pA (picoamperes)-mu A(microamperes) range | |
| Cochems et al. | In-circuit-measurement of parasitic elements in high gain high bandwidth low noise transimpedance amplifiers | |
| US5034698A (en) | Dual-path wideband and precision data acquisition system | |
| US4885528A (en) | Apparatus which uses a simulated inductor in the measurement of an electrical parameter of a device under test | |
| US3971984A (en) | Wide-range logarithmic responding translation circuit | |
| O'haver et al. | A versatile, solid state, constant bandwidth recording nanoammeter | |
| Štubian et al. | Fast low-noise transimpedance amplifier for scanning tunneling microscopy and beyond | |
| GB2393865A (en) | A transimpedance amplifier for detecting very small chromatograph or mass spectrometer currents with a greater bandwidth | |
| Achtenberg et al. | Transformer-based low frequency noise measurement system for the investigation of infrared detectors’ noise | |
| Petersons | A self-balancing high-voltage capacitance bridge | |
| US3321628A (en) | Baseline stabilization system for ionization detector | |
| EP0366453A2 (en) | Variable gain current-to-voltage amplifier with gain independent test mode operation | |
| US4538066A (en) | Modulated voltage metastable ionization detector | |
| US4780668A (en) | Method and an electronic circuit for measuring very weak signals at high voltage | |
| Conrad et al. | A recommended standard resistor-noise test system | |
| CN112904111B (en) | Ion signal detection circuit | |
| Scandurra et al. | A simplified topology for the design of low noise voltage amplifiers for low frequency noise measurements | |
| CN113030689B (en) | Test circuit, test method and test device of operational amplifier | |
| US4529936A (en) | Supply circuit including a pair of amplifying channels for an eddy current probe with two windings | |
| Wojciechowski et al. | The development of measuring setup for electrical noise research in two-dimensional semiconductor structures for quantum Hall resistance standard | |
| Kolobov et al. | A High-Dynamic-Range Counting− Analog Detector System Based on a Secondary Electron Multiplier | |
| GB2210744A (en) | D C offset voltage nulling circuit | |
| Re et al. | Instrumentation for high accuracy noise characterisation of front-end devices in detector applications |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| COOA | Change in applicant's name or ownership of the application | ||
| PCNP | Patent ceased through non-payment of renewal fee |
Effective date: 20160803 |