FR3014260A1 - METHOD AND SYSTEM FOR CONTROLLING A BIDIRECTIONAL CHARGER OF A MOTOR VEHICLE BATTERY. - Google Patents

Abstract

A method of controlling a bidirectional charger of a motor vehicle battery comprising a rectifier stage connected at the input to a power supply network and at the output of the conversion stage (2) and at a capacitor, the conversion stage (2 ) being connected to the battery, the conversion stage (2) comprising a first set of transistors (T1, T2, T3, T4) connected in series output to a resonant circuit (L1, C1) and to the transformer (6) , the transformer (6) being connected to a second set of transistors (T5, T6, T7, T8) connected to the battery. The method comprises the following steps: controlling the first set (T1, T2, T3, T4) at the natural frequency of the resonant circuit, if the battery voltage is lower than a threshold voltage, controlling the transformer (6) of so that a first transformation ratio is achieved, otherwise the transformer (6) is controlled so that a second transformation ratio, greater than the first ratio, is achieved.

Description

Method and system for controlling a bidirectional charger of a motor vehicle battery

The invention relates to the technical field control of the charge of motor vehicle batteries. As part of the development of low-cost electric vehicles, the charging system has, like the electric powertrain (GMPe) or the high-voltage battery (HT), a significant cost item that should be reduced. The limitation of the charging power represents a first way to reduce the cost of the charging system. It is thus appropriate to associate the electric vehicle a recharging system said "slow", that is to say that takes its energy on the single-phase network with a power less than or equal to 7kW. Such a charger typically absorbs 10A, 16A or 32A on the single-phase network and is compatible with a household outlet. Furthermore, the chargers of electric vehicles are currently non-reversible, that is to say that the flow of energy flows only from the electrical network to the battery. In the future, it will be interesting to have two-way chargers, that is to say also able to circulate energy from the battery to the power grid. Such chargers make it possible, for example, to smooth the load curve of the electrical network by using the energy storage represented by the battery of the connected vehicle. The battery is thus charged during consumption hollows during which the infrastructure and the means of production are underutilized, and provides energy during peaks to avoid the use of little used and expensive means of production and than overloading the power grid. In this case, the massive deployment of the electric vehicle does not imply additional investments of production or infrastructure, but on the other hand allows to smooth peaks of expensive consumptions to provide (with a generally very carbonaceous energy). Two-way chargers facilitate the deployment of renewable energies by optimally adapting the consumption of energy produced to the production hazards specific to green energy and benefit from a better performance in terms of carbon dioxide (CO2) emissions from the environment. electric vehicle while facilitating the introduction of renewable energy production. Bi-directional chargers also enable the creation of new activities related to the control and optimization of energy consumption without degradation of service. Today, the charger is required to perform the energy conversion between the network and the battery. This conversion requires the charger to adapt to wide input voltage ranges (90V to 250V AC) and battery voltage (250V to 400V DC).

FIG. 1 illustrates the main elements of a charger which comprise an input rectifier stage 1 connected to the network 3 and providing the power factor correction function, denoted PFC (acronym for "Power Factor Corrector"), and a stage 2 DC / DC conversion, allowing the galvanic isolation of the battery 5. An electromagnetic interference filter noted EMI (acronym for "Electromagnetic Interferences") can be arranged between the network 3 and the floor input rectifier, while an output filter can be arranged between the DC-DC conversion stage and the battery 5.

The difficulty lies in being able to adapt to all of the voltage levels mentioned above without sacrificing either the efficiency or the cost of the conversion stage while keeping the possibility of carrying out a bidirectional load. With regard to the bi-directional charge of electric vehicles, the state of the art today relies essentially on non-isolated topologies, that is to say that there is no galvanic isolation between the network and battery. Regarding the design of insulated chargers, in general, the first stage consists of a boost topology for performing the PFC function while being voltage booster. In order to limit switching losses and frequency increase (synonymous with cost reduction, volume, weight reduction of passive elements), it is strongly recommended that the second stage uses a resonant circuit. Between these two stages, an electrolytic capacitance is positioned in order to smooth the current in constant current and to manage a constant power flow and no longer variable in sine. Since the first stage has a voltage booster function, the DC voltage (DC bus) between the two stages is generally higher than the battery voltage. This implies that the DC / DC stage must have a voltage gain of less than unity. With a resonant isolated circuit, the total voltage gain is obtained by the product between the transformer transformation factor and the gain of the resonant circuit. The transformation factor is fixed and dependent on the design. It can not help adjust the gain during a vehicle load. However, when a vehicle is recharged, the battery voltage varies according to its state of charge SoC (state of charge), this requires in real time that the DC / DC stage adapts its gain for allow to convert the DC bus voltage to the battery voltage. This voltage adaptation can be made only by the operation of the resonant circuit.

Generally, the use of the series-parallel resonant topology LLC (acronym referring to the use of a circuit comprising the combination of two inductances each denoted L and a capacity denoted C) is applied in the DC converter. DC. The variation of the switching frequency during charging makes it possible to vary the voltage gain of the resonant circuit. The frequency used is always greater than the natural frequency fo of the resonant circuit, which then operates in a hyper-resonance regime. The use of a resonant circuit in hyper-frequency makes it possible to have ZVS (Zero Voltage Switching) zero switching switching soft switching when closing the transistors. However, this solution has two important problems. A first problem lies in the fact that the control dynamics of the DC / DC conversion resonating in hyper-frequency depends on the flow of power passing through it. To solve this problem, an electrolytic capacitance (voluminous, expensive and unreliable) must be put in place between the PFC and the DC / DC in order to smooth the flow of power. Such a capacity is noted 4 in FIG.

A second problem lies in the fact that the hyper-frequency operation makes it possible to limit the losses during the closing of the transistors but not when they are opened. If the switching frequency of the switches is equal to the natural frequency fo, it is possible to combine the ZVS closure with zero opening loss limitation in zero current switching ZCS (acronym for "Zero Current Switching"). From the state of the art, we know the following documents. US20110273130 discloses a bi-directional insulated charger which is however not used for vehicle charging applications. However, like all chargers of this type, it includes a current rectification stage performed with an electrolytic capacitance which has a significant cost. Patent application number FR1255070 discloses a resonant topology for charging a battery without using power factor correction. However, the control of such a topology is variable frequency, inducing operation in ZVS. Such an operation generates non-negligible switching losses limiting the operating frequency. This limitation indirectly involves additional costs. Document US 577-116-5A discloses a two-stage converter of the PFC boost and DC continuous conversion (DC / DC) type. However, this converter uses an electrolytic capacitance between the two stages so as to smooth the power flow in the converter. In addition, the DC / DC stage does not make it possible to combine the ZVS and ZCS switches at the transistors located before the resonant circuit. An object of the invention is a method of controlling a bidirectional charger of a motor vehicle battery, the charger comprising a rectifier stage connected at the input to a power supply network and at output to a continuous conversion stage. continuous and at a capacity, the DC-DC conversion stage being connected to the battery. The DC-DC conversion stage comprises a first set of transistors connected at input to the rectifier stage and in series output to a resonant circuit and to a transformer, the transformer being connected to a second set of transistors, the second set being otherwise connected to the battery. The method comprises the following steps: controlling the switching of the first set of transistors to the natural frequency of the resonant circuit, controlling the second set of transistors in an on state, determining whether the battery voltage is greater than a threshold voltage if this is the case, the transformer is controlled so that a first transformation ratio is achieved, if this is not the case, the transformer is controlled so that a second transformation ratio is achieved, the second transformation ratio being greater than the first transformation ratio. The threshold voltage can be determined as the square root of the product of the maximum value and the minimum value of the battery voltage. A switch can be connected by a terminal between one end of a first primary winding of the transformer and one end of the transformer. a second primary winding of the transformer, a switch being connected by a terminal to the other end of the second primary winding, the switches being connected by their other terminal to the collector of the fourth transistor and to the emitter of the third transistor. To control the transformer so that the first transformation ratio is achieved, the switches can be controlled so that only the first primary winding of the transformer is connected to the first set of transistors, and to control the transformer so that the second gear In order to achieve the transformation, the switches can be controlled so that the first primary winding and the second primary winding of the transformer are connected to the first set of transistors. The first transformation ratio can be determined as the ratio of the sum of the peak value of the grid voltage and an offset voltage by the threshold voltage.

The second transformation ratio can be determined as the ratio of the sum of the maximum of the peak value of the mains voltage and an offset voltage by the minimum of the battery voltage. The switches may be controlled by high frequency modulation so that the obtained transformation ratio is equal to the average of the first transformation ratio and the second transformation ratio weighted by their respective switching times. Another object of the invention is a control system of a bidirectional charger of a motor vehicle battery, the charger comprising a rectifier stage connected at input to a power supply network and at output to a continuous conversion stage -continuous and a capacity, the DC-DC conversion stage being connected to the battery. The DC-DC conversion stage comprises a first set of transistors connected at input to the rectifier stage and in series output to a resonant circuit and to a transformer, the transformer being connected to a second set of transistors, the second set being otherwise connected to the battery. The system comprises: a control means able to control the switching of the first set of transistors to the natural frequency of the resonant circuit, and able to control the second set of transistors in an on state, a means for comparing the battery voltage to a threshold voltage, able to control the transformer so that a second transformation ratio is achieved, if the battery voltage is greater than a threshold voltage and a first transformation ratio is achieved, if the battery voltage is not greater than a threshold voltage, the second transformation ratio being greater than the first transformation ratio. The system may include means for determining the threshold voltage as a function of the maximum value and the minimum value of the battery voltage.

A first switch may be connected by a terminal between one end of a first primary winding of the transformer and one end of a second primary winding of the transformer, a second switch being connected by a terminal to the other end of the second primary winding, the first switch and the second switch being connected by their other terminal to the collector of the fourth transistor and to the emitter of the third transistor, the first switch and the second switch being able to modify the transformation ratio of the transformer as a function of the number of connected primary windings.

The number of windings of the first primary winding of the transformer divided by the number of windings of the secondary winding of the transformer may be equal to the first transformation ratio, itself equal to the ratio of the sum of the peak value of the transformer. mains voltage and offset voltage by the threshold voltage. The sum of the number of windings of the first primary winding of the transformer and the second primary winding of the transformer divided by the number of windings of the secondary winding of the transformer may be equal to the second ratio of transformation, itself equal to the ratio the sum of the maximum of the peak value of the mains voltage and an offset voltage by the minimum of the battery voltage. The switches may be capable of being controlled by high frequency modulation so that the obtained transformation ratio is equal to the average of the first transformation ratio and the second transformation ratio weighted by their respective switching times. The first switch and / or the second switch can be a triac or a relay. The first switch and / or the second switch may be a reversible switch comprising two terminals, a first terminal being connected to the cathode of a ninth diode and to the anode of a tenth diode, the anode of the ninth diode being connected to the anode of an eleventh diode and to the source of a first MOSFET transistor, the cathode of the eleventh diode and the drain of the first MOSFET transistor being connected to the second terminal, the cathode of the tenth diode being connected to the cathode of a twelfth diode and the drain of a second MOSFET transistor, the anode of the twelfth diode and the source of the first MOSFET transistor being connected to the second terminal. The method and the control system have the advantage of allowing, for the output capacity of the rectifier stage, to employ a film capacitor in place of an electrolytic capacitance. This improves the cost and reliability of the conversion stage. The method and the control system have the advantage of allowing the transistors to be switched to the natural frequency of the resonant circuit. This makes it possible to limit switching losses as much as possible at both opening and closing, which increases the efficiency of the converter and makes it possible to limit the costs associated with the installation of a consequent hydraulic cooling system. . Other objects, features and advantages of the invention will become apparent on reading the following description, given solely by way of nonlimiting example and with reference to the appended drawings in which: FIG. 1 illustrates the main elements of a bidirectional charger, - figure 2 illustrates the main elements of a continuous DC conversion stage, - figure 3 illustrates the main elements of a DC continuous conversion stage according to the invention, - figure 4 illustrates the profile of the DC bus voltage as a function of the battery voltage Vbat, and - Figure 5 illustrates the main elements of a Mosfets reversible switch. Fig. 1 shows the bidirectional charger comprising an input rectifier stage 1 and an output DC-DC conversion stage 2. The input rectifier stage 1 generates a DC bus voltage and ensures absorption of a sinusoidal current on the power supply network 3. The input rectifier stage 1 is connected in parallel with a capacitor 4 and a DC continuous output conversion stage 2, the conversion stage being connected at the output to a battery 5. Unlike the state of the prior art, the input rectifier stage 1 of the bidirectional charger according to FIG. The invention is also able to regulate the charge of the battery through the variation of the DC bus voltage (325V-400V). In FIG. 2, it can be seen that the output DC-DC conversion stage 2 is series resonant and galvanically isolated via a transformer 6, the assembly operating at a constant switching frequency.

A first input of the conversion stage is connected to the collector of a first transistor T1 and to the collector of a third transistor T3, a second input of the conversion stage being connected to the emitter of a second transistor T2 and to the emitter of a fourth transistor T4. The emitter of the first transistor T1 and the collector of the second transistor T2 are connected to a first intermediate point, the emitter of the third transistor T3 and the collector of the fourth transistor T4 being connected to a second intermediate point.

The first intermediate point is connected in series with a first capacitor C1, a first inductor L1, at a first end of the first winding of a transformer 6, the second end of the first winding of the transformer 6 being connected to the second intermediate point.

One end of the second winding of the transformer 6 is connected to the emitter of a fifth transistor T5 and to the collector of a sixth transistor T6, the collector of the fifth transistor T5 being connected to the collector of a seventh transistor T7 and to a first output of the conversion stage. The emitter of the sixth transistor T6 is connected to the emitter of an eighth transistor T8 and to a second output of the conversion stage. The other end of the second winding is connected to the emitter of the seventh transistor T7 and to the collector of the eighth transistor T8. A first diode D1 is connected by its anode to the emitter of the first transistor Ti and by its cathode to the collector of the first transistor Ti. A second diode D2 is connected by its anode to the emitter of the second transistor T2 and by its cathode to the collector of the second transistor T2.

A third diode D3 is connected by its anode to the emitter of the third transistor T3 and by its cathode to the collector of the third transistor T3. A fourth diode D4 is connected by its anode to the emitter of the fourth transistor T4 and its cathode to the collector of the fourth transistor T4. A fifth diode D5 is connected by its anode to the emitter of the fifth transistor T5 and its cathode to the collector of the fifth transistor T5.

A sixth diode D6 is connected by its anode to the emitter of the sixth transistor T6 and its cathode to the collector of the sixth transistor T6. A seventh diode D7 is connected by its anode to the emitter of the seventh transistor T7 and by its cathode to the collector of the seventh transistor T7. An eighth diode D8 is connected by its anode to the emitter of the eighth transistor T8 and by its cathode to the collector of the eighth transistor T8.

The gates of the transistors receive control signals, in particular pulse width modulation transmitted at a constant frequency. The second set of transistors (T5, T6, T7, T8) allows bidirectional operation. During the discharge of the battery to the power supply network, the behavior of the first set of transistors (T1, T2, T3, T4) and second set of transistors (T5, T6, T7, T8) are exchanged with respect to operation when charging the battery. The control method makes it possible to maintain a fixed voltage gain on the resonant circuit by switching the first transistor T1, the second transistor T2, the third transistor T3 and the fourth transistor T4 to the natural frequency of the resonant circuit. To achieve the voltage adaptation required for charging the battery, the voltage gain is adjusted between the DC bus (Vred) and the battery by a transformation ratio of the variable transformer. The voltage Vred is measured at the output of the rectifier stage 1, at the terminals of the capacitor 4. According to one particular embodiment illustrated in FIG. 3, the DC-DC conversion stage 2 differs from that illustrated by FIG. 2 in that it comprises a transformer with two primary windings and switches I1 and 12 for switching these windings depending on the operating point. Referring to the description of FIG. 2 and FIG. 3, it can be seen that the impedance L1 is connected to one end of the first primary winding of the transformer 6. The other end of the first primary winding of the transformer 6 is connected to one end of the second primary winding of the transformer 6 and to a terminal of a first switch 11. The other end of the second primary winding of the transformer 6 is connected to a terminal of a second switch 12. The other terminals of the first switch I1 and the second switch 12 are connected together and at the second intermediate point.

The structure of the circuit connecting the transistors, the diodes remain identical to those described in connection with Figure 2. The switches I1 and 12 may be triacs or relays. The switches I1 and 12 operate in an open loop while the transistors T1 to T4 operate at a switching frequency set at the value of the resonant frequency of the circuit L-C. It is recalled that at its resonance frequency, a series LC resonant circuit has a voltage gain equal to 1. Since the DC-DC conversion stage 2 operates at resonance, the entire gain of this stage is based on the gain brought by the transformation ratio. Depending on the state of charge of the battery and therefore its voltage Vbat, the DC bus voltage, noted Vred, must be adapted to allow conversion. FIG. 4 shows the profile of the voltage Vred as a function of the battery voltage Vbat. The different phases of operation presented in this figure are associated with the different transformation ratios used and will be discussed below. The assumption is made that at a threshold voltage Vth, the relay is switched. The transformer then changes from a ratio k1 to a ratio k2 greater than kl. More precisely, if the battery voltage Vbat is greater than the threshold voltage Vth, the ratio kl is used. If the battery voltage Vbat is lower than the threshold voltage Vth, the ratio k2 is used. In order for this type of operation to be implemented, it is necessary to check the minimum value Vred min and the maximum value Vred max of the voltage at the terminals of the capacitor 4.

At any time, it is necessary that the voltage Vred is greater than the maximum voltage of the network Vres_max since this voltage is generated by the rectifier stage 1 which is a voltage booster stage, said boost stage. Indeed, a voltage booster stage may have difficulty in obtaining an output voltage equal to the input voltage due to the imperfect nature of the components used. To account for this, it is believed that a minimum offset must be maintained between the output voltage and the input voltage so that the step-up stage can operate under acceptable conditions. The reduction of electrical losses involves reducing the leakage current of the transistors. For this, it is important to consider components whose breakdown voltage is as close as possible to the operating point of the charger, especially at the level of the Vred continuous bus voltage. For example, a limit of this voltage can be set at 600 V for the application presented hereinafter. Those skilled in the art may, however, adapt the following examples for a different limit. With a limit of 600V, it is necessary that at any time, the voltage Vred is lower than 550V. It is also important to minimize the maximum voltage Vred in order to limit the losses in the inverter. The following control method makes it possible to take these constraints into account and thus allow the use of this type of transformer for bidirectional charger applications for electric or hybrid vehicles. Throughout the load, the following equations govern the system: Vred = k.Vbat (Eq. 1) With Vbat = the battery voltage k = the transformation ratio used Vred min = Vres. = Vres_max (Eq. 2) With Vred min = the minimum value of the voltage Vred Vres max = the maximum value of the mains voltage Vres For reasons of control stability, the minimum voltage of the DC bus, denoted Vred min, is greater than the voltage Vres max of an offset value AV, the offset value AV being equal to 40V. Indeed, the DC bus voltage Vred is obtained at the output of the rectifier stage 1. However, this stage can produce a voltage greater than the voltage supplying it, in this case the voltage of the network Vres. In addition, a gain equal to unity is particularly difficult to obtain and maintain. For these reasons, an offset value is considered in order to allow easy operation of the rectifier stage 1. The transformation ratio k2 is a parameter which is determined by the operating conditions of the charger. Indeed, the charger must be able to charge a completely discharged battery (250V) on a network of 250VAC. This imposes the following equation: Vres max + AV = k2.Vbat min (Eq.3) With Vbat min = the minimum value of the battery voltage Vbat Vres max = the maximum value of the voltage Vres of the supply network electric. The equation Eq. 3 can be reformulated to determine the transformation ratio k2. Vres + AV k2 = (Eq.4) Vbat min By choosing Vres max = 250V, Vbat min = 250V and AV = 40V, we obtain a transformation ratio k2 which takes the value 1.574.

To limit the voltage level Vres on the DC bus, the following two conditions are applied: kl.Vth = Vres + AV (Eq.4) k2.Vth = kl.Vbat max (Eq.5) With Vbat max = la maximum value of the battery voltage Vbat The equation (Eq.4) means that, at the transition point C of FIG. 4, an AV voltage offset is imposed at the minimum voltage Vred (determined by the equation Eq.2 ). The equation (Eq.5) indicates that it is desired to obtain a voltage Vred at point B of FIG. 4 equal to the value of Vred at point D in FIG. 4. This condition makes it possible to minimize the maximum value of Vred on the total range of voltages. The transformation ratio k1 is a parameter which is also determined by the boundary conditions. In practice, a network of up to a voltage of 250 V AC is used. The two equations governing the system are: kl .Vth = VresmIXW. D +40 (Eq.6) k2.Vth = kl.Vbat. (Eq.7) From these two equations, one can obtain the expression of the theoretical tension Vth, by the successive developments presented below. k2.Vth = Vresn '..' 1 + 40 .Vbat. Vth (Eq.8) Vth2 = Vres + 40 Vbat. k 2 (Eq.9) 16 Vresina ,, .- +40 vb (Eq.10) Vth 2 - t Vbat Vresi. 40. a max - mm Vth VVbat .Vbatm (Eq. 11) By choosing Vbat max = 400V, Vbat min = 250V, a threshold value Vth equal to 316V is obtained. Using the equations Eq. 11 and Eq. 6, we can then determine a value of the first transformation ratio kl equal to 1.25 In order to design the transformer, the number of windings n1 of the first primary winding, the number of windings n2 of the second primary winding and the number of windings n3 of the secondary winding must be determined, and this, in accordance with the transformation ratios kl and k2.

An example of winding numbers is n1 = 20, n2 = 5, and n3 = 16. We find again the value of the first transformation ratio k1 = n1 / n3 = 1.25 and the second transformation ratio k2 = (nl + n2 ) / n3 = 1.56. It thus appears that since the second transformation ratio k2 is dimensioned by the limit conditions of the continuous DC conversion stage, the maximum voltage at the output of the rectifier stage 1 without adaptation of the transformation ratio would be greater than the limit of 600V. With the adaptation of the transformation ratio, it is therefore possible to use the continuous continuous conversion stage 2 pure resonance while retaining 600V components that generate less losses. With the adaptation of the transformation ratio, the DC bus voltage (Vred in the figure) is 400V instead of 600V for the conventional solution.

We note that the design is based on the case of maximum network voltage, and the efficiency will be optimized on the maximum network voltage (which is the most common case in France). In the previous example, the transformation ratio is modified at the end of a switching period of the transistors (T1, T2, T3, T4), making it possible to go from k1 = 20/8 to k2 = 40/8. By performing this switching of the switches I1 and 12 at the end of each switching period of the transistors (T1, T2, T3, T4), or at the end of a multiple of switching periods, an equal average transformation ratio is obtained. at k = 30/8. This control principle can be extended by modifying the ratio between the duration during which the switches (11,12) make it possible to obtain a ratio k1 and the duration during which the switches (11,12) make it possible to obtain a ratio k2. The transformation ratio obtained can be determined by the following equation: k = (tl * t2) * (kl I tl + k2 t2) (Eq.12) With tl: the duration during which the switches (11, 12) are switched so as to obtain the transformation ratio k1, and t2: the duration during which the switches (11, 12) are switched so as to obtain the transformation ratio k2. To achieve this, a high frequency modulation of the switches is used ( 11, 12) which is superimposed on the main modulation produced by the transistors (T1, T2, T3, T4). By varying the driving pattern of the switches (11,12) so as to vary the ratio of the times t1 and t2 of the equation 12, it is possible to obtain a continuously varying transformation ratio of the value k1 = 20/8 at the value k2 = 40/8. In addition, it is necessary to switch the switches (11,12) zero currents, all transistors (T1, T2, T3, T4) must be blocked. However, triacs are limited in frequency. Instead of each switch (11, 12), it is preferable to use a switch based on Mosfet transistors (metal oxide semiconductor field effect transistor) arranged in a structure that allows reversible behavior. In FIG. 5, it can be seen that such a switch comprises two terminals, a first terminal being connected to the cathode of a ninth diode D9 and to the anode of a tenth diode D10. The anode of the ninth diode D9 is connected to the anode of an eleventh diode D11 and to the source of a first MOSFET transistor referenced M1. The cathode of the eleventh diode D11 and the drain of the first MOSFET transistor M1 are connected to the second terminal. The cathode of the tenth diode D10 is connected to the cathode of a twelfth diode D12 and to the drain of a second transistor of MOSFET type referenced M2. The anode of the twelfth diode D12 and the source of the first MOSFET transistor M2 are connected to the second terminal.

Claims (14)

REVENDICATIONS1. A method of controlling a bidirectional charger of a battery (5) of a motor vehicle, the charger comprising a rectifier stage (1) connected at the input to a power supply network (3) and at output to a conversion stage ( 2) DC-DC and at a capacitance (4), the DC-DC conversion stage (2) being connected to the battery (5), the DC-DC conversion stage (2) comprising a first transistor set (T1, T2, T3, T4) connected in input to the rectifier stage (1) and in series output to a resonant circuit (L1, C1) and to a transformer (6), the transformer (6) being connected to a second set of transistors (T5, T6, T7, T8), the second set (T5, T6, T7, T8) being furthermore connected to the battery (5), characterized in that it comprises the following steps: the first set of transistors (T1, T2, T3, T4) is switched to the natural frequency of the resonant circuit, the second set of emble of transistors (T5, T6, T7, T8) in an on state, it is determined whether the battery voltage is greater than a threshold voltage. If this is the case, the transformer (6) is controlled so that a first transformation ratio is achieved, if this is not the case, the transformer (6) is controlled so that a second transformation ratio is performed, the second transformation ratio being greater than the first transformation ratio. 2. Control method according to the preceding claim, wherein the threshold voltage is determined as the square root of the product of the maximum value and the minimum value of the battery voltage. 3. Control method according to any one of the preceding claims, whereina switch (I1) is connected by a terminal between an end of a first primary winding of the transformer (6) and an end of a second primary winding of the transformer (6), a switch (I2) being connected by a terminal to the other end of the second primary winding, the switches (I1) and (I2) being connected by their other terminal to the collector of the fourth transistor (T4) and to the emitter of the third transistor (T3), for controlling the transformer (6) so that the first transformation ratio is achieved, the switches (I1) and (I2) are controlled, so that only the first primary winding of the transformer ( 6) is connected to the first set of transistors (T1, T2, T3, T4), and to control the transformer (6) so that the second transformation ratio is achieved, it controls the switches 11 and 12 of s that the first primary winding and the second primary winding of the transformer (6) are connected to the first set of transistors (T1, T2, T3, T4). 4. Method according to claim 3, wherein the first transformation ratio is determined as the ratio of the sum of the peak value of the mains voltage and an offset voltage by the threshold voltage. 5. A method according to any one of claims 3 or 4, wherein the second transformation ratio is determined as the ratio of the sum of the maximum of the peak value of the mains voltage and an offset voltage by the minimum of the battery voltage. 6. Control method according to any one of the preceding claims, wherein the switches (11, 12) are controlled by a high frequency modulation so that the conversion ratio obtained is equal to the average of the first transformation ratio and of the second transformation ratio weighted by their respective switching times. 7. A control system for a bidirectional charger of a battery (5) of a motor vehicle, the charger comprising a spreader (1) input connected to a power supply network (3) and output to a conversion stage (2) DC-DC and at a capacitance (4), the DC-DC conversion stage (2) being connected to the battery (5), the DC-DC conversion stage (2) comprising a first set of transistors (T1, T2, T3, T4) connected at input to the rectifier stage (1) and in series output to a resonant circuit (L1, C1) and to a transformer (6), the transformer (6) being connected to a second set of transistors (T5, T6, T7, T8), the second set (T5, T6, T7, T8) being furthermore connected to the battery (5), characterized in that it comprises: a means control device adapted to control the switching of the first set of transistors (T1, T2, T3, T4) at the natural frequency of the resonant circuit, and able to control the e second set of transistors (T5, T6, T7, T8) in an on state, a means for comparing the battery voltage to a threshold voltage, adapted to control the transformer (6) so that a second ratio of transformation is performed, if the battery voltage is greater than a threshold voltage and a first transformation ratio is achieved, if the battery voltage is not greater than a threshold voltage, the second transformation ratio being greater than at the first transformation report. 8. Control system according to claim 7, comprising means for determining the threshold voltage as a function of the maximum value and the minimum value of the battery voltage. 9. Control system according to any one of claims 7 or 8, wherein a first switch (I1) is connected by a terminal between one end of a first primary winding of the transformer (6) and one end of a second primary winding of the transformer (6), a second switch (12) being connected by a terminal to the other end of the second primary winding, the first switch (I1) and the second switch (12) being connected by their other terminal to the collector the fourth transistor (T4) and the emitter of the third transistor (T3), the first switch (I1) and the second switch (I2) being able to modify the transformation ratio of the transformer (6) as a function of the number of connected primary windings. A control system according to claim 9, wherein the number of windings of the first primary winding of the transformer (6) divided by the number of windings of the secondary winding of the transformer (6) is equal to the first transformation ratio. itself equal to the ratio of the sum of the peak value of the mains voltage and an offset voltage by the threshold voltage. The system of any one of claims 9 or 10, wherein the sum of the number of windings of the first primary winding of the transformer (6) and the second primary winding of the transformer (6) divided by the number of windings of the secondary winding of the transformer (6) is equal to the second transformation ratio, itself equal to the ratio of the sum of the maximum of the peak value of the grid voltage and a lag voltage by the minimum of the battery voltage. 12. Control system according to any one of claims 7 to 11, wherein the switches (I1, 12) are capable of being controlled by a high frequency modulation so that the conversion ratio obtained is equal to the average of the first transformation ratio and the second transformation ratio weighted by their respective switching times. 13. System according to any one of claims 7 to 12, wherein the first switch (I1) and / or the second switch (I2) is a triac or a relay. 14. System according to any one of claims 7 to 12, wherein the first switch (I1) and / or the second switch (I2) is a reversible switch comprising two terminals, a first terminal being connected to the cathode of a ninth diode (D9) and the anode of a tenth diode (D10), the anode of the ninth diode (D9) being connected to the anode of an eleventh diode (D11) and to the source of a first MOSFET transistor (M1), the cathode of the eleventh diode (D11) and the drain of the first MOSFET transistor (M1) being connected to the second terminal, the cathode of the tenth diode (D10) being connected to the cathode of a twelfth diode (D12) and the drain of a second MOSFET transistor (M2), the anode of the twelfth diode (D12) and the source of the first MOSFET transistor (M2) being connected to the second terminal.