EP4378061A1

EP4378061A1 - Converter circuit for generating an isolated dc voltage

Info

Publication number: EP4378061A1
Application number: EP22758480.2A
Authority: EP
Inventors: Michael Kokes
Original assignee: Hochschule Heilbronn
Current assignee: Hochschule Heilbronn
Priority date: 2021-07-30
Filing date: 2022-07-29
Publication date: 2024-06-05
Also published as: DE102021208278A1; WO2023006949A1

Abstract

The invention relates to a converter circuit (10) for generating an isolated DC voltage (u_out), having: a switchable input converter (30), a switchable resonant circuit converter (40), an m-phase transformer (50), and a rectifier bridge (60), wherein the input converter (30) is configured so as to be connected to an n-phase power network (20) and generate a pulsed output current (I_DC) from n input currents (i₁; i₂; i₃) at an input converter switch frequency f0; the resonant circuit converter (40) has m phases and is configured so as to convert the pulsed output current (I_DC) of the input converter (30) into a resonant circuit current (i_a; i_b; i_c) in each of the m phases at a resonant circuit converter switch frequency f₀/m; the m resonant circuit currents (i_a; i_b; i_c) are shifted relative to one another by the phase angle in each of the m phases; each of the m phases of the resonant circuit converter (40) has a resonant circuit capacitor (C_p) and forms a respective parallel resonant circuit together with a main inductor (L_h) of the m-th phase of the transformer (50); each resonant circuit capacitor (C_p) of the m-th phase of the resonant circuit converter (40) is configured so as to provide a magnetization current for the m-th phase of the transformer (50), and the transformer (50) is configured so as to generate m secondary-side output voltages (u_ab; u_bc; u_ca) in an isolated manner from the primary side of the transformer (50); the m phases of the transformer (50) are connected to the rectifier bridge (60) on the secondary side, and the rectifier bridge (60) is configured so as to rectify m secondary-side output voltages (u_ab; u_bc; u_ca) of the transformer (50) and generate an isolated DC voltage (u_out) at the output; and n equals a positive whole number, and m equals a positive whole number which is greater than 2.

Description

Power converter circuit for generating a potential-separated DC voltage

Description

The invention relates to a converter circuit and a converter circuit system for generating a potential-separated DC voltage and its use, as well as a method for generating a potential-separated DC voltage by a converter circuit.

Mobile electrical applications, especially in vehicle technology, require ever more powerful traction batteries and accumulators in order to have sufficient electrical energy supplied over a longer period of time. For charging such high-performance traction batteries with direct current, because of the high protection requirements for such charging stations, electrical isolation is required between the AC mains supply and the direct voltage to be applied to a traction battery. In the prior art, this electrical isolation occurs either on the network side by means of transformers or within a power converter by using a high-frequency clocked converter. These converters in the power converters usually have a single-phase design and are fed from an upstream DC link. For high charging capacities over 100 kW, charging stations with a line-side transformer or a power converter with a DC link and converter are very expensive. A major reason for this is the cost of the coiled goods and the capacitors for intermediate energy storage, which are required for such electrical circuits of the charging stations. In this case, suitable mains transformers cannot be manufactured automatically and, in addition, high production costs arise due to the high material costs for iron cores and loose goods. When converters are used in this power range, the switching frequencies to be used are comparatively low and therefore require manual and therefore cost-intensive wiring of the converter circuits. However, if the field of electromobility is to be used not only in the commercial environment but also in private use, powerful rapid charging devices must become significantly cheaper.

It is therefore an object of the present invention to provide a power converter circuit and a method for generating a potential-separated DC voltage which causes lower costs for the winding goods and energy storage devices to be used and enables largely automated production.

This problem is solved by the subjects of the independent claims. Advantageous embodiments are the subject matter of the dependent claims.

One aspect of solving the problem relates to a converter circuit for generating a potential-separated DC voltage, having: a switchable input converter, a switchable oscillating circuit converter, an m-phase transformer and a rectifier bridge; wherein the input converter is configured to be connected to an n-phase power grid and to generate a pulse-shaped output current from n input currents at an input converter switching frequency fo; wherein the tank converter has m phases and is configured to convert the pulsed output current of the input converter into a tank current in each of the m phases at a tank converter switching frequency fo/m; wherein the m resonant circuit currents in the m phases are each shifted in phase angle to one another; each of the m phases of the oscillating circuit converter having an oscillating circuit capacitor and forming a respective parallel oscillating circuit with a main inductance of the m-th phase of the transformer; wherein each tank circuit capacitor of the m-th phase of the tank circuit converter is configured to provide a magnetizing current for the m-th phase of the transformer and the transformer is configured to generate m secondary-side output voltages isolated from the primary side of the transformer; wherein the m phases of the transformer are connected to the rectifier bridge on the secondary side and the rectifier bridge is configured to rectify the m secondary-side output voltages of the transformer and to provide the isolated DC voltage at an output; and where n is a positive integer and where m is a positive integer greater than or equal to 2.

The power converter circuit offers the advantage that higher switching frequencies are possible compared to conventional power converters with a DC link and converter. This is accompanied by a reduction in the size of the windings used for input inductances and transformers. The use of large and therefore expensive capacitors, such as electrolytic capacitors for intermediate storage of energy in the DC voltage intermediate circuit, can be dispensed with entirely. The smaller coils and capacitors that can be used for the converter circuit can also be integrated more easily in an automated production process for the converter circuit and, in addition to the lower material costs, also make the production process cheaper.

The converter circuit does not require a DC voltage intermediate circuit, but has an input converter, an m-phase oscillating circuit converter, an m-phase transformer and a rectifier bridge, viewed from the input side towards the output, and the converter circuit is connected to an n-phase power supply system on the input side and can be connected and/or connected to an accumulator to be charged on the output side. In particular, the converter circuit has no intermediate circuit capacitor between an input converter output and an oscillating circuit converter input. In this case, the input converter output is preferably connected directly to the resonant circuit converter input via at least one electrical line, without a further electronic component being configured between the input converter output and the resonant circuit converter input.

The input converter and the m-phase oscillating circuit converter are each configured to be switchable. This means that the input converter and the m-phase resonant circuit converter each have at least one switching device which provides an input signal in a switched-on state for a switch-on period TEI _P at its output and which supplies the input signal in a switched-off state for a period TOFF make them unavailable at their exit. The switching devices of the input converters and the m-phase oscillating circuit converter can each be switched on and off by applying a control signal to a control input. The control signal can be designed to switch the switching device to a switched-on state when the control voltage is not equal to 0V and to switch the switching device to a switched-off state when the control voltage deviates from this. The control voltage can preferably be a positive voltage for switching on the switching device and a voltage of approximately 0V for switching off the switching device. Switching on the switching device with a control voltage of approximately 0V and switching off the switching device with a positive control voltage is also possible.

A periodic, hard switching on and off of the at least one switching device with a predetermined input converter switching frequency fo can be used to generate an output signal with a pulsed profile from a continuous input signal. In this case, hard switching means very fast switching of the switching device in order to obtain an output signal with a very steep edge profile during the switching process. The switching frequency fo of the switching devices of the input converter can preferably be constant and can be designed in particular according to the components used in the converter circuit. The input currents can be regulated via pulse width modulation (P WM) of the input converter. The selection of the switching frequency fo is limited only by the switching losses of the input converter.

An m-phase transformer is always described below as a transformer configuration in which a primary side of the transformer has m phases, each with a primary winding, and a secondary side of the transformer has m phases, each with a secondary winding. In this case, the primary winding of the m-th phase of the primary side is always magnetically coupled to the secondary winding of the m-th phase of the secondary side. In other words, the transformer has m primary windings and m secondary windings. The m-phase transformer can preferably have a common core for all m phases, to be used more effectively, and the core may preferably be made of ferrite or amorphous metals. However, the m-phase transformer can also be configured by connecting m single-phase transformers together. The transformer core can also be made of any soft magnetic material. The primary side is isolated from the secondary side of the m-phase transformer. The use of the m-phase transformer in the converter circuit thus causes potential isolation between the input and the output of the converter circuit.

The power grid to which the converter circuit can be connected and/or to which the converter circuit can be connected can be a 1-phase power grid or a multi-phase power grid with any number n of phases. In this case, the converter circuit can preferably be connected on the input side to an n-phase AC network, the n network currents preferably being sinusoidal. Another AC shape such as a triangular shape is also possible. However, the converter circuit can also be connected to an n-phase direct current network and/or to n batteries.

The power converter circuit for a multi-phase AC network can preferably be connected to a symmetrical AC network, since the input power converter can then also be constructed symmetrically and the circuit design is simplified. The input power converter can have n input phases, with each input phase having its own connection terminal. In particular, the number of input phases of the input power converter and the number of phases of the AC network can be identical. However, the number n of input phases of the input converter can also be larger or smaller than the number of phases of the power grid, in which case not all phases of the power grid can be connected with a smaller number of input phases or not all input phases with one phase with a larger number of input phases of the power grid can be connected. In such a configuration, only the phases of the input converter that are actually connected are processed further. The flexibility in the design of the power converter circuit allows it to be used in a wide variety of applications not specific to one Type of power grids are limited. When connecting the power converter circuit to the power grid, it should be noted that an input inductance can be or is connected between each phase of the power grid and each input, ie on each input phase of the input power converter. The size of the input inductances depends on an input converter switching frequency fo of the switching devices used in the input converter. In this case, the input converter switching frequencies for an input converter without a subsequent DC link can be configured higher than for an input converter with a subsequent DC link, since charging and discharging a large capacitance in the DC link can be omitted. This means that smaller and less expensive input inductances can also be selected for input converters without a subsequent DC voltage intermediate circuit.

During operation, the input converter hard-switches its switching devices at an input converter switching frequency fo and a period of To = 1/ fo with To = TEIP + TOFF for one switching cycle and generates an input current for each input phase n, with the input currents at the output of the input converter becoming one pulsed output current are combined and / or added. When connecting to an n-phase AC network with n sinusoidal network currents, a sinusoidal input current is generated for each input phase. The pulse-shaped output current of the input converter is not fed into a DC voltage intermediate circuit and smoothed, as is the case with conventional converters, but is converted directly in its m phases into m oscillating circuit currents by the m-phase oscillating circuit converter. The phase angles of the m resonant circuit currents in the m phases of the resonant circuit converter are shifted relative to one another. For this purpose, all switching devices of the resonant circuit converter are switched with a resonant circuit converter switching frequency fo/m with a period duration of mTo=m/fo for one switching cycle, the switching devices of an mth phase being switched in such a way that oscillation of the resonant circuit currents in the resonant circuit of the m th phase is effected. Each m-th phase of the resonant circuit converter preferably has at least one switching device for a positive branch of the resonant circuit converter and at least one switching device for a negative branch of the oscillating circuit converter, the switching devices of the positive branch being switched with a time offset and preferably at the times when the current in the intermediate circuit assumes the value zero. The switching times for switching the m phases on and off differ and are shifted by a phase angle. In this case, the number of phases m of the resonant circuit converter is at least two and preferably an odd number of phases. The output signal of the input converter is provided or switched on at the input of each of the m phases by all switching devices of the respective phase during the period mTo for a switch-on period less than or equal to To as an input signal in a switched-on state and for a switch-off period greater than or equal to (m-1) To not made available or switched off in a switched-off state of the respective phase. The resonant circuit converter is preferably configured in such a way that the output signal of the input converter is always provided or switched on at exactly one of the m phases for a switch-on period less than or equal to To, while the output signal of the input converter is not made available at the other m-1 phases or is is switched off. In this way, the on periods of the m phases do not overlap. In addition, all switching devices of the resonant circuit converter are configured in such a way that they are preferably switched on and/or switched off at a point in time when the switching devices of the input converter are in a switched-off state during the time period TOFF. This prevents the switching devices of the resonant circuit converter from being overloaded during the switching operations. The switching on and off of the switching devices of the input converter and/or the m-phase oscillating circuit converter can be controlled and/or regulated by a control circuit. This control circuit can be designed as an independent control circuit for switching the switching devices, or it can be integrated into an overall control circuit.

In the oscillating circuit converter, an oscillating circuit consisting of an oscillating circuit capacitor and a main inductance of the transformer is configured for each phase, the main inductance being at least part of a primary-side winding in one phase of the m-phase transformer. In this case, the parallel resonant circuits are preferably operated at a resonant frequency or close to the resonant frequency. The oscillating circuit capacitors and also the main inductances can also be configured differently from one another, depending on the requirements placed on the power converter circuit. In general, resonant circuit capacitors can be configured on the primary and secondary side. In this case, the resonant circuit capacitors can be distributed centrally on the primary side or distributed on the primary side and the secondary side. The total capacitance of the oscillating circuits results from the entirety of the oscillating circuit capacitors. For the transmission from the primary side of the transformer to its secondary side, a magnetizing current is supplied per phase from the respective oscillating circuit capacitor into the winding transmitting on the primary side. The output voltages of the m-phase transformer, which are transmitted in isolated form to the secondary side of the transformer, are then rectified by the rectifier bridge and combined to form a common, isolated direct voltage. An accumulator to be charged can then be and/or connected to this DC voltage in order to be charged.

In a preferred embodiment of the converter circuit, the converter circuit can be connected to a 3-phase AC network and/or the resonant circuit converter and the transformer have three phases. Optionally, the main inductances of the transformer are configured in a star or delta arrangement.

The 3-phase alternating current network to which the converter circuit can be connected can preferably be a symmetrical three-phase network with a network frequency of 50Hz or 60Hz, which is available almost everywhere in both commercial and private environments. The phase angles of the three alternating voltages and currents of the three-phase system each differ by 120°. A three-phase configuration of the tank circuit converter and the transformer also simplifies the circuit design of the converter circuit. Here, the primary-side windings of the 3-phase transformer can be in a delta or Delta arrangement, in a star arrangement or in a zigzag arrangement. In addition, the secondary side windings of the 3-phase transformer can be configured in a delta or delta arrangement or in a star arrangement. Preferably, both the primary and secondary windings of the 3-phase transformer are configured the same and both sides of the transformer have the delta or delta configuration, or both sides of the transformer have the star configuration. However, mixed arrangements on the primary side and the secondary side of the 3-phase transformer are also possible. The main inductances of the three phases can each make up the entire winding or just part of it.

In a further preferred embodiment of the converter circuit, the converter circuit has a protective diode between the input converter and the m phases of the oscillating circuit converter and/or the pulsed output current of the input converter is configured to have a current intensity of 0 A when the m phases of the oscillating circuit converter are respectively switched. In this case, the output current of the input converter has a current intensity of 0 A when the switching devices of the input converter are in a switched-off state during the time period TOFF. For configurations in which the current intensity can deviate from 0A when switching the m phases of the oscillating circuit converter, a configuration with protective diodes should preferably be used to protect the converter circuit.

In a further preferred embodiment of the converter circuit, the m main inductances of the m-phase transformer can each be adjusted via an air gap. The m main inductances are set with the air gap during manufacture. In this way, the secondary-side output voltages of the m-phase transformer and the isolated DC voltage at the output of the converter circuit can be easily adjusted.

In a further preferred embodiment of the converter circuit, the rectifier bridge is designed as a midpoint circuit or as a B6 rectifier bridge configured. In this case, the ripple of the DC voltage at the output of the rectifier bridge can be further reduced by a smoothing capacitor. When the power converter circuit is configured as a midpoint circuit, there is only one diode in the current path during operation, as a result of which the losses at the diodes can be reduced. Therefore, the midpoint circuit is preferred for smaller output voltages, such as 48V. The diodes of the midpoint circuit have twice the blocking voltage of the diodes of the B6 rectifier bridge.

In a further preferred embodiment of the power converter circuit, the input power converter has a control circuit and the control circuit is configured to generate pulse patterns for switching the input power converter, the pulse patterns controlling the input power converter in such a way that the reactive power of the input power converter is minimized. The use of a reactive power-optimized control circuit can thus also increase the efficiency of the converter circuit. The control circuit is preferably also configured to generate pulse patterns for switching the switching devices of the oscillating circuit converter. The control circuit can be designed as an independent control circuit for controlling the reactive power, or it can be integrated into an overall control circuit.

In a further preferred embodiment of the converter circuit, the input converter switching frequency fo of the converter circuit can be adjusted before or during operation to optimize overvoltages and efficiency or to compensate for parameter deviations. In this case, the input power converter switching frequency fo can preferably be adjusted via the control circuit.

A further aspect for solving the problem relates to a converter circuit system for generating a potential-separated DC voltage, having: a plurality j of converter circuits as described above, the plurality j of converter circuits being arranged in parallel with one another, and j being a positive integer greater than 1. On the input side, the individual power converter circuits can be connected in parallel. In a single-phase supply on the mains side, the converter circuits can also be arranged in a series connection to reduce the voltage stress on the semiconductors and/or to reduce the distortions. On the secondary side of the converter circuits, there is always the possibility of parallel connection, series connection or a combination of both variants in order to minimize the distortion of the output current. When the converter circuits are connected in parallel, there is no power transmission via the resonant circuit capacitors and the power transmission also takes place in parallel, with the individual converter circuits needing to transmit less power than without the parallel connection. Therefore, in turn, the components of the individual power converter circuits can be made smaller. This in turn facilitates automated production of the entire power converter circuit system. When a plurality of converter circuits are connected in series, the power transfer across each of the converter circuits is much higher than when the converter circuits are connected in parallel.

In a preferred embodiment of the converter circuit system, the plurality j of converter circuits are configured in such a way that the pulse patterns of the input converters of the parallel-connected converter circuits are shifted by a phase angle 2 tt/j relative to one another with the same input converter switching frequency fo in relation to their period duration To. In this case, a phase angle shift is not limited to 2 π/j, but can also have a different phase angle, which preferably improves and/or optimizes the transformation ratio at the transformers. With this configuration, the ripple of the input current and the output current of the converter circuit system is reduced. The permissible ripple or current ripple of the output current is standardized here. The ripple can be kept below the standardized value by a sufficiently high number of phase-shifted power converter circuits arranged in parallel. With a sufficiently high number of parallel arranged, phase-shifted power converter circuits, the input inductances can be further reduced and the production of the power converter circuit system can in turn be automated more easily. Such a parallel connection is only possible due to the electrical isolation in the output stage.

A further aspect of solving the problem relates to the use of a converter circuit or a converter circuit system as described above for generating a potential-separated DC voltage.

Another aspect of solving the problem relates to a method for generating a potential-separated DC voltage using a converter circuit, the converter circuit having an m-phase oscillating circuit converter with one oscillating circuit capacitor per phase and an m-phase transformer, the oscillating circuit capacitor of the m-th phase of the oscillating circuit converter each forming a parallel resonant circuit with a main inductance of the m th phase of the transformer, and the method comprises: receiving n input currents; generating a pulsed current at a switching frequency fo from the n input currents; Converting the pulsed current into m resonant circuit currents at a respective switching frequency fo/m, the phase angle of the resonant circuit currents being shifted with respect to one another; providing a respective magnetizing current for the mth phase of the transformer through the resonant circuit capacitor of the mth phase of the resonant circuit converter; generating m secondary-side output voltages which are electrically isolated from the primary side of the transformer; rectifying the m secondary output voltages of the transformer and generating the isolated DC voltage; and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

A further aspect relates to a circuit for a power converter circuit for generating a potential-separated DC voltage, the circuit being electrically connectable to an m-phase transformer, the m-phase transformer being configured m secondary-side output voltages to generate electrically isolated from the primary side of the transformer, the m phases of the m-phase transformer being connected or connectable to a rectifier bridge on the secondary side and the rectifier bridge being configured to rectify the m secondary-side output voltages of the transformer and to generate the electrically isolated DC voltage at an output, wherein the circuit comprises: a switchable input power converter; and a switchable resonant circuit converter; wherein the input converter is configured to be connected to an n-phase power grid and to generate a pulse-shaped output current from n input currents at an input converter switching frequency f0; wherein the tank converter has m phases and is configured to convert the pulsed output current of the input converter into a tank current in each of the m phases at a tank converter switching frequency fO/m; wherein the m resonant circuit currents in the m phases are each shifted in phase angle to one another; wherein each of the m phases of the tank circuit converter has a tank circuit capacitor and, when connected to the m-phase transformer, forms a respective parallel tank circuit with a magnetizing inductance of the m-th phase of the transformer; wherein each m th phase tank capacitor of the tank circuit converter is configured, when connected to the m-phase transformer, to provide a magnetizing current for the m th phase of the transformer; and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

The statements made above or below regarding the embodiments of the respective other aspects also apply to the above-mentioned aspects and in particular to embodiments that are preferred in this regard.

In the following, individual embodiments for solving the task described by way of example with reference to the figures. Some of the individual embodiments described have features that are not absolutely necessary to implement the claimed subject matter, but which provide desired properties in certain applications. Thus, embodiments that do not have all the features of the embodiments described below are also to be regarded as being disclosed as falling under the technical teaching described. Furthermore, in order to avoid unnecessary repetition, certain features are only mentioned in relation to individual embodiments described below. It is pointed out that the individual embodiments should therefore not only be considered individually, but should also be viewed as a whole. Based on this synopsis, the person skilled in the art will recognize that individual embodiments can also be modified by incorporating individual or multiple features of other embodiments. It is pointed out that a systematic combination of the individual embodiments with individual or multiple features that are described in relation to other embodiments can be desirable and useful and should therefore be considered and should also be regarded as covered by the description.

Brief description of the drawings

Figure 1 shows a schematic representation of a

Converter circuit according to a preferred

embodiment of the present invention.

FIG. 2 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without protective diodes.

FIG. 3 shows a time course of input currents, an output current and an output voltage in an input converter according to a preferred embodiment of the present invention.

FIG. 4 shows a detailed time course of an output current in an input converter and of oscillating circuit currents in an oscillating circuit converter according to preferred embodiments of the present invention.

FIG. 5 shows a detailed time profile of switching signals and an output current in an input converter and of switching signals and oscillating circuit currents in an oscillating circuit converter according to a preferred embodiment

Embodiments of the present invention.

FIG. 6 shows a time curve of the signals from FIG. 5 spread over time with a comparison of the switching signals at switching times in the oscillating circuit converter according to preferred embodiments of the present invention.

Figure 7 shows a detailed time course of a resonant circuit current and associated voltages of a

Resonant circuit converter according to the preferred embodiment of the present invention from Figure 1.

Figure 8 shows a detailed time course of a resonant circuit current and associated voltages of a

Resonant circuit converter according to the further preferred embodiment of the present invention without

Protection diodes from Figure 2.

FIG. 9 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without

Protective diodes and an alternative primary-side transformer circuit. Figure 10 shows a schematic representation of a

Converter circuit according to a further preferred

Embodiment of the present invention without

Protective diodes and an alternative transformer circuit in a delta arrangement.

Figure 11 shows a schematic representation of a

Converter circuit according to a further preferred

Embodiment of the present invention without

Protection diodes and an alternative

rectifier bridge circuit.

Figure 12 shows a schematic representation of a

Converter circuit according to a further preferred

Embodiment of the present invention without

protection diodes and an alternative rectifier bridge with center point circuit.

Figure 13 shows a schematic representation of a

Converter circuit according to a further preferred

Embodiment of the present invention with an alternative transformer and rectifier bridge circuit.

Figure 14 shows a schematic representation of a

Converter circuit according to a further preferred

Embodiment of the present invention with an alternative transformer circuit and an alternative neutral point rectifier bridge.

Figure 15 shows a schematic representation of a

Converter circuit according to a further preferred

Embodiment of the present invention with an alternative transformer circuit. Figure 16 shows a detailed time course of mains currents of the AC mains, an output current of the input converter, an oscillating circuit current of the oscillating circuit converter, phase-to-phase transformer voltages of the transformer and a module output current according to the preferred embodiment of the present invention from Figure 15.

FIG. 17 shows a block diagram of a control circuit of an input converter according to a preferred embodiment of the present invention.

FIG. 18 shows a time course of the input voltages and the input currents, as well as the active and reactive power consumed in the event of a setpoint jump in the mains current, according to a preferred embodiment of the present invention.

FIG. 19 shows a schematic representation of a power converter circuit system with a plurality of power converter circuits connected in parallel according to a preferred embodiment of the present invention.

FIG. 20 shows a time profile of input currents in a converter circuit system with a different number of converter circuits connected in parallel, each with a time phase offset according to a preferred

embodiment of the present invention.

FIG. 21 shows a time profile of output currents in a converter circuit system with a different number of converter circuits connected in parallel, each with a time phase offset according to a preferred embodiment of the present invention.

FIG. 22 shows a schematic illustration of a converter circuit system with a plurality of converter circuits connected in parallel according to a further preferred embodiment of the present invention.

FIG. 23 shows a time profile of input currents in a converter circuit system with a different number of converter circuits connected in parallel, each with a time phase offset according to a preferred

embodiment of the present invention.

FIG. 24 shows a time profile of output currents in a converter circuit system with a different number of converter circuits connected in parallel, each with a time phase offset according to a preferred

embodiment of the present invention.

Detailed description of the figures

FIG. 1 shows a schematic representation of a power converter circuit 10 according to a preferred embodiment of the present invention. The converter circuit has an input converter 30, a resonant circuit converter 40, a transformer 50 and a rectifier bridge 60, viewed from the input side in the direction of the output. The power converter circuit 10 can be connected and/or connected to an AC network 20 on the input side and to a rechargeable battery 70 to be charged on the output side. It should be noted here that the transformer 50 in FIGS. 1, 2, 7 to 10 and 14 is shown in an idealized manner, ie without coil resistances, hysteresis losses in the core and stray inductances.

The AC network 20 in Figure 1 is designed as a 3-phase, symmetrical Three-phase network shown, but can also be a 1-phase AC network or a multi-phase AC network with any number of phases. In order to be able to connect the AC network 20 to the input converter 30 on the input side, an input inductance LE is preferably interposed between each phase of the AC network 20 and each input of the input converter 30 . In other words, an input inductor LE can be connected to each input, ie to each input phase of the input converter 30, via which the input converter can be connected to the AC network. The size of the input inductances LE depends on a switching frequency fo of the switching devices used in the input converter 30 and can be designed to be correspondingly smaller for a higher input converter switching frequency fo. The input converter 30 itself has two symmetrically arranged switching devices 30i and 30-1, 302 and 30-2, as well as 303 and 30-3 for each phase connected to its inputs, the input converter switching frequency fo of which is controlled via a control input by a control circuit 80 (Not shown in Figure 1) can be regulated. The switching devices are shown here as MOSFETs, but they can also be replaced by other conventional field effect transistors or bipolar transistors. The symmetrically arranged switching devices form a positive and a negative branch for each phase, and the outputs of the positive branch of all phases are combined and/or added in a common positive output. The outputs of the negative branch of all phases are also combined and/or added in a common negative output. During operation, the input converter 30 switches the switching devices hard, or with a very steep edge, at an input converter switching frequency fo and generates a sinusoidal input current ii, h and h for each phase at an input voltage ui, U2 and U3. During operation, a pulse-shaped output voltage UDC, which approximately represents a direct voltage, is present between the common positive and negative outputs of the input converter 30 . The combined and/or added output current IDC also has a pulsed profile.

The pulse-shaped output current IDC is not as in the usual Power converters fed into a DC link, but serves as input current for the oscillating circuit converter 40. The oscillating circuit converter 40 in Figure 1 here has three phases, but can also be divided into two or more phases. For each phase, the resonant circuit converter 40 has two symmetrically arranged switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3, each with an upstream protective diode 42, in which a switching device 40i, 402 and 403 with the positive branch of the output of the input converter 30 and the other switching device 40-1, 40-2 and 40-3 is connected to the negative branch of the output of the input converter 30. The switching devices are switched with a resonant circuit converter switching frequency fo/3 and thereby generate per phase an oscillating circuit current i _a , ib and _ic at a respective voltage of u _a , Ub and u _c between the three phases and the negative branch of the input converter 30. No switching losses occur when switching the switching devices of the oscillating circuit converter 40, since the Switching devices each gescha in a currentless state of the pulsed output current IDC of the input converter 30 become old. For this reason, the protective diodes 42 are not absolutely necessary if the pulsed output current IDC is regulated in a stable manner. In the oscillating circuit converter 40, an oscillating circuit consisting of an oscillating circuit capacitor CP and a main inductance LH is configured per phase, the main inductance LH being part of a primary-side winding in one phase of a 3-phase transformer. In this case, in FIG. 1, the main inductance LH per phase is arranged in parallel with a second primary-side inductance, and the three-phase transformer is constructed in a star shape on the primary and secondary sides. The voltage between the neutral point of the transformer and the negative branch of the input converter 30 is designated by the voltage uo-. In FIG. 1, only the second inductance on the primary side transfers to the secondary side of transformer 50. For the transfer from the primary side of transformer 50 to its secondary side, a magnetizing current is supplied by the respective resonant circuit capacitor CP for each phase. This results for each phase from the difference between the resonant circuit currents i _a , ib and ic and currents ih _a , _ihb and ihc through the main inductances LH. In order to fix the resonant circuits in the respective phases, the main inductances LH can be configured to be adjustable via an air gap. The output voltages u _ab , Ubc and u _ca of the 3-phase transformer, which are transmitted to the secondary side of the transformer with potential isolation, are rectified by the rectifier bridge 60 and an output current iout is provided at an output voltage u _0U t at the output of the converter circuit 10 . In this case, in FIG. 1, the rectifier bridge 60 is configured in a B6 configuration and the output voltage is smoothed at the output by a smoothing capacitor CG.

Figure 2 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42. The configuration of the converter circuit 10 is identical to the configuration in Figure 1 apart from the omission of the optional protective diodes 42. Omitting the protective diodes 42 can take place with small leakage inductances in the transformer 50 and enables a reduction in the power loss in the converter circuit 10 and a cost reduction in the layout of the converter circuit 10.

FIG. 3 shows a time profile of input currents ii and b, an output current IDC and an output voltage UDC in an input converter 30 according to a preferred embodiment of the present invention. The behavior of the converter circuits 10 from Figures 1 and 2 in the further description of the figures is always shown with a mains frequency of 500 Hz, with an input inductance LE of 200 mH, with an input converter switching frequency fo of the input converter 30 of 83 kHz, with a capacitance of the resonant circuit capacitor CP of 188nF, a main inductance LH of 169mH and a voltage at the accumulator 70 of 800V are simulated. The axis diagrams for the time curves in Figure 3 always show the time curve on the x-axis (0.0002s steps) and on the y-axis the input currents ii, b and b (20A steps), the output current IDC (10A steps) and the output voltage UDC (100V steps). Here, for Figures 1 and 2, the phase-shifted, sinusoidal input currents ii, and b, have a jagged course with the switching frequency of 83 kHz of the input converter 30 due to the hard switching of the switching devices of the input converter 30 on. The pulse-shaped output voltage UDC has an approximately DC voltage profile and the pulse-shaped output current IDC is composed of the combination and/or addition of the input currents ii, h and h. The invention is not limited to the above frequencies, voltages and currents. Rather, these examples merely serve to illustrate the inventive principle.

Figure 4 shows a detailed time course of an output current IDC in an input converter 30 and of resonant circuit currents i _a , ib and _ic in a resonant circuit converter 40 according to the preferred embodiments of the present invention from Figures 1, 2, 9 to 15, 19 and 22. The axis diagrams for the time curves in FIG. 4 always show the extended time curve on the x-axis (0.00005s steps) and on the y-axis the output current IDC (10A steps) and the resonant circuit currents i _a , ib and i _c ( 20A steps). In this case, the output current IDC of the input converter 30 is shown with a higher temporal resolution than in FIG. 3 in order to clarify the pulse-shaped profile. The oscillating circuit currents i _a , ib and _ic of the oscillating circuit converter 40 result from the output current IDC of the input converter 30 by phase-shifted switching of the switching devices of the oscillating circuit converter 40 at a switching frequency of fo/3. The oscillating circuit currents i _a , ib and _ic have a symmetrical, pulsed course with negative and positive amplitude values.

FIG. 5 shows a detailed time course of switching signals at the switching devices 30i and 30-1, 302 and 30-2, and 303 and 30-3 and an output current IDC in an input converter 30 and of switching signals at the switching devices 40i and 40-1, 402 and 40-2, as well as 403 and 40-3 and resonant circuit currents i _a , ib and i _c in a resonant circuit converter 40 according to the preferred embodiments of the present invention from FIGS the time curves in FIG. 5 always show the time curve on the x-axis (0.00002s steps) and on the y-axis the control voltages of the switching devices 30i and 30-1, 302 and 30-2, and 303 and 30- 3 of the input converter 30 (step: 0V to control voltage), the output current IDC (50A steps), the control voltages of Switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 des

Oscillating circuit converter 40 (step: OV to control voltage) and the

Resonant circuit currents i _a , ib and _ic (100A steps). This creates the

Control circuit 80 sends switching signals and uses a control input of the respective switching device to control the switching of switching devices 301 and 30-1, 302 and 30-2, as well as 303 and 30-3 of input converter 30, and the switching of switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 of the resonant circuit converter 40. The switching signals are each as

Pulse pattern generated by pulse width modulation.

The pulse patterns for the control inputs of the switching devices 30i and 30-1, 302 and 30-2, and 303 and 30-3 of the input converter 30 each have a control voltage for switching on the switching device (e.g 5V) and for a switch-off period TOFF a control voltage for switching off the switching device (e.g. OV). These pulse patterns are respectively switched on and off complementarily for the two symmetrically arranged switching devices 30i and 30-1, 302 and 30-2, and 303 and 30-3. In addition, the pulse widths and switching times of the pulse patterns (TEI _P ) of the three phases of the input converter 30 are matched to one another in such a way that a pulse-shaped output current bc is generated from the sum of the sinusoidal input currents h and b.

The pulsed output current bc of the input converter 30 is provided at the input of the m phases of the oscillating circuit converter 40 at the switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3. The pulse patterns for the control inputs of the switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 of the oscillating circuit converter 40 each have a control voltage during a period duration 3To=3/fo for a switch-on period less than or equal to To Turning on the switching device (e.g. 5V) and a control voltage for turning off the switching device (e.g. OV) for an off period greater than or equal to 2To. The switching processes at the switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 always take place at times at which the pulsed output current bc des Input converter 30 has a value of 0A, so no current at the input of the resonant circuit converter 40 flows. In this way, there are no switching losses at the switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 of the resonant circuit converter 40. In addition, the switch-on periods of the three switching devices 40i, 402 and 403 are preferably on the positive branch of the inputs am Oscillating circuit converter 40 each offset in time by To, so that only at the input of one phase of the three phases is a positive amount of the pulsed output current bc of the input converter 30 present. Likewise, the switch-on periods of the three switching devices 40-1, 40-2 and 40-3 on the negative branch of the inputs on the oscillating circuit converter 40 are preferably offset in time by To, so that only at the input of one phase of the three phases is there a negative amount of the pulsed output current bc of the input converter 30 is applied. In this case, the switch-on times of the symmetrical switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 are preferably offset in time by 1.5 To. The sum of the currents through the respective symmetrical switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 then result in the oscillating circuit currents i _a , ib and _ic in the oscillating circuit converter 40.

Figure 6 shows a time profile of the signals from Figure 5 stretched out over time with a comparison of the switching signals at switching times in resonant circuit converter 40 according to the preferred embodiments of the present invention from Figures 1 and 2. The axis diagrams for the time profiles in Figure 6 always show the extended time course on the x-axis (0.000005s steps) and on the y-axis the same representations as in Figure 5. In particular, Figure 6 shows that switching operations on the switching devices 40i and 40-1, 402 and 40-2, as well 403 and 40-3 always take place at points in time at which the pulsed output current bc of the input converter 30 has a value of 0A.

FIG. 7 shows a detailed time profile of a resonant circuit current i _a and associated voltages of a resonant circuit converter 40 according to the preferred embodiment of the present invention from FIG. 1 with protective diodes 42. The axis diagrams for the time profiles in FIG always the time course on the x-axis (0.00002s steps) and on the y-axis the resonant circuit current i _a (20A steps), the negative voltages u _a -, Ub- and Uc- of the resonant circuit converter 40 (200V steps) , the secondary-side output voltages u _a b, Ubc and u _ca at the transformer 50 (200V steps) and the voltage uo- between the star point of the transformer 50 and the negative branch of the input converter 30 (100V steps). In doing so, he will

Oscillating circuit current i _a compared to FIG. 4 with a higher temporal resolution. In addition, the resulting time profiles of the negative voltages u _a -, Ub- and u _c - of the resonant circuit converter 40, the secondary-side output voltages u _a b, Ubc and u _ca at the transformer 50 and the voltage uo- between the star point of the transformer 50 and the negative branch of the input converter 30 is shown.

Figure 8 shows a detailed time profile of a resonant circuit current i _a and associated voltages of a resonant circuit converter 40 according to the further preferred embodiment of the present invention without protective diodes from Figure 2. The axis diagrams for the time profiles in Figure 8 correspond to the axis diagrams from Figures 7 and always show the time course on the x-axis (0.00002s steps) and on the y-axis the oscillating circuit current i _a (20A steps), the negative voltages u _a -, Ub- and Uc- of the oscillating circuit converter 40 ( 100V steps), the secondary-side output voltages u _a b, Ubc and u _ca at the transformer 50 (200V steps) and the voltage uo- between the star point of the transformer 50 and the negative branch of the input converter 30 (100V steps). In this case, the omission of the protective diodes 42 in FIG. 2 results in a smoothing of the voltages, as can be seen in comparison with FIG.

Figure 9 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative primary-side transformer circuit 50. The embodiment in Figure 9 is based on the embodiment from Figure 2. In contrast to the primary side of the transformer 50 in FIG. 2, no second primary-side inductance is configured in parallel with the main inductance LH per phase, but rather the main inductance LH is also used for transmission im Transformer 50 to the secondary side.

FIG. 10 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative transformer circuit 50 in a delta arrangement. The embodiment in FIG. 10 is based on the embodiment from FIG. 9. In contrast to the transformer 50 in FIG. 9 with a star-shaped configuration of the main inductances LH, the main inductances LH in FIG. 10 are configured in a delta arrangement. In this case, the resulting capacitance of the resonant circuit capacitors CP facing the input converter 30 is greater in the delta arrangement than in the star connection. Among other things, this leads to lower overvoltages in the converter circuit 10.

Figure 11 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative rectifier bridge circuit 60. The embodiment in Figure 11 is based on the embodiment from Figure 9. In contrast to the rectifier bridge 60 in Figure 9 with In a B6 configuration, FIG. 11 shows a rectifier bridge with additional resonant circuit capacitors CP on the secondary side. These oscillating circuit capacitors CP form an oscillating circuit on the secondary side with the main inductance of the transformer 50 . The division of the oscillating circuit capacitors CP serves to protect individual components in the converter circuit 10. The total capacitance of the oscillating circuits in the converter circuit 10 results from the entirety of the oscillating circuit capacitors CP.

FIG. 12 shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative rectifier bridge 60 with a midpoint connection. Here, the embodiment in FIG. 12 is based on the embodiment from FIG. 9. In contrast to the rectifier bridge 60 in FIG. 9 with a B6 configuration, a rectifier bridge with a midpoint connection is shown in FIG. The output voltage u _0U t of Centers of the secondary main inductances to one side of the main inductances and thus has half of the output voltage Uout in a B6 configuration of the rectifier bridge 60.

In order to allow the output voltage u _0U t to be reduced further independently of the dimensioning of the transformer 50, a step-down converter circuit can additionally be configured between the rectifier bridge 60 and the accumulator 70.

In order to feed back electrical energy from the accumulator 70 into the AC network 20, a step-up converter circuit can additionally be configured between the rectifier bridge 60, if this is designed as an active bridge with transistors, and the accumulator 70.

Figure 13 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention with an alternative transformer circuit 50 and rectifier bridge circuit 60. The embodiment in Figure 13 is based on the embodiment from Figure 1, but represents the transformer circuit 50 as an equivalent circuit diagram of a real transformer 50 with leakage inductances. A distributed arrangement of the oscillating circuit capacitors CP can integrate parasitic capacitances and/or inductances better into the circuit concept, depending on the configuration. In this case, the specific configuration depends on the structure of the overall arrangement.

FIG. 14 shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention with an alternative transformer circuit 50 and an alternative rectifier bridge 60 with a midpoint connection as already explained for FIG. In addition, protection diodes 42 are shown at the input of the tank circuit converter 40 as an optional configuration. The midpoint circuit in turn reduces the losses of the output rectification. However, the required blocking voltage of the diodes is doubled. Such a configuration can be used for a power converter circuit 10 with smaller output voltages. FIG. 15 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention with an alternative transformer circuit 50. In addition, protective diodes 42 are shown at the input of the oscillating circuit converter 40 as an optional configuration. Here, the transformer 50 is shown as a real transformer, ie with coil resistances and leakage inductances.

The converter circuit 10 from Figure 15 is operated with a 3-phase input converter 30, oscillating circuit converter 40 and transformer 50 in a further simulation on a 3-phase, symmetrical three-phase network with a mains frequency of 50Hz and a mains voltage of 230V per phase and 400V overall and represents a test design for a 50 kW converter circuit 10. An inductance of 40 mH is selected as the input inductance LE for each phase of the input converter 30 and the input converter 30 is operated with an input converter switching frequency fo of 83 kHz. The capacitance of the tank circuit capacitor CP per phase of the tank circuit converter 40 is set at 195 nF, the magnetizing inductance LH of the transformer 50 at 175 mH, and the total leakage inductance of the transformer 50 at 2 mH. The transformation ratio of the transformer is ü=1. For an output filter of the rectifier bridge 60, a capacitor parallel to the output has a value of 10pF and an inductance in series with the output has a value of 5mH. assumed. The voltage at the accumulator is fixed at 800V. In the case of the topology without protective diodes, the costs and the power loss in the converter circuit 10 are in turn reduced. However, the leakage inductances must not become too great in this case.

Figure 16 shows a detailed time course of mains currents of the AC network 20, an output current of the input converter 30, a resonant circuit current of the resonant circuit converter 40, line-to-line transformer voltages of the transformer 50 and a module output current according to the preferred embodiment of the present invention from Figure 15. The axis diagrams for the time courses in 16 always shows the time course on the x-axis (0.00001s steps) and on the y-axis the input currents h, b and h (50A steps), the output current bc (20A steps), the resonant circuit current i _a (50A steps), the phase-to-phase transformer voltages UTrafo (500V steps) and the module output current iout (50A steps). FIG. 16 shows the real time profiles with a leakage inductance in the transformer 50, which are always present in a real power converter circuit 10. In detail, the phase-shifted, sinusoidal line currents ii, b and b have a jagged course with the switching frequency of 83 kHz of the input converter 30 due to the hard switching of the switching devices of the input converter 30 . The pulsed output current IDC is made up of the combination and/or addition of the input currents ii and b. The oscillating circuit current i _a of the oscillating circuit converter 40 results from the output current IDC of the input converter 30 by phase-shifted switching of the switching devices of the oscillating circuit converter 40 at a switching frequency of fo/3. In this case, the oscillating circuit current i _a has a symmetrical, pulse-shaped profile with negative and positive amplitude values. The phase-to-phase transformer voltages of the transformer 50 result from the interaction of the oscillating circuit currents i _a , ib and _ic. At the output of the rectifier bridge 60 there is a rippled module output current, which is smoothed out by a filter circuit before it is fed to the accumulator 70 .

FIG. 17 shows a block diagram of a control circuit 80 of an input converter 30 according to a preferred embodiment of the present invention. In this case, the reactive power consumed by the input converter 30 is to be regulated to zero with the control circuit 80 . For this purpose, the input voltages ui, U2 and U3 and the input currents ii and b are converted in a space vector converter 82 into mains voltage and current space vectors. The reciprocal amount of ÜN is calculated in a reciprocal converter 84 from the ab components UN and UNP of the grid voltage space vector. This reciprocal absolute value is then multiplied by the ab components of the mains voltage in two first multipliers 86, resulting in ab components with the amplitude 1 one. These ß components of the voltage are then multiplied by a current setpoint INSOII of the superimposed control in two second multipliers 88 and thereby the nominal values iNasoii and iNßsoii of the ß mains currents are formed. These current setpoints iNasoii and iNßsoii are then phase-shifted in a current setpoint phase-shifting unit 90, which will be discussed again below. These phase-shifted desired current values are compared with the actual current values INO and iNß in a comparison element 92 and supplied to ß current regulators 94 . To improve the dynamics, at the output of the down-current controller 94, a disturbance variable feed-in 96 is carried out with the down-components of the mains voltage. Desired control voltages USRQSOII and usRßsoii are generated as an output signal and are made available to the input converter 30 for pulse pattern generation. The desired control voltages USRQSOII and usRßsoii are desired values of the average power converter output voltage over a period. The target voltages are converted into the corresponding pulse widths of the pulse width modulation for the switching devices 30i to 303. A reactive power control takes place in the core element of the control circuit 80 . For this purpose, the reactive power consumed is calculated from the ß components of the network variables in a reactive power calculation unit 98 and compared with the reactive power value of zero. A reactive power arises when there is a phase shift between the input voltages ui, U2 and U3 and the input currents ii, b and h. Due to the control of alternating variables, there is always a phase shift between the actual current value and the current setpoint. A reactive power controller 100 generates a phase angle cp which shifts the phase of the target value of the mains currents in such a way that no reactive power is consumed. Due to the possibility of using very high switching frequencies, the control circuit 80 has high control dynamics with a very stable control behavior at the same time. In the simulation data shown here, for example, a switching frequency fo of the input converter 30 of 83 kHz is used, which allows the use of SIC components with a simplified design of the converter circuit 10.

In addition to controlling the reactive power consumed by input converter 30, control circuit 80 preferably has additional control units which switch switching devices 30i and 30-1, 302 and 30-2, and 303 and 30-3 of input converter 30 and 303 and 30-3 on and off controls and/or regulates the switching devices 40i and 40-1, 402 and 40-2, and 403 and 40-3 of the 3-phase resonant circuit converter 40 on and off. The control units for switching the switching devices of the input converter 30 and the oscillating circuit converter 40 on and off can also be designed as a separate control circuit separate from the control circuit 80 .

As an alternative or in addition to the regulation of the switching frequencies fo and fo/3 by the regulation circuit 80, an L-C-L filter can be connected upstream on the mains side. In this way, voltage-impressing control is possible instead of current-impressing control.

FIG. 18 shows a time course of the input voltages ui, U2 and U3 and the input currents ii, h and h, as well as the consumed active power p(t) and reactive power q(t) in the event of a setpoint jump in the mains current according to a preferred embodiment of the present invention. The axis diagrams for the time curves in Figure 18 always show the time curve on the x-axis and on the y-axis the input currents i-i, and h, the input voltages ui, U2 and U3, the active power consumed p(t) and the consumed Reactive power q(t). It is clear here that the control circuit 80 controls the input converter 30 of the converter circuit in such a way that the reactive power q(t) consumed is again at zero shortly after the setpoint jump. A mains frequency of the feeding mains of 500 Hz was selected for the simulation in FIG. 18 for better representation.

FIG. 19 shows a schematic representation of a converter circuit system 1 with a plurality j of parallel-connected converter circuits 10 according to a preferred embodiment of the present invention. Here, the embodiments of the power converter circuit 10 described above are configured in parallel between the AC grid 20 and the accumulator 70 . This parallel connection is possible due to the potential isolation in the output stage of the individual power converter circuits 10 . The pulse pattern of the input converter 30 of the parallel-connected converter circuits 10 at same input power converter switching frequency fo each shifted by a phase angle 2 tt/j relative to each other with respect to their period duration To. With the phase-shifted switching sequence of the parallel power converter circuits 10, the ripple of the input currents ii and h can also be reduced with the same physical size of the input inductances LE. If the ripple is optimized, it is possible to further reduce the overall size of the input inductances LE.

FIG. 20 shows a time profile of input currents 11 in a converter circuit system 1 with a different number of converter circuits 10 connected in parallel, each with a time phase offset according to a preferred embodiment of the present invention. The

Axis diagrams for the time courses in FIG. 20 always show the

Time history on the x-axis (0.000005s steps) and on the y-axis three input currents 11 from three parallel connected power converter circuits 10 (0.5A steps), the combined input current 11 from three parallel connected power converter circuits 10 (0.2A steps) and the combined input current I1 from six parallel-connected converter circuits 10 (0.1A steps). Here, the figure 20a shows the parallel time course of three individual input currents 11 of three parallel-connected power converter circuits 10 in a diagram, the input currents 11 of the parallel

Power converter circuits 10 by phase-shifted circuit of

Input converter 30 are generated. FIG. 20b now shows the combined input current 11 of the current circuit system 1 consisting of the three added input currents 11 of the parallel, phase-shifted input converters 30, whose amplitude was divided by the number of three parallel converter circuits 10. The time offset of the phases results in To/3. FIG. 20c shows the combined input current 11 of a current circuit system 1 consisting of six converter circuits 10 connected in parallel. The time offset of the phases results in To/6. In comparison to FIG. 20b, it is clear here that the ripple of the combined input current 11 can be reduced by a higher number of converter circuits 10 connected in parallel. FIG. 21 shows a time curve of output currents iout in a converter circuit system 1 with a different number of converter circuits 10 connected in parallel, each with a time phase offset according to a preferred embodiment of the present invention. The axis diagrams for the time curves in FIG. 21 always show the time curve on the x-axis and on the y-axis three output currents iout from three parallel-connected converter circuits 10 (5A steps), the combined output current iout from three parallel-connected converter circuits 10 ( 2A steps) and the combined output current iout of six parallel-connected converter circuits 10 (2A steps). In this case, FIG. 21a, comparable to FIG. 20a, shows the parallel time profile of three individual output currents iout from three converter circuits 10 connected in parallel in a diagram. FIG. 21b then shows the addition of the three individual output currents iout from FIG. 21a. Comparable to FIG. 20c, FIG. 21c again shows the addition of six individual output currents iout of a current circuit system 1 consisting of six converter circuits 10 connected in parallel. Here, too, the ripple of the combined output current iout is reduced by a higher number of converter circuits 10 connected in parallel.

FIG. 22 shows a schematic representation of a converter circuit system 1 with a plurality of parallel-connected converter circuits 10 according to a further preferred embodiment of the present invention. Figure 22 shows a complete equivalent circuit diagram of the power converter circuit 10 from Figure 19.

FIG. 23 shows a time profile of input currents ii and h in a converter circuit system 1 with a different number of converter circuits 10 connected in parallel, each with a time phase offset according to a preferred embodiment of the present invention. The axis diagrams for the time curves in FIG. 23 always show the time curve on the x-axis (0.002s steps) and on the y-axis three input currents ii and h from a converter circuit 10 (50A steps), the input currents ii, h and h of three converter circuits connected in parallel 10 (100A steps) and the combined input currents ii, and b of six parallel-connected converter circuits 10 (200A steps). It is clear here that an increase in the number of converter circuits 10 connected in parallel results in an increase in the reduction in current ripple.

FIG. 24 shows a time course of output currents in a converter circuit system 1 with a different number of converter circuits 10 connected in parallel, each with a time phase offset according to a preferred embodiment of the present invention. The axis diagrams for the time curves in FIG. 24 always show the time curve on the x-axis (0.000005s steps) and on the y-axis the individual output currents iout of three parallel-connected converter circuits 10 (50A steps), the combined output current iout of three power converter circuits 10 connected in parallel (50A steps) and the combined output current iout of six power converter circuits 10 connected in parallel (50A steps). In comparison to FIG. 21, stray inductances in the transformer 50 are taken into account. It is again clear that with an increase in the number of converter circuits 10 connected in parallel, an increase in the current ripple reduction in the output current is achieved.

Reference List

1 power converter circuit system

10 converter circuit

20 AC power grid

30 input converters

301..m switching devices of the input converter, positive branch

30-1 -m switching devices of the input converter, negative branch 40 resonant circuit converter

401..m switching devices of the oscillating circuit converter, positive branch

40-1 -m switching devices of the oscillating circuit converter, negative Ast42 Protection diodes 50 transformer

60 rectifier bridge

70 accumulator

80 control circuit

82 space vector converter

84 reciprocal converter

86 first multipliers

88 second multipliers

90 current command phase shift unit

92 comparator

94 ate current regulator

96 Disturbance control

98 reactive power calculation unit

100 reactive power controllers

LE Input inductance of the input power converter

C _P resonant circuit capacitor

CG smoothing capacitor

Lh main inductance of the transformer ii..n input current of the nth phase of an AC network ui..n input voltage of the nth phase of an AC network

IDC Pulsed output current of the input converter

UDC Pulse-shaped output voltage of the input converter ia.b.c Resonant circuit currents in the m-phases of the resonant circuit converter iha.hb.hc Currents through the main inductances in the m-phases

Ua.b.c Voltages between the mth phase of the resonant circuit converter and the negative branch of the input converter Uab.bc.ca Secondary-side output voltages at the transformer uo- Voltage between the star point of the transformer and the negative branch of the input converter iout Output current of the converter circuit

Uout Output voltage of the power converter circuit

UNa.ß ß components of the input voltage space vector

UNa.ßsoii Target values of the ß components of the input voltage space vector ΐNa,b ab-components of the input current space vector iNa.ßsoii target values of the ass-components of the input current space vector ÜN reciprocal amount of the ab-components des

input voltage space vector TNSOII current setpoint usRa.ßsoii setpoint control voltage for pulse pattern generation in the input converter f phase angle

Q reactive power p(t) active power consumed q(t) reactive power consumed n number of input phases of the input converter m number of phases of the oscillating circuit converter and the transformer j number of converter circuits connected in parallel fo input converter switching frequency

To period at an input converter switching frequency fo fo/m resonant circuit converter switching frequency

Claims

patent claims

1. Converter circuit (10) for generating a potential-separated DC voltage (u ₀ ut), comprising: a switchable input converter (30); a switchable resonant circuit converter (40); an m-phase transformer (50); and a rectifier bridge (60); wherein the input converter (30) is configured to be connected to an n-phase power grid (20) and to generate a pulse-shaped output current (IDC) ZU from n input currents (h; ; b) at an input converter switching frequency fo; wherein the tank converter (40) has m phases and is configured to convert the pulsed output current (IDC) of the input converter (30) in each of the m phases at a tank converter switching frequency fo/m into a tank current (i _a ; b; ic) to convert wherein the m resonant circuit currents (i _a ; b; ic) in the m phases are each shifted in phase angle to one another; each of the m phases of the oscillating circuit converter (40) having an oscillating circuit capacitor (C _P ) and forming a respective parallel oscillating circuit with a main inductance (U) of the m th phase of the transformer (50); wherein in each case the resonant circuit capacitor (C _P ) of the mth phase of the resonant circuit converter (40) is configured to provide a magnetizing current for the mth phase of the transformer (50) and the transformer (50) is configured to output m secondary-side output voltages (u _ab ; to generate Ub _G ; u _ca ) isolated from the primary side of the transformer (50); wherein the m phases of the transformer (50) are connected to the rectifier bridge (60) on the secondary side and the rectifier bridge (60) is configured to rectify the m secondary-side output voltages (u _ab ; Ub _G ; u _ca ) of the transformer (50) and at one to generate the potential-separated direct voltage (u _0Ut ) at the output; and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

2. Converter circuit (10) according to claim 1, wherein the converter circuit (10) can be connected to a 3-phase AC network (20); and/or wherein the oscillating circuit converter (40) and the transformer (50) have three phases and optionally the main inductances (U) of the transformer (50) are configured in a star configuration or in a delta configuration.

3. converter circuit (10) according to claim 1 or 2, wherein the converter circuit (10) between the input converter (30) and the m phases of the resonant circuit converter (40) each having a protective diode (42); and/or wherein the pulse-shaped output current (IDC) of the input converter (30) is configured to have a current intensity of 0 A when the m phases of the oscillating circuit converter (40) are respectively switched.

4. Power converter circuit (10) according to any one of claims 1 to 3, wherein the m main inductances (U) of the m-phase transformer (50) are each adjustable via an air gap; and/or wherein the rectifier bridge (60) is configured as a midpoint circuit or as a B6 rectifier bridge.

5. converter circuit (10) according to any one of claims 1 to 4, wherein the input converter (30) has a control circuit (80), and the control circuit (80) is configured to generate pulse patterns for switching the input converter (30); and wherein the pulse patterns regulate the input converter (30) in such a way that the reactive power (Q) of the input converter (30) is minimized.

6. converter circuit (10) according to one of claims 1 to 5, wherein the input converter switching frequency fo of the converter circuit (10) can be adjusted to optimize overvoltages and efficiency or to compensate for parameter deviations before or during operation.

7. converter circuit system (1) for generating a potential-separated DC voltage (u ₀ ut), comprising: a plurality j of converter circuits (10) according to any one of claims 1 to 6; wherein the j plurality of power converter circuits (10) are arranged in parallel with one another, and wherein j is a positive integer greater than 1.

8. The power converter circuit system (1) according to claim 7, wherein the plurality j of power converter circuits (10) are configured in such a way that the pulse patterns of the input power converters (30) of the parallel-connected power converter circuits (10) at the same input power converter switching frequency fo each with respect to their period To by one Phase angle 2 tt / j are shifted from each other.

9. Use of a converter circuit (10) or a converter circuit system (1) according to any one of claims 1 to 8 for generating a potential-separated DC voltage (u ₀ ut).

10. A method for generating a potential-separated DC voltage (u _0U t) by a converter circuit (10), the converter circuit (10) having an m-phase oscillating circuit converter (40) with an oscillating circuit capacitor (C _P ) per phase and an m-phase transformer ( 50), wherein the oscillating circuit capacitor (C _P ) of the mth phase of the oscillating circuit converter (40) forms a respective parallel oscillating circuit with a main inductance (U) of the mth phase of the transformer (50), and the method has:

receiving n input streams (ii; 12; h);

generating a pulsed current (IDC) at a switching frequency fo from the n input currents (h; 12; h);

Converting the pulsed current (IDC) into m oscillating circuit currents (i _a ; it _> ; ic) each at a switching frequency fo/m, the phase angle of the oscillating circuit currents (i _a ; it _> ; ic) being shifted relative to one another;

Providing a magnetizing current for the m-th phase of the transformer (50) through the tank circuit capacitor (C _P ) of the m-th phase of the tank circuit converter (40);

Generation of m secondary-side output voltages (u _at> ; Ub _C ; u _ca ), which are electrically isolated from the primary side of the transformer (50); rectifying the m secondary-side output voltages (u _ab ; Ub _C ; u _ca ) of the transformer (50); and

Generation of the isolated DC voltage (u ₀ ut); and wherein n is a positive integer and m is a positive integer greater than or equal to 2.