EP0459571A1 - Microstrip slow wave transmission line and circuit including such a line - Google Patents

Abstract

Wave transmission line, in slow wave mode, of the so-called microstrip type, including a first so-called lower conductive layer (11) acting as earth plane, a second so-called upper conductive layer (12) in the form of a strip with specific longitudinal and transverse dimensions, and a third non-conductive material (1, 2) arranged between these two conductive layers. This transmission lines has, longitudinally, a periodic structure, each period, of length l, being formed of a so-called bridge (4) followed by a so-called pillar (13). Each bridge consists of a segment of the upper conductive strip (12), of length l1 < l, arranged at the surface of a so-called first part (1) of the third material, which part is dielectric in nature. Moreover each pillar (13) is a capacitance which can be an active or passive element. Furthermore, the first conductive layer (11) can be provided with recesses (5) beneath each bridge. A directional coupler (50) can be produced by means of such slow wave lines, and used to produce an integrated transceiver device with single antenna. Application: monolithic microwave integrated circuits. <IMAGE>

Description

The invention relates to a wave transmission line, in slow wave mode, of the so-called microstrip type, including a first so-called lower conductive layer acting as ground plane, a second so-called upper conductive layer in the form of a ribbon of transverse dimensions and longitudinal specific, and a third non-conductive material disposed between these two conductive layers.

The invention also relates to couplers formed from such lines.

The invention also relates to circuits including such a line.

The invention relates, among these circuits, to a transceiver device including an integrated circuit comprising a frequency duplexer for transmitting a first signal and receiving a second signal on a single pole.

The invention particularly finds its application in the production of integrable transmission lines, that is to say that can be included in integrated circuits, and more particularly in monolithic and microwave integrated circuits known under the name of MMIC's (of English: Monolithic Microwave Integrated Circuits).

In general, the invention finds its application in the miniaturization of transmission lines and allows the increase in the integration density of integrated circuits including these lines, and / or the increase in the operating performance of these circuits. .

In the case where the integrated circuit comprising the frequency duplexer is used, the invention finds its application in transmission and reception in the microwave domain by means of a single antenna, the transmitted signals being isolated from the signals transmitted by this single antenna by means of the integrated duplexer.

A microstrip type transmission line is described in the publication entitled: "Properties of Microstrip Line on Si-So₂ System", by HIDEKI HASEGAWA, et alii, in "IEEE Transactions on Microwave Theory and Techniques, vol.MTT-19, N ^° 11, November 1971, pp. 869-881 ".

According to the cited document, a microstrip type line consists of a stacked structure formed of a metal layer acting as a ground plane, of a semiconductor layer of silicon (Si), of a dielectric layer of silica (SiO₂ ) and a metallic ribbon of predetermined transverse dimension.

This document teaches that such a line admits the propagation of three fundamental modes. The first is a quasi-TEM mode, the second is a so-called "skin effect" mode, and the third is a so-called "slow wave" mode.

The higher the resistivity of the semiconductor layer, the closer the propagation mode is to a conventional TEM mode.

The third mode called "slow wave" appears when the operating frequency is low, of the order of 10 to 10³ MHz, and when the resistivity of the semiconductor layer is also low, of the order of 10⁻⁴ to 10² Ω.cm. In this "slow wave" mode, magnetic energy is distributed in the semiconductor layer, while electrical energy is stored in the dielectric layer. The sum of these energies is transmitted perpendicular to the layers, through the thin dielectric layer of silica (SiO₂). The phase speed therefore decreases due to the transfer of energy to the semiconductor-dielectric interface (Si / SiO₂).

The phase constant is expressed in terms of normalized wavelengths: λg / λ₀, a ratio which is equal to the speed of propagation in the line divided by the speed of light in a vacuum. The upper limit frequency strongly depends on the resistivity of the semiconductor layer and becomes maximum when the resistivity reaches 10⁻¹ Ω.cm, this frequency remaining less than GHz.

On the other hand, the phase constant and the characteristic impedance of the line are also very dependent on the transverse dimension of the ribbon, and on the thickness of the semiconductor + dielectric layers separating the ground plane of the ribbon.

In conclusion, this document teaches that the operation in slow wave mode has high losses which could be reduced by constructing a multilayer structure between the ground plane and the ribbon, this multilayer structure being formed by the alternation of semiconductor layers and of layers. thin dielectrics to reduce skin effect losses. If such a multilayer structure were used to make a microstrip line operating in slow wave mode, then the dimension of the line could be reduced, which would make it possible to reduce the dimensions of integrated circuits with the line operating in the frequency domain of the order of GHz or lower.

A technical problem which currently arises is the monolithic integration of microwave circuits on semi-insulating substrate. Indeed, if a microwave circuit is not integrated monolithically, it is less efficient due to losses in the links between substrates, it operates at lower frequencies due to the parasitic capacitances which appear, it shows a higher consumption, and it is more expensive because it requires larger surfaces of semi-insulating substrates, and more manufacturing steps.

However, the known transmission lines necessary for producing the microwave circuits, for example the microstrip lines operating in quasi-TEM mode, today occupy a large surface area on the substrates, making monolithic integration difficult, as soon as the circuit becomes complex.

The technical problem of monolithic integration of Microwave Integrated Circuits (MICs) can only be solved if the problem of miniaturization of transmission lines is solved beforehand, while taking into account that their realization must remain in synergy of manufacture with the other elements of the circuit, for example the transistors and the interconnection lines and taking into account that the losses in the lines must not increase and that the operating frequency must be that of the circuits microwave.

• le domaine de fréquences exploré est trop bas et non compatible avec les MMICs ;
• le substrat présente une résistivité trop faible qui n'est pas compatible avec la réalisation des autres éléments des circuits MMICs, ou qui au moins limite leurs performances ;
• la génération d'ondes lentes est très dépendante de la résistivité du substrat, ce qui entraîne que le dopage du substrat doit être très bien optimisé. Cette optimisation rend le procédé de réalisation d'un circuit incluant une telle ligne coûteux, avec néanmoins des risques de dispersion dans les performances ;
• le dispositif formé par la ligne nécessite un plan de masse à l'arrière du substrat, ce qui résulte en difficultés technologiques pour réaliser des interconnexions ;
• les pertes dans le fonctionnement en mode ondes lentes avec une seule couche semiconductrice sont très élevées ;
• si l'on veut diminuer les pertes, pour profiter de l'avantage présenté par les lignes ondes lentes relatif à la diminution de leur dimension, alors la technologie de fabrication du substrat incluant des couches alternées semiconductrices-diélectriques rend le dispositif encore plus difficile à réaliser, plus coûteux et moins compatible avec l'intégration monolithique.

However, the device known from the state of the art does not meet these requirements. Indeed: either it works in quasi TEM mode and in this case the dimensions of the lines are too large, or else it works in slow wave mode with the advantage of a significant phase shift and smaller dimensions, but in this case it has the following disadvantages, among others:

• the frequency range explored is too low and not compatible with MMICs;
• the substrate has too low a resistivity which is not compatible with the production of the other elements of the MMICs circuits, or which at least limits their performance;
• the generation of slow waves is very dependent on the resistivity of the substrate, which means that the doping of the substrate must be very well optimized. This optimization makes the process for producing a circuit including such a line expensive, with nevertheless risks of dispersion in performance;
• the device formed by the line requires a ground plane at the rear of the substrate, which results in technological difficulties in making interconnections;
• losses in slow wave operation with a single semiconductor layer are very high;
• if we want to reduce losses, to take advantage of the advantage presented by slow wave lines relating to the reduction in their size, then the technology for manufacturing the substrate including alternating semiconductor-dielectric layers makes the device even more difficult to more expensive and less compatible with monolithic integration.

It therefore follows from the teaching of the cited article that the lines operating in slow wave mode are attractive for the realization of integrated monolithic circuits because their dimensions could be minimized compared to these lines operating in TEM or quasi-TEM mode. conventional, but that on the other hand, their field of operation, their performance, and their production technology are incompatible with those required for MMIC circuits.

The object of the present invention is therefore to propose a transmission line in slow wave mode of the MICRORUBAN type, in which the propagation structure is fully compatible with integrated circuits, for example with microwave integrated circuits and in particular with MMICs.

To this end, an object of the invention is to propose a transmission line in slow wave mode of the MICRORUBAN type whose characteristics are independent of the characteristics of the substrate.

An object of the invention is to provide such a line devoid of ground plane on the rear face of the substrate.

An object of the invention is to propose such a line whose losses are not higher than those of microstrip lines operating in TEM or quasi-TEM conventional mode.

An object of the invention is to provide such a line, the dimensions of which are several times smaller than those of lines operating in conventional TEM or quasi-TEM mode, for identical line characteristics.

An object of the invention is to propose such a line capable of being associated with microwave circuits.

An object of the invention is to propose such a line, the production method of which is in complete synergy with the production methods of all conventional integrated circuits whatever the semiconductor substrate chosen for this circuit, without increasing the number of steps required. processes, and using only layers or materials used in said processes.

According to the invention, the problems are solved by means of a circuit as described in the preamble of claim 1, characterized in that the transmission line has, longitudinally, a periodic structure, each period, of length ℓ, being formed of a said bridge followed by a said pillar, in that each bridge consists of a section of the upper conductive tape, of length ℓ₁ <ℓ, disposed on the surface of said first part of the third material, which is of dielectric nature , and that each pillar is a capacity.

The line according to the invention can then be included in an MMIC circuit with all the advantages already mentioned which result therefrom.

Another object of the invention is to provide a slow wave transmission line, the principle of which is based on such a periodic structure, the dimensions of which are further reduced and the performance of which is also improved, all by simply changing the design. in the step of drawing the masks of the integrated circuit.

This object is achieved by means of the aforementioned line, further characterized in that the first conductive layer serving as ground plane has at least one recess respectively under each bridge.

This line has the property of having a higher deceleration than the previous line at equal frequency. This property makes it possible to produce, for the same application, even shorter lines, therefore even more easily integrated. When we know the problems linked to the integration of microwave lines, this result constitutes a first-rate industrial advantage, without any great additional technological difficulty.

On the other hand, the transmission line obtained being shorter, the losses are reduced compared to those which occur in the known line.

Another object of the invention is to provide a coupler of the so-called Lange coupler type which is easily integrated, and in particular which is synergistically produced with current microwave integrated circuits, and whose performance is also improved compared to that we can expect known devices.

A Lange coupler is known to those skilled in the art by the publication "Integrated Stripline quadrature Hybrids", IEEE, MTT, Dec. 1969, pp.1150-1151.

This coupler is produced in microstrip technology, that is to say by means of microstrip conductors arranged on a first face of a substrate of given thickness, the second face of which receives the ground plane. Therefore, by this production method, this coupler is not fully compatible with current integrated circuit technologies.

This known coupler consists of an odd number, that is to say at least 3, of parallel transmission lines, connected 2 to 2 alternately to form an interdigitated structure. The middle line is called the line main, and the coupler is completely symmetrical about the middle of the main line. In particular, its inputs and outputs are symmetrical.

The length L of the main line defines the operating frequency band of this coupler. This length L is of the order of a quarter of the wavelength λ of the transported signal.

The operation of the Lange coupler is based on the following principle: an electromagnetic field coupling is formed between the parallel lines. This coupling is of the capacitive or inductive type depending on the relationships between the length L of the main line and the wavelength λ of the signals which propagate in the coupler.

le couplage est capacitif,
   si $\text{λ/4 = L}$

le couplage est à la fois capacitif et inductif,
   si $\text{λ/4 > L}$

le couplage est inductif.Yes $\text{λ / 4 <L}$

the coupling is capacitive,
if $\text{λ / 4 = L}$

the coupling is both capacitive and inductive,
if $\text{λ / 4> L}$

the coupling is inductive.

.On the other hand, there is a phase shift Δφ between the signals carried by the two outputs. This phase shift Δφ is equivalent to 90 ° in a frequency band centered on that where $\text{λ = 4L}$

.

Because the operating wavelength is linked to the dimensions of the coupler, it seemed a priori impossible to modify these dimensions, for a given wavelength, and in a chosen technology.

However, as we have seen previously, the designer of integrated circuits poses the problem of the ever-deeper reduction of the dimensions of the components, in order to achieve a greater integration density.

One of the aims of the invention is therefore to provide a Lange coupler whose design is compact and whose dimensions are minimized compared to those of known devices.

These aims are achieved when a Lange coupler is produced by means of transmission lines of the aforementioned slow wave type.

In addition, an integrated duplexer, or active duplexer, is known from the publication entitled: "Distributed amplifiers as duplexer / low cross talk bidirectional element in S band" by OP LEISTEN, RJCOLLIER AND RNBATES in "Electronics Letters March 3, 1988 , Vol.24, N ° 5, p.264-265 ".

It must first be recalled that the technical problem which arises for a person skilled in the art who wishes to use only one antenna for the transmission and for the reception of two signals at different frequencies, with different amplitudes, is the realization of a signal splitter, also called duplexer, which makes it possible to avoid crosstalk, that is to say the intermodulation of the signals transmitted and received.

Another technical problem which arises to those skilled in the art is the production of such a duplexer in integrated form. The resolution of these problems makes it possible to reduce manufacturing costs, which is an important advantage in particular in the fields of consumer products, such as the field of television or automotive electronics for example.

It is known from the document cited above that the problem of the separation of the transmitted and received signals can be solved by an active duplexer constituted by an integrable distributed amplifier working at microwave frequencies.

• il est certes intégrable mais il occupe une surface importante ; bien que cette surface puisse être diminuée lorsque le circuit est conçu pour travailler dans le domaine des hyperfréquences (60 GHz), elle est cependant considérée par le concepteur de circuit intégré comme trop importante dans tous les cas ;
• ce circuit est complexe à réaliser ;
• la diaphonie due à ce circuit est encore trop importante ; notamment elle est beaucoup plus importante que celle de circulateurs hybrides non intégrables ; en effet l'intermodulation est due, dans le circuit décrit dans le document cité, à la non linéarité des éléments actifs ;
• ce circuit est bruyant.

However, the distributed amplifier described in the cited publication has some drawbacks:

• it is certainly integrable but it occupies an important surface; although this surface can be reduced when the circuit is designed to work in the microwave domain (60 GHz), it is nevertheless considered by the integrated circuit designer to be too large in all cases;
• this circuit is complex to perform;
• the crosstalk due to this circuit is still too great; in particular it is much more important than that of non-integrable hybrid circulators; indeed intermodulation is due, in the circuit described in the cited document, to the non-linearity of the active elements;
• this circuit is noisy.

These problems are solved according to the invention by a transceiver device including an integrated circuit comprising a frequency duplexer for transmitting a first signal and receiving a second signal on a single pole, characterized in that the integrated frequency duplexer is a coupler directional of the aforementioned type, having two said first poles connected by electromagnetic coupling to two said second poles, in that one of said first poles constitutes an input for the first signal from a first amplifier, and the other says first an output for the second signal, which propagates towards the input of a second amplifier, and in that one of the said second poles constitutes an output for the first signal and an input for the second signal and the other said second poles is isolated.

• le duplexeur de fréquences, nécessaire à son fonctionnement, est intégrable, avec une surface occupée très inférieure à celle de l'amplificateur distribué connu ;
• la diaphonie est quasiment nulle ;
• le bruit est minimisé.

The transceiver device according to the invention then has the following advantages:

• the frequency duplexer, necessary for its operation, can be integrated, with an occupied surface area much smaller than that of the known distributed amplifier;
• the crosstalk is almost zero;
• noise is minimized.

• la figure 1a qui représente une ligne de transmission en mode ondes lentes de type microruban, vu du dessus, dans l'exemple I de réalisation ;
• la figure 1b qui représente une telle ligne de transmission dans l'exemple II de réalisation ;
• la figure 1c qui représente une telle ligne de transmission dans l'exemple V de réalisation ;
• la figure 1d qui représente une telle ligne de transmission vue du dessus dans l'exemple VI de réalisation ;
• la figure 2a qui représente la ligne de la figure 1a en coupe transversale selon l'axe A-A' de cette figure 1a ;
• la figure 2b qui représente la ligne de la figure 1a en coupe longitudinale selon l'axe B-B' de cette figure 1a ;
• la figure 2c qui représente la ligne de la figure 1a en coupe transversale selon l'axe C-C' de cette figure 1a ;
• la figure 3 qui représente le schéma équivalent à une ligne selon la figure 1 ;
• la figure 4 qui représente le facteur de ralentissement (ou d'ondes lentes) λ₀/λ_g en fonction de la fréquence F de propagation exprimée en GHz dans l'exemple I de réalisation ;
• la figure 5 qui représente d'une part la partie réelle Re de l'impédance caractéristique Z_c de la ligne, et d'autre part la partie imaginaire Im de cette impédance, dans l'exemple I, et en fonction de la fréquence F en GHz ;
• la figure 6 qui représente d'une part les pertes α en dB/cm en fonction de la fréquence F en GHz, et d'autre part les pertes α' en dB par rapport à la longueur d'onde λ_g en fonction de ladite fréquence F ;
• la figure 7 qui représente la ligne de la figure 1b en coupe longitudinale selon l'axe B-B' de cette figure 1b dans l'exemple II ;
• la figure 8 qui représente la ligne décrite dans l'exemple III en coupe longitudinale ;
• la figure 9 qui représente la ligne décrite dans l'exemple IV en coupe longitudinale ;
• la figure 10 qui représente le facteur de ralentissement (ou d'ondes lentes) λ₀/λ_g en fonction de la fréquence F de propagation exprimée en GHz dans l'exemple V de réalisation ;
• la figure 11 qui représente la ligne décrite dans l'exemple VI selon la coupe CC' de la figure 1d.
• la figure 12 qui représente, vue du dessus, schématiquement une ligne coplanaire connectée à une ligne ondes lentes selon l'invention ;
• la figure 13 qui représente à titre exemplatif un circuit utilisant le dispositif de la figure 12 ;
• la figure 14a montre une ligne à ondes lentes vue du dessus dans l'exemple de réalisation X ;
• la figure 14b montre cette ligne en coupe grossie, selon l'axe BB' de la figure 14a ;
• la figure 14c montre une ligne à ondes lentes vue du dessus, dans l'exemple XI de réalisation ;
• la figure 14d montre cette ligne en coupe grossie, selon l'axe BB' de la figure 14c ;
• la figure 14e montre la ligne de la figure 15a, ou bien de la figure 14c, en coupe selon l'axe AA' ;
• la figure 15a montre deix cpirbes rerésentatives du facteur de ralentissement R de lignes hyperfréquences, en fonction de la fréquence F, la courbe A concernant une ligne microruban selon la figure 1a sans évidements sous les ponts, et la seconde B concernant une ligne microruban munie d'évidements dans le plan de masse sous les ponts, tels que par exemple montrée sur les figures 14a ou 14c ;
• la figure 15b montre 3 courbes représentatives du facteur de ralentissement R d'une ligne hyperfréquences correspondant au type de la figure 15a, en fonction de la fréquence F, et pour différentes valeurs du paramètre constitué par la hauteur e₁ de diélectrique 1 sous les ponts, la courbe C correspondant à e₁ = 2 µm, la courbe D à e₁ = 2,4 µm et la courbe E à e₁ = 2,8 µm ;
• la figure 15c montre 3 courbes représentatives du facteur de ralentissement R d'une ligne hyperfréquences conforme au type de la figure 15a, en fonction de la période ℓ, pour différentes valeurs du paramètre constitué par le rapport ℓ₁/ℓ₂ où ℓ₁ est la longueur des ponts et ℓ₂ la longueur des piliers, à une valeur fixe de la fréquence F = 12 GHz ;
• la figure 16a montre un coupleur de Lange représenté schématiquement ;
• la figure 16b représente un coupleur de Lange vu du dessus, réalisé au moyen de lignes conformes à celles de la figure 15a, dans une technologie de circuit intégré ;
• la figure 16c représente une partie agrandie d'un tel coupleur réalisé selon un premier exemple de mise en oeuvre ;
• la figure 16d représente une partie agrandie d'un tel coupleur lorsqu'il est réalisé selon un second exemple de mise en oeuvre ;
• la figure 17 représente deux courbes, l'une K du coefficient de couplage en dB en fonction de la fréquence F et l'autre M du coefficient d'accord en dB en fonction de la fréquence pour un coupleur du type de la figure 16b.
• la figure 18 qui représente schématiquement un dispositif émetteur-récepteur à une seule antenne ;
• la figure 19 qui représente schématiquement un dispositif émetteur-récepteur muni d'un coupleur de Lange ;
• la figure 20 qui représente un coupleur à branches ;
• la figure 21 qui représente un circuit de tête hyperfréquence de module réception-émission d'un radar ;
     De nombreuses variantes de la ligne ondes lentes selon l'invention sont possibles. Toutes ces variantes ont en commun les éléments essentiels de l'invention qui vont être mis en lumière dans la description d'un premier exemple de réalisation, choisi parmi d'autres pour sa simplicité.

The invention is described below in detail, with reference to the appended schematic figures among which:

• FIG. 1a which represents a transmission line in slow wave mode of microstrip type, seen from above, in Example I of embodiment;
• Figure 1b which shows such a transmission line in the embodiment II;
• Figure 1c which shows such a transmission line in the embodiment V;
• Figure 1d which shows such a transmission line seen from above in the exemplary embodiment VI;
• Figure 2a which shows the line of Figure 1a in cross section along the axis AA 'of this Figure 1a;
• Figure 2b which represents the line of Figure 1a in longitudinal section along the axis BB 'of this Figure 1a;
• Figure 2c which shows the line of Figure 1a in cross section along the axis CC 'of this Figure 1a;
• FIG. 3 which represents the diagram equivalent to a line according to FIG. 1;
• FIG. 4 which represents the deceleration factor (or slow wave) λ₀ / λ _g as a function of the propagation frequency F expressed in GHz in example I of embodiment;
• FIG. 5 which represents on the one hand the real part Re of the characteristic impedance Z _c of the line, and on the other hand the imaginary part Im of this impedance, in example I, and as a function of the frequency F in GHz;
• FIG. 6 which represents on the one hand the losses α in dB / cm as a function of the frequency F in GHz, and on the other hand the losses α ′ in dB relative to the wavelength λ _g as a function of said frequency F;
• Figure 7 which shows the line of Figure 1b in longitudinal section along the axis BB 'of this Figure 1b in Example II;
• Figure 8 which shows the line described in Example III in longitudinal section;
• FIG. 9 which represents the line described in Example IV in longitudinal section;
• FIG. 10 which represents the deceleration factor (or slow wave) λ₀ / λ _g as a function of the propagation frequency F expressed in GHz in the embodiment V;
• FIG. 11 which represents the line described in Example VI according to section CC 'of FIG. 1d.
• FIG. 12 which represents, seen from above, schematically a coplanar line connected to a slow wave line according to the invention;
• Figure 13 which shows by way of example a circuit using the device of Figure 12;
• FIG. 14a shows a slow wave line seen from above in the embodiment X;
• Figure 14b shows this line in enlarged section, along the axis BB 'of Figure 14a;
• FIG. 14c shows a slow wave line seen from above, in exemplary embodiment XI;
• Figure 14d shows this line in enlarged section, along the axis BB 'of Figure 14c;
• Figure 14e shows the line of Figure 15a, or of Figure 14c, in section along the axis AA ';
• FIG. 15a shows ten cpirbs representing the deceleration factor R of microwave lines, as a function of frequency F, curve A relating to a microstrip line according to FIG. 1a without recesses under the bridges, and the second B relating to a microstrip line provided with 'recesses in the ground plane under the bridges, such as for example shown in Figures 14a or 14c;
• FIG. 15b shows 3 curves representative of the deceleration factor R of a microwave line corresponding to the type of FIG. 15a, as a function of the frequency F, and for different values of the parameter constituted by the height e₁ of dielectric 1 under the bridges, curve C corresponding to e₁ = 2 µm, curve D at e₁ = 2.4 µm and curve E at e₁ = 2.8 µm;
• FIG. 15c shows 3 curves representative of the deceleration factor R of a microwave line conforming to the type of FIG. 15a, as a function of the period ℓ, for different values of the parameter constituted by the ratio ℓ₁ / where ℓ₁ is the length of the bridges and ℓ₂ the length of the pillars, at a fixed value of the frequency F = 12 GHz;
• Figure 16a shows a Lange coupler shown schematically;
• FIG. 16b represents a Lange coupler seen from above, produced by means of lines conforming to those of FIG. 15a, in an integrated circuit technology;
• FIG. 16c represents an enlarged part of such a coupler produced according to a first example of implementation;
• FIG. 16d represents an enlarged part of such a coupler when it is produced according to a second example of implementation;
• FIG. 17 shows two curves, one K of the coupling coefficient in dB as a function of frequency F and the other M of the tuning coefficient in dB as a function of frequency for a coupler of the type of FIG. 16b.
• FIG. 18 which schematically represents a transmitter-receiver device with a single antenna;
• FIG. 19 which schematically represents a transmitter-receiver device provided with a Lange coupler;
• FIG. 20 which represents a branch coupler;
• FIG. 21 which represents a microwave head circuit of a radar reception / emission module;
  Many variants of the slow wave line according to the invention are possible. All these variants have in common the essential elements of the invention which will be brought to light in the description of a first exemplary embodiment, chosen from among others for its simplicity.

EXAMPLE I

This exemplary embodiment is illustrated by FIGS. 1 and 2 to 6.

Figure la shows a slow wave line seen from the above, of MICRORUBAN structure.

This line is produced on a substrate 10 which can be of any material whatsoever. For example: completely insulating, fully conducting, semi-insulating or semiconducting; this unlimited choice of materials for producing the substrate makes it possible to apply the invention to all kinds of circuits, in all conceivable technologies, when the circuit comprises a transmission line.

• une couche conductrice 11, par exemple en un métal bon conducteur pouvant faire office de plan de masse de dimension transversale W1 ;
• une couche diélectrique 2, de permitivité relative ε_r2 et d'épaisseur e₂, de longueur totale au moins égale à celle de la couche 11, et de dimension transversale W₃ ;
• un ruban en un matériau conducteur, par exemple un métal bon conducteur 12 ; ce ruban 12, de faible dimension transversale W₂, forme avec les couches précédentes une structure périodique, de périodicité ℓ ; à cet effet, le ruban conducteur 12 comprend des parties 3 en contact avec la couche diélectrique 2, ces parties 3 étant de dimension longitudinale ℓ₂ (parallèlement à l'axe BB'), et des parties 4 suspendues entre deux parties 3, ces parties 4 suspendues ayant une dimension longitudinale ℓ₁ (parallèlement à l'axe B-B'), de sorte que :
  
  $\text{ℓ = ℓ₁ + ℓ₂ ;}$
  
  
  
  
• les dimensions transversales des couches 11, 2, 12, sont telles que : W₂ ≦ W₃ ≦ W₁

On the substrate 10, the line comprises the succession of:

• a conductive layer 11, for example of a good conductive metal which can act as a ground plane of transverse dimension W1;
• a dielectric layer 2, of relative permitivity ε _r2 and of thickness e₂, of total length at least equal to that of layer 11, and of transverse dimension W₃;
• a ribbon made of a conductive material, for example a good conductive metal 12; this ribbon 12, of small transverse dimension W₂, forms with the preceding layers a periodic structure, of periodicity ℓ; for this purpose, the conductive tape 12 comprises parts 3 in contact with the dielectric layer 2, these parts 3 being of longitudinal dimension ℓ₂ (parallel to the axis BB '), and parts 4 suspended between two parts 3, these parts 4 suspended with a longitudinal dimension ℓ₁ (parallel to the axis B-B '), so that:
  
  $\text{ℓ = ℓ₁ + ℓ₂;}$
  
  
  
  
• the transverse dimensions of the layers 11, 2, 12, are such that: W₂ ≦ W₃ ≦ W₁

Figure 2b shows a longitudinal section along the axis BB 'of the line of Figure 1a. This figure shows that, in example I, to make the contact of the parts 3 of the tape 12 with the dielectric layer 2, the tape 12 is collapsed at the level of the parts 3. On the contrary, in the suspended parts 4, the strip 12 is raised by a height e₁ with respect to the upper surface of the dielectric layer 2.

The hanging parts 4 are the parts in which the propagation takes place. In these parts, the strip 12 is suspended above a dielectric 1, of relative permissiveness ε _r1 .

• PONTS les parties 4 du ruban 12 suspendues au-dessus du diélectrique 1, les ponts 4 ayant une longueur ℓ₁, et constituant les régions de propagation ;
• PILIERS les parties 13 formées de l'associationde la couche conductrice inférieure 12, de la couche diélectrique 2 d'épaisseur e₂ et des parties 3 du ruban 12, les piliers 13 formant une structure MIM (métal-isolant-métal) de longueur ℓ₂.

For reasons of language simplification we will call below:

• BRIDGES the parts 4 of the ribbon 12 suspended above the dielectric 1, the bridges 4 having a length ℓ₁, and constituting the propagation regions;
• PILLARS the parts 13 formed from the combination of the lower conductive layer 12, the dielectric layer 2 of thickness e₂ and the parts 3 of the tape 12, the pillars 13 forming a MIM (metal-insulator-metal) structure of length ℓ₂.

Figure 2a shows a cross section of the line along the axis AA 'of Figure 1a, at a bridge 4, and Figure 2c shows a cross section of the line along the axis CC' of Figure 1a , at the level of a pillar 13.

• une structure de ligne MICRORUBAN comprenant une couche conductrice inférieure 11, un ruban conducteur supérieur 12 et une partie intermédiaire diélectrique 1, 2 ;
• le fait que cette structure est périodique, de période ℓ, formée de PONTS 4 suspendus sur un premier diélectrique 1, de permitivité relative ε_r1, de longueur ℓ₁, ces ponts dans lesquels se fait la propagation de l'onde étant disposés entre deux PILIERS 13 formés d'une structure capacitive (dans cet exemple I, la structure capacitive est une structure MIM constituée de la couche inférieure conductrice 11, de la couche diélectrique 2, de permitivité ε_r2 et du ruban conducteur 12, les piliers ayant une longueur ℓ₂ telle que $\text{ℓ₂ + ℓ₁ = ℓ}$
  
  ) ;
• les valeurs des paramètres : ε_r1, ε_r2, ℓ, ℓ₁, e₁, la valeur de la capacité et W₁, W₂ de la structure de la ligne sont liées entre elles pour résulter en la propagation d'ondes lentes et fournir un déphasage important sur une longueur totale Δ de ligne de transmission courte. (Dans l'exemple I, la valeur de la capacité est liée à ℓ₁ et e₂).

From this exemplary embodiment I, it appears that the essential elements for producing a slow wave line reside in:

• a MICRORUBAN line structure comprising a lower conductive layer 11, an upper conductive tape 12 and a dielectric intermediate part 1, 2;
• the fact that this structure is periodic, of period ℓ, formed of BRIDGES 4 suspended on a first dielectric 1, of relative permitivity ε _r1 , of length ℓ₁, these bridges in which the wave propagation takes place being placed between two PILLARS 13 formed of a capacitive structure (in this example I, the capacitive structure is a MIM structure consisting of the lower layer conductor 11, the dielectric layer 2, with permissiveness ε _r2 and the conductive tape 12, the pillars having a length ℓ₂ such that $\text{ℓ₂ + ℓ₁ = ℓ}$
  
  );
• the values of the parameters: ε _r1 , ε _r2 , ℓ, ℓ₁, e₁, the value of the capacitance and W₁, W₂ of the structure of the line are linked together to result in the propagation of slow waves and provide a significant phase shift over a total length Δ of short transmission line. (In example I, the value of the capacity is linked to ℓ₁ and e₂).

• le pas ℓ de la structure périodique peut être constant ou non. On décrira plus loin un exemple de réalisation de ligne à pas non constant ;
• le matériau choisi pour réaliser le substrat n'a aucune influence sur le fonctionnement de la ligne ; le substrat ne sert que de support ;
• le dessin de la ligne peut être linéaire, en méandre, en spirale ; tout autre dessin imaginable est possible.
• la capacitè peut être un élément passif ou actif.
• en outre la couche diélectrique de la structure MIM peut être éventuellement formée de deux couches diélectriques superposées (2a, 2b). Ce genre de structure a deux couches diélectriques est connue de l'homme de métier et n'est donc pas représentée sur les dessins.

Apart from these essential elements:

• the step ℓ of the periodic structure can be constant or not. An exemplary embodiment of a line with non-constant pitch will be described later;
• the material chosen to make the substrate has no influence on the operation of the line; the substrate only serves as a support;
• the line drawing can be linear, meandering, spiral; any other design imaginable is possible.
• the capacity can be a passive or active element.
• in addition, the dielectric layer of the MIM structure can optionally be formed of two superimposed dielectric layers (2a, 2b). This type of structure with two dielectric layers is known to those skilled in the art and is therefore not shown in the drawings.

It is these characteristics which lead to numerous variants of the slow wave transmission line, which are particularly easy to produce, particularly efficient and, inter alia, especially applicable to the production of MMIC circuits.

Indeed, the slow wave operation of the line, producing significant phase shifts over a short length Δ results in the fact that these lines are more easily integrated than the known MICRORUBANS lines.

In order to evaluate the performance of such a line, it is necessary to evaluate the propagation constant γ in the length ℓ of the line.

   Z₁, Z₂ les impédances caractéristiques respectivement dans les parties PONTS 4 et PILIERS 13.
La constante de propagation γ est liée aux pertes α dans la ligne et à la constante de phase β par la relation :

$\text{γ = α + jβ.}$

La constante de phase β dans la ligne est liée à la longueur d'onde λ_g de propagation dans la ligne par la relation :

${\text{β = 2π/λ}}_{\text{g}}$

La permitivité effective ε_reff est liée à la longueur d'onde normalisée λ_g/λ₀ déjà définie précédemment par :
${\text{ ε}}_{\text{reff}}{\text{= (λ₀/λ}}_{\text{g}}\text{)² = (1/R)²}$

où R est le facteur d'ondes lentes.We will call hereafter:
γ₁, γ₂ the propagation constants respectively in the BRIDGE part 4, and in the PILLAR part 13.
ℓ₁, ℓ₂ the BRIDGE, PILLAR lengths already defined as $\text{ℓ₁ + ℓ₂ = ℓ}$

Z₁, Z₂ the characteristic impedances respectively in the BRIDGES 4 and PILLARS 13 parts.
The propagation constant γ is linked to the losses α in the line and to the phase constant β by the relation:

$\text{γ = α + jβ.}$

The phase constant β in the line is linked to the wavelength λ _g of propagation in the line by the relation:

${\text{β = 2π / λ}}_{\text{g}}$

The effective permitivity ε _reff is linked to the normalized wavelength λ _g / λ₀ already defined previously by:
${\text{ε}}_{\text{reff}}{\text{= (λ₀ / λ}}_{\text{g}}\text{) ² = (1 / R) ²}$

where R is the slow wave factor.

FIG. 3 represents the equivalent diagram of a unit cell of the line, that is to say comprising a half BRIDGE, a PILLAR and a second half BRIDGE.

   et $\text{ϑ₂ = γ₂ ℓ₂}$

D'autre part, B est la susceptance de la discontinuité entre le PONT 4 sur le diélectrique 1 et le PILIER 13 MIM.We define $\text{ϑ₁ = γ₁ ℓ₁}$

and $\text{ϑ₂ = γ₂ ℓ₂}$

On the other hand, B is the susceptance of the discontinuity between BRIDGE 4 on dielectric 1 and PILLAR 13 MIM.

où $\text{K± = (1±K)}$

avec $\text{K = Z₂/Z₁ + Z₁/Z₂ = B² Z₂Z₁}$

Using a classical calculation method applicable to periodic structures, the propagation constant γ is linked to the other parameters of the line defined previously for the unit cell of the equivalent diagram in Figure 3, by the relation:

$\text{ch (γ.ℓ) = {K⁺ ch (ϑ₁ + ϑ₂) + K⁻h (ϑ₁-ϑ₂) - B / 2 (Z₁ + Z₂) sh (ϑ₁ + ϑ₂) - B / 2 (Z₁-Z₂) sh (ϑ₁-ϑ₂)}}$

or $\text{K ± = (1 ± K)}$

with $\text{K = Z₂ / Z₁ + Z₁ / Z₂ = B² Z₂Z₁}$

This relation allows the calculation of the phase constant β. It follows from these calculations that by choosing:
ℓ₁, ℓ₂
ε _r1 , ε _r2
e₁ and e₂
W₁ and W₂
appropriately, the phase speed of the line is low. Hence the existence of the so-called slow wave mode.

• le substrat 10 est semi-isolant de manière à intégrer la ligne dans un circuit MMIC,
• le diélectrique 1 sous les PONTS 4 est l'air de permitivité relative ε_r1 = 1
• le diélectrique 2 dans les piliers 13 de structure MIM est choisi entre la silice (SiO₂) et le nitrure de silicium (Si₃N₄) ; dans ces conditions, la permitivité relative de la couche diélectrique 2 a une valeur de l'ordre de 6 pour la silice (SiO₂) et une valeur de l'ordre de 7 pour le nitrure de silicium (Si₃N₄) ; on réalisera ces couches 2 dans les conditions technologiques très strictes, propres aux circuits intégrés, de manière à obtenir pour les permitivités ε_r2 ces valeurs élevées ; si les conditions technologiques sont moins strictes, les valeurs peuvent être moins élevées, de l'ordre de 4 ;
• les couches conductrices 11 et 12 sont choisies parmi les métaux qui constituent d'ordinaire le premier niveau d'interconnexion d'un circuit intégré pour la couche conductrice inférieure 11, et le second niveau d'interconnexion d'un circuit intégré pour la couche conductrice supérieure 12 formant le ruban.

To meet the conditions established by these calculations, in this example I, a slow wave line was produced where:

• the substrate 10 is semi-insulating so as to integrate the line into an MMIC circuit,
• the dielectric 1 under the BRIDGES 4 is the air of relative permissiveness ε _r1 = 1
• the dielectric 2 in the pillars 13 of MIM structure is chosen between silica (SiO₂) and silicon nitride (Si₃N₄); under these conditions, the relative permeability of the dielectric layer 2 has a value of the order of 6 for silica (SiO₂) and a value of the order of 7 for silicon nitride (Si₃N₄); these layers 2 will be produced under very strict technological conditions, specific to integrated circuits, so as to obtain these high values for the permitivities ε _r2 ; if the technological conditions are less strict, the values may be lower, of the order of 4;
• the conductive layers 11 and 12 are chosen from the metals which usually constitute the first level of interconnection of an integrated circuit for the lower conductive layer 11, and the second level of interconnection of an integrated circuit for the conductive layer upper 12 forming the ribbon.

Thus, in this exemplary embodiment I, the line is in complete manufacturing synergy with an integrated circuit MMIC.

However, it is obvious that other choices can be made for the materials.

Table I below brings together the preferred values of the parameters for implementing the line in this example I.

FIG. 1a also shows that the dielectric 2 has a length slightly greater than that of the ground plane 11 (which can be connected to ground by studs 21) to allow the realization of an input E by a stud 22a, and of an output O of the slow wave line by a pad 22b.

Figures 4, 5 and 6 give curves showing the performance of a line, obtained under the conditions where the elements of the line have the values given in table I.

Figure 4 shows the slow wave factor λ₀ / λ _g as a function of frequency F in GHz. From this figure, we deduce that the relative effective permissiveness ε _reff is very high at low frequencies, frequencies for example less than 4 GHz, then remains almost constant between 4 and 20 GHz, with a value of the order of 20. This value must be compared with effective relative permissivity values known to a person skilled in the art for conventional MICRORUBANS lines, which are of the order of 6 to 8 when the line is made on alumina (Al₂O₃) or on a semiconductor.

FIG. 5 represents the real and imaginary parts, respectively Re (Z _c ), and Im (Z _c ), of the characteristic impedance Z _c of this line. The real part of the impedance Z _c is extremely small. This line according to Example I will therefore find very interesting applications in the production of a low impedance line for an impedance transformer.

FIG. 6 shows on the one hand the losses α in the line, expressed in dB / cm, as a function of the frequency F in GHz, and on the other hand the losses α ′ in dB relative to the wavelength λg as a function of said frequency F. These losses per cm are slightly higher than those of a conventional MICRORUBAN line.

However, because the phase speed is low, the slow wave line has a total length Δ reduced by approximately 2 times compared to a conventional MICRORUBAN line. As a result, the performance of the slow wave line is not deteriorated compared to a conventional MICRORUBAN line, while on the contrary it has the advantage of being shorter, and therefore more easily integrated.

EXAMPLE II

This example is illustrated by FIG. 1b seen from above and by FIG. 7 which is a section along the axis BB 'of FIG. 1b.

In previous Example I, the dielectric layer 2 was continuous from one end to the other of the line. In this example II, on the other hand, the layer 2 is eliminated under the BRIDGES. However, it is essential for producing the MIM structure of the pillars 13. In fact, it was considered that, in example I, its influence under bridges 4 was negligible.

EXAMPLE III

This example is illustrated by FIG. 1b and by FIG. 8.

The slow wave line does not show any changes in the schematic representation seen from above and can therefore be illustrated by FIG. 1b.

Figure 8 is a section along the axis B-B 'of Figure 1b in this embodiment. According to the section of FIG. 8, the dielectric 2 of the MIM structure of the pillars 13 has the same thickness as the dielectric 1 placed under the bridges 4. On the other hand, the layer of dielectric 2 which could remain under the BRIDGES 4 in the example I, should be excluded in this example III, as the possibility was shown in example II.

les autres paramètres vont varier considérablement par rapport à ceux qui sont présentés dans le tableau I. Plus particulièrement, les rapports des longueurs ℓ₁ et ℓ₂ seront très différents. Par contre, les permitivités respectivement ε_r1 et ε_r2 peuvent être les mêmes que dans l'exemple I, et par conséquent les diélectriques 1 et 2 peuvent être identiques à ceux de cet exemple.To obtain operation in slow wave mode, because we have chosen here:

$\text{e₁ = e₂}$

the other parameters will vary considerably compared to those presented in table I. More particularly, the ratios of the lengths ℓ₁ and ℓ₂ will be very different. On the other hand, the permitivities respectively ε _r1 and ε _r2 can be the same as in example I, and consequently the dielectrics 1 and 2 can be identical to those of this example.

EXAMPLE IV

This example can be illustrated by FIG. 1a, seen from above and by FIG. 9.

The slow wave line does not show any change in the schematic representation of FIG. 1a seen from above.

.Figure 9 is a section along the axis BB 'of Figure 1a in this embodiment. According to the section of FIG. 9, the dielectric 1 and the dielectric 2 are produced using the same material and therefore have the same relative permissiveness: ${\text{ε}}_{\text{r1}}{\text{= ε}}_{\text{r2}}$

.

To obtain operation in slow wave mode, the other parameters of the line are then very different from those whose values are given in table I.

More particularly, the ratios between the thicknesses e₁ and e₂, the ratios between the lengths ℓ₁ and ℓ₂ will be very different.

EXAMPLE V

This example is illustrated by Figures 1c and 10.

In all the previous examples, the curve in FIG. 4, representing the deceleration factor, could remain substantially valid, by adjusting the values of the different parameters.

As it was sought, in all cases a constant slowing factor was obtained in the medium and microwave frequencies (4 to 20 GHz). This resulted in a variation in phase shift β as a function of frequency F.

On the contrary, by means of the slow wave line produced according to the principle of the invention in example V, it is possible to obtain a phase shift β which remains constant as a function of the wavelength. It suffices for this to produce a slow wave line structure in which the slowing factor λ₀ / λ _g varies, for example this slowing factor showing a growth which approaches a hyperbolic form, as shown on the curve of the figure. 10.

deviendra sensiblement constant en fonction de la fréquence F,dans la bande de fréquences 4 à 20 GHz.Under these conditions, the phase shift ${\text{β = 2π / λ}}_{\text{g}}$

will become substantially constant as a function of frequency F, in the frequency band 4 to 20 GHz.

This result is obtained by means of the slow wave line structure shown diagrammatically seen from above in FIG. 1c.

The main characteristic of this line is that the periodicity ℓ shows a growth and in particular a geometric growth. The growth factor can be included between 1 (1 being not included since we would then be in the case of the previous examples) and approximately 3.

As regards the technology proper of such a line of periodicity ℓ not constant, the person skilled in the art can preferably adopt that of Example I which is particularly easy to implement. But nothing prevents the creation of new variants by applying to this example V the teaching drawn from examples II to IV.

EXAMPLE VI

This example is illustrated by FIG. 1d seen from above and by FIG. 11.

In the previous example, a person skilled in the art had the possibility of acting on the phase shift β by implementing a particular structure of the slow wave line.

In this example VI, a structure is proposed giving the possibility of acting electronically on said phase shift β.

As shown seen from above in FIG. 1d, the conductive layer 11 itself has a periodic structure, of period ℓ. In the regions 13 ′ corresponding to the PILLARS 13 in FIG. 1a for example, a diode 13 ′ polarized by a DC bias voltage V _{DD has been produced} which can have different values.

In example VI, the DIODE 13 ′ is more conveniently a field effect transistor with a Schottky gate, whose source S and the drain D short-circuited are brought to the DC bias voltage V _DD and whose gate G is brought to ground M. Obviously, in the region of the transistor or DIODE 13 ', the substrate 10 is no longer arbitrary, as in the previous examples, but must include an active area 10a, of a semiconductor material, for example conductivity type N, the rest of the substrate 10b on either side of the active layer 10a being semi-insulating. Regions 10a and 10b can be layers of material chosen from semiconductors such as: silicon (Si) or gallium arsenide (GaAs) for example. The Schottky gate transistor 13 ′ is produced for example as follows:
A semi-insulating layer 10b and regions 10a called active zones are produced by any means known to those skilled in the art of integrated circuits. The active zones 10a are produced with a periodicity ℓ chosen for the slow wave line. The active areas 10a must have the dimensions necessary and sufficient to receive a Schottky gate field effect transistor. This technology is known to anyone skilled in the art of integrated circuits.

The conductive layer 11 is then produced. outside the active regions 10a, the conductive layer 11, the material of which is preferably chosen from metals capable of forming a Schottky grid, has the transverse dimension W₁ determined as in the previous examples.

In the active regions 10a, the metal layer 11 is on the other hand narrowed (see FIG. 1d).
Longitudinally, along the axis BB 'of FIG. 1d, it has a dimension known as the gate width of the Schottky transistor and perpendicular to the axis BB', it has a small dimension of the order of µm known as the gate length of the transistor Schottky. Then ohmic contacts of a material 14 forming source pads S and drain D are disposed on either side of the gate G according to a conventional scheme of field effect transistor with Schottky gate. The Schottky gate transistor 13 ′ is illustrated in FIG. 11 in section along the axis CC ′ in FIG. 1d.

The ribbon 12 is then produced, showing bridges 4 in the regions of the metal layer 12, where the latter has the dimension W₁.

To make the electrical contacts between the strip 12 and the ohmic contacts 14 of source S and of drain D of each field effect transistor 13 ', in a particularly interesting embodiment, the ribbon 12 is divided into two parts 12a and 12b, the part 12a coming to establish the surface contact of the ohmic contact of source S, and the part 12b coming to establish the contact on the surface of the ohmic drain contact D for example . The device is symmetrical with respect to the axis BB 'as well as with respect to the axis CC' of Figure 1d.

In order to avoid short circuits between the strip 12 and the metal layer 11, the parts 12a and 12b may consist of air bridges, or else a thin insulating dielectric layer such as the layer 2 described in the preceding examples may be provided at the same time under the bridges 4 and slightly projecting from the metal layer 11 in the Schottky grid regions, while leaving the ohmic contacts stripped on which the ribbon parts 12a and 12b come to rest and establish the electrical contact.

By this method, the sources S and drain D of the transistors 13 ′ are short-circuited and the Schottky gate G is grounded M via the metal layer 11.

It then suffices to provide a connection line 15 to connect at least one ohmic contact S or D to an adjustable bias voltage V _DD .

As has been said previously, the strip 12, its parts 12a and 12b can be produced by any metal suitable for producing the second interconnection levels of the integrated circuits. Consequently, the connection line 15 which connects the ohmic contacts can be produced using the same technology.

The phase β of the slow wave line is then electronically adjustable by adjusting the bias voltage V _DD which varies the gate-source capacitance of the transistor 13 '.

EXAMPLE OF IMPLEMENTATION VII

This example is illustrated by FIG. 12 schematically seen from above.

• elle présente une surface réduite ;
• elle est réalisable sur la face principale du circuit intégré ;
• ses connexions sont compatibles avec des éléments de circuits planaires ;
• ses connexions sont compatibles avec les éléments réalisés sur la face principale du circuit intégré ;
• cette ligne est notamment basse impédance.

The slow wave transmission line, the essential elements of which have been given, and of which a certain number of examples among the many possible variants has been described in examples I to VI, solves, as we have seen, among other two problems crucial techniques for the implementation of integrated circuits in general and MMIC's in particular, namely:

• it has a reduced surface;
• it can be performed on the main face of the integrated circuit;
• its connections are compatible with elements of planar circuits;
• its connections are compatible with the elements made on the main face of the integrated circuit;
• this line is notably low impedance.

FIG. 12 represents the connection of such a low impedance slow wave line and of reduced surface area, with a high impedance coplanar line.

By coplanar line is meant a line made on the main face of the integrated circuit or MMIC, showing a central conductive tape of small transverse dimension disposed between two conductive tapes of larger transverse dimension. The impedance of the coplanar line depends on the transverse dimension of the central conductive tape in which the distance which separates it from the two other tapes generally connected to a reference potential or mass is propagated. The phase shift (generally expressed in wavelength, for example λ / 4, λ / 2) depends on the length of the line.

Other factors are involved in the actual calculation of the characteristics of the line such as: the thickness of the ribbon, the nature of the substrate.

Coplanar lines can be used for both high impedance lines and low impedance lines. But, if the high impedance coplanar lines have dimensions compatible with integrated circuits, on the other hand, the low-impedance coplanar lines have dimensions, notably transverse, which occupy an enormous surface of the integrated circuit, which is entirely unfavorable for monolithic integration.

The low impedance slow wave line then makes it possible, by calculating its length and its characteristics appropriately, to form a line having for example the same phase shift as a coplanar line, (λ / 4, λ / 2).

Therefore, when the problem arises of making a low impedance line, those skilled in the art have every interest in adopting the structure of one of the slow wave lines according to the invention, as described above.

On the other hand, when the problem arises of making an impedance transformer, those skilled in the art have every interest in adopting the structure shown seen from above in FIG. 12, showing the connection between a high impedance coplanar line (by example λ / 4) and a low impedance slow wave line according to the invention (for example λ / 4 also).

• une largeur réduite d'un facteur = 10 ;
• une longueur réduite d'un facteur ≃ 2 à 4.

Indeed, compared to a coplanar line of the same characteristic, the low impedance slow wave line according to the invention will have:

• a width reduced by a factor = 10;
• a length reduced by a factor ≃ 2 to 4.

As shown in FIG. 12, the part P₁ delimited by broken lines is the low impedance slow wave line according to the invention, and the part P₂ is a high impedance coplanar line as known to those skilled in the art.

On the substrate 10, a first metallization level will form the ground plane 11 of the slow wave line P₁ separating into two ribbons to form the ground lines 11a and 11b of the coplanar line P₂.

The slow wave line P₁ will include, produced on the conductive layer 11, a dielectric layer 2, as already described, extending beyond the ground plane 11 of the slow wave line P₁ in the regions necessary to avoid short circuits between the ground plane 11 and the line 12 produced subsequently.

Then, the slow wave line P₁ will include the ribbon 12, realizing as already described pillars 13 and BRIDGES 4, ribbon 12 which continues directly on the substrate 10 between the ground lines 11a and 11b to form the coplanar structure of the line P₂. To this end, it is generally necessary for the dielectric layer 2 to extend beyond the ground plane 11 of the slow wave line P₁ on the side of the coplanar line P₂ to avoid short circuits between the ground plane 11 and the line 12.

If an output O is desired for the slow wave line P₁, on the side opposite to its connection with the coplanar line P₁, the dielectric layer 2 is also extended beyond the ground plane 11, and the strip 12 is provided with a output O as shown in Figures 1a, 1b, 1c.

EXAMPLE VIII

This example is not illustrated.

We have seen that the low impedance slow wave line had a conductive plane 11, connectable to ground, in contact with the upper main surface of the substrate.

If necessary, contact with another ground plane made on the second face of the substrate, or rear face of the substrate, can be made as is known to those skilled in the art under the name of "metallized hole".

EXAMPLE IX

In this example, illustrated by FIG. 13, we show an example of application of the impedance transformer described in example VII, to an integrated circuit.

As shown in Figure 13, the circuit includes a transistor, for example T₁ field effect, having a gate G grille for receiving a signal F signal in a band of given frequencies, having a drain D₁ connected to a DC bias voltage V _D1 through a load R₁, having an output O₁ for said signal and having a source S₁ for example connected to ground M.

A circuit based on an impedance transformer P₁ + P₂ can be applied to the gate G₁ of the transistor T₁.

A high impedance line P₂, for example λ / 4 is connected by one end to the gate G₁ and by its other end both to a low impedance line P₁ slow waves according to the invention and to a DC bias voltage V _G1 .

The low impedance line P₁ is therefore connected at one end to both P₂ and V _G1 , and its other end is open in this application.

The slow wave line according to the invention has wide applications in integrated circuits of all kinds as well as in MMICs (microwave) because its operation can be, as we said, indifferent to the substrate, which it presents small dimensions compared to other lines having the same characteristics, and that it is compatible with all the integrated circuit technologies used to date.

EXAMPLE X

This exemplary embodiment is illustrated by FIGS. 14a, 14b, 14e, 2c.

FIG. 14a shows a slow wave line seen from above, of MICRORUBAN structure, having first characteristics identical to those of the line of example II.

Thus, this line is produced on a substrate 10 which can be of any material whatsoever. For example: completely insulating, fully conducting, semi-insulating or semiconducting.

• une couche conductrice 11, par exemple en un métal bon conducteur pouvant faire office de plan de masse M dedimension transversale W1 ;
• une couche diélectrique 2, de permitivité relative ε_r2 et d'épaisseur e₂, de dimension transversale W₃ ;
• un ruban en un matériau conducteur, par exemple un métal bon conducteur 12 ; ce ruban 12, de faible dimension transversale W₂, forme avec les couches précédentes une structure périodique, de périodicité ℓ : à cet effet, le ruban conducteur 12 comprend des parties 3 en contact avec la couche diélectrique 2, ces parties 3 étant de dimension longitudinale ℓ₂ (parallèlement à l'axe BB'), et des parties 4 suspendues entre deux parties 3, ces parties 4 suspendues ayant une dimension longitudinale ℓ₁ (parallèlement à l'axe B-B'), de sorte que :
  
  $\text{ℓ = ℓ₁ + ℓ₂ ;}$
  
  
  
  
• les dimensions transversales des couches 11, 2, 12, sont telles que : W₂ ≦ W₃ ≦ W₁.

On the substrate 10, the line comprises the succession of:

• a conductive layer 11, for example of a good conductive metal which can act as a ground plane M of transverse dimension W1;
• a dielectric layer 2, of relative permitivity ε _r2 and of thickness e₂, of transverse dimension W₃;
• a ribbon made of a conductive material, for example a good conductive metal 12; this strip 12, of small transverse dimension W₂, forms with the preceding layers a periodic structure, of periodicity ℓ: for this purpose, the conductive strip 12 comprises parts 3 in contact with the dielectric layer 2, these parts 3 being of longitudinal dimension ℓ₂ (parallel to the axis BB '), and parts 4 suspended between two parts 3, these suspended parts 4 having a longitudinal dimension ℓ₁ (parallel to the axis B-B'), so that:
  
  $\text{ℓ = ℓ₁ + ℓ₂;}$
  
  
  
  
• the transverse dimensions of the layers 11, 2, 12 are such that: W₂ ≦ W₃ ≦ W₁.

Par exemple la valeur de ℓ₃ peut approcher celle de ℓ₁ à quelques % près, ou être égale.The structure also comprises, with respect to Example II, an essential element consisting of parts 5 in which the layer 11 of the ground plane, like the dielectric layer 2, are hollowed out under the suspended parts 4, so that the surface of the substrate 10 appears. In this example X, the recess 5 is unique under each suspended part 4, and the longitudinal dimension of the recess 5 is:

$\text{ℓ₃ ≦ ℓ₁}$

For example, the value of ℓ₃ can approach that of ℓ₁ to within a few%, or be equal.

The structure of the line according to Example X appears more clearly in the enlarged schematic representation shown in FIG. 14b, in longitudinal section along the axis BB 'of the line in FIG. 14a. This figure shows that, to make the contact of the parts 3 of the ribbon 12 with the dielectric layer 2, the ribbon 12 is collapsed at the level of the parts 3. On the contrary, in the suspended parts 4, the ribbon 12 is raised by a height e ₁ with respect to the upper surface of the substrate which appears in the recess 5.

The hanging parts 4 are the parts in which the propagation takes place. In these parts, the strip 12 is suspended above a single dielectric 1, of relative permissiveness ε _r1 .

• PONTS les parties 4 du ruban 12 suspendues au-dessus du diélectrique 1, les ponts 4 ayant une longueur ℓ₁ ≃ ℓ₃ et constituant les régions de propagation ;
• PILIERS les parties 13 formées de l'associationde la couche conductrice inférieure 12, de la couche diélectrique 2 d'épaisseur e₂ et des parties 3 du ruban 12, les piliers 13 formant une structure MIM (métal-isolant-métal) de longueur ℓ₂.

As in Example II, the following is called:

• BRIDGES the parts 4 of the ribbon 12 suspended above the dielectric 1, the bridges 4 having a length ℓ₁ ≃ ℓ₃ and constituting the regions of propagation;
• PILLARS the parts 13 formed from the combination of the lower conductive layer 12, the dielectric layer 2 of thickness e₂ and the parts 3 of the tape 12, the pillars 13 forming a MIM (metal-insulator-metal) structure of length ℓ₂.

Figure 14e shows a cross section of the line along the axis AA 'of Figure 14a, at a bridge 4, and Figure 2c remains valid to show a cross section of the line along the axis CC' of FIG. 14a at the level of a pillar 13.

• une structure de ligne MICRORUBAN comprenant une couche conductrice inférieure 11 formant plan de masse M, un ruban conducteur supérieur 12 et, une partie intermédiaire diélectrique 1, 2 ;
• le fait que cette structure est périodique, de période ℓ, formée de PONTS 4 suspendus, de longueur ℓ₁, ces ponts dans lesquels se fait la propagation de l'onde étant disposés entre deux PILIERS 13 formés d'une structure capacitive. Dans cet exemple X, la structure capacitive est une structure MIM constituée de la couche inférieure conductrice 11, de la couche diélectrique 2, de permitivité ε_r2 et du ruban conducteur 12, les piliers ayant une longueur ℓ₂ telle que $\text{ℓ₂ + ℓ₁ = ℓ}$
  
  qui est la période de la structure :
• le fait que sous les ponts 4, est formé dans la couche diélectrique 2 et le plan de masse 11 au moins un évidement 5 ayant une longueur :
  
  $\text{ℓ₃ ≦ ℓ₁}$
  
  
  
  
• les valeurs des paramètres : ε_r1, ε_r2, ℓ₁, ℓ₂, ℓ₃, e'₁, la valeur de la capacité et W₁, W₂, W₃ de la structure de la ligne sont liées entre elles pour résulter en la propagation d'ondes lentes et fournir un déphasage important sur une longueur totale Δ de ligne de transmission courte.

From this exemplary embodiment X, it appears that the essential elements for producing a slow wave line reside in:

• a MICRORUBAN line structure comprising a lower conductive layer 11 forming a ground plane M, an upper conductive strip 12 and, a dielectric intermediate part 1, 2;
• the fact that this structure is periodic, of period ℓ, formed of suspended BRIDGES 4, of length ℓ₁, these bridges in which the propagation of the wave takes place being arranged between two PILLARS 13 formed of a capacitive structure. In this example X, the capacitive structure is a MIM structure consisting of the conductive lower layer 11, the dielectric layer 2, of permissiveness ε _r2 and of the conductive tape 12, the pillars having a length ℓ₂ such that $\text{ℓ₂ + ℓ₁ = ℓ}$
  
  which is the period of the structure:
• the fact that under the bridges 4, is formed in the dielectric layer 2 and the ground plane 11 at least one recess 5 having a length:
  
  $\text{ℓ₃ ≦ ℓ₁}$
  
  
  
  
• the values of the parameters: ε _r1 , ε _r2 , ℓ₁, ℓ₂, ℓ₃, e'₁, the value of the capacitance and W₁, W₂, W₃ of the structure of the line are linked together to result in the propagation of waves slow and provide a significant phase shift over a total length Δ of short transmission line.

In this example X, the value of the MIM capacities of the parts 13 is linked to ℓ₂, to e₂ and ε _r2 . On the other hand, the recesses 5 arranged in the bridge regions 4 play the role of inductors, making it possible to obtain an increase in the characteristic impedance of the line.

• le pas ℓ de la structure périodique peut être constant ou non.
• le matériau choisi pour réaliser le substrat n'a aucune influence sur le fonctionnement de la ligne ; le substrat ne sert que de support ;
• le dessin de la ligne peut être linéaire, en méandre, en spirale ; tout autre dessin imaginable est possible.
• la capacité peut être un élément passif ou actif. Dans l'exemple X, on a préféré un élément passif pour rendre la ligne plus compacte ; les lignes qui incluent des éléments actifs présentent d'autres propriétés explicitées précédemment.
• en outre la couche diélectrique de la structure MIM peut être éventuellement formée de deux couches diélectriques superposées. Ce genre de structure à deux couches diélectriques pour réaliser une capacité est à la portée de l'homme de métier et n'est donc pas représentée sur les dessins.

Apart from these essential elements:

• the step ℓ of the periodic structure can be constant or not.
• the material chosen to make the substrate has no influence on the operation of the line; the substrate only serves as a support;
• the line drawing can be linear, meandering, spiral; any other design imaginable is possible.
• capacity can be a passive or active element. In example X, we preferred a passive element to make the line more compact; the lines which include active elements have other properties explained above.
• additionally the dielectric layer of the structure MIM can optionally be formed from two superimposed dielectric layers. This kind of structure with two dielectric layers for producing a capacitance is within the reach of those skilled in the art and is therefore not shown in the drawings.

All these characteristics which lead to numerous variants of the slow wave transmission line, particularly easy to perform, particularly efficient, as already described for example in the embodiments derived from Examples I and II, such as Examples III, IV , V.

In this example X, compared with examples I or II, the increase in the deceleration factor, in conjunction with that of the characteristic impedance of the line, actually allows an optimal reduction in the dimensions of the lines.

In order to evaluate the performance of such a line, it is necessary to evaluate the propagation constant γ in the length ℓ of the line, or period.

   ℓ₃ la longueur des évidements sous les ponts équivalents à ℓ₁,
   Z₁, Z₂ les impédances caractéristiques respectivement dans les parties PONTS 4 et PILIERS 13.We will call hereafter:
γ₁, γ₂ the propagation constants respectively in the BRIDGE part 4, and in the PILLAR part 13.
ℓ₁, ℓ₂ the BRIDGE, PILLAR lengths already defined as $\text{ℓ₁ + ℓ₂ = ℓ}$

ℓ₃ the length of the recesses under the bridges equivalent to ℓ₁,
Z₁, Z₂ the characteristic impedances respectively in the BRIDGES 4 and PILLARS 13 parts.

The calculation of the phase constant β is carried out in the same manner as described in Example I. It follows from these calculations that by choosing:
ℓ₁, ℓ₂, ℓ₃
ε _r1 , ε _r2
e'₁ and e₂
W₁ and W₂
appropriately, the phase speed of the line is low. Hence the existence of the so-called slow wave mode already described in Example I.

However, it turned out that the way in which the person skilled in the art could act on ε _r1 and on e₁ which are essential parameters, was limited by the fact that, to date, the microstrip type propagation lines had always comprising the superposition of three layers: a ground plane M, a dielectric layer and a microstrip conductor, as it is precisely described in Example I.

This 3-layer structure was the result of constant teaching of the state of the art, and this teaching was an obstacle to an evolution making it possible to obtain an improvement compared to the slow wave structure mentioned above.

The problem was therefore to find an electronic solution to increase the slow wave factor in order to further reduce the dimensions of the lines, which makes it possible both to reduce the losses by wavelengths and to further increase the densities. integration, all without adding considerable technological difficulties.

en fonction de la fréquence F, pour différentes valeurs k1 de la hauteur e'₁ de diélectrique 1 sous les ponts, à savoir :
pour la courbe C, e'₁ = 2 µm
pour la courbe D, e'₁ = 2,4 µm
pour la courbe E, e'₁ = 2,8 µm
que le facteur de ralentissement R augmente lorsque l'épaisseur e'₁ de diélectrique 1 augmente, pour une même valeur de la fréquence F.Experiments have shown, as it appears on the curves of figure 15b which represents the variations of the deceleration factor ${\text{R = λ₀ / λ}}_{\text{g}}$

as a function of the frequency F, for different values k1 of the height e'₁ of dielectric 1 under the bridges, namely:
for curve C, e'₁ = 2 µm
for curve D, e'₁ = 2.4 µm
for curve E, e'₁ = 2.8 µm
that the deceleration factor R increases when the thickness e ₁ of dielectric 1 increases, for the same value of the frequency F.

However, if it is desirable to increase the height e ₁ of the bridges, the person skilled in the art quickly encounters a crippling technological problem, because, if one chooses air as the dielectric 1, for the reason than air is the best dielectric, so it becomes random to make the bridges above a certain value of e₁, maximum value which obviously also depends on the length ℓ₁ and the width W₁ of the conductor 11.

To solve this problem satisfactorily, according to the invention, an increase in the characteristic line impedance has been achieved, by forming the recesses 5 in the ground plane M under the bridges, recesses 5 which increase the role inductive of the line constituting the bridge.

We then have in addition parameters on which we could act according to Example I to increase the slowing factor, that is to say e₁, ε _r1 , possibilities for further improvement thanks to the self effect of these recesses.

de lignes en fonction de la fréquence F ;

• la courbe A représente ce facteur R, dans le cas d'une ligne selon l'exemple I : sans évidements ;
• la courbe B représente ce facteur R dans le cas d'une ligne selon l'exemple X : avec évidements 5.

De ces courbes, il apparaît très nettement que l'effet dû aux évidements est très important et bénéfique. Rien ne laissait prévoir à l'homme du métier que la réalisation de tels évidements sous les ponts produirait l'effet d'augmentation supplémentaire du facteur de ralentissement R et ceci en outre sans conduire à des désavantages plus grands que les avantages que l'on escomptait de l'augmentation de ce paramètre R, comme par exemple des pertes supplémentaires ou des perturbations non voulues de l'onde. En effet l'homme du métier sait bien que dès que l'on fait varier 1 paramètre dans un système qui comprend un nombre assez grand de paramètres, il devient difficile de prévoir l'effet exact obtenu, même dans le cas où l'on peut réaliser des simulations au moyen de programmes d'ordinateur. En effet dans ce dernier cas, on est toujours amené à supposer certains paramètres négligeables en théorie, lesquels s'avèrent dans la pratique précisément non négligeables.Figure 15a shows the deceleration factor ${\text{R = λ₀ / λ}}_{\text{g}}$

lines as a function of frequency F;

• curve A represents this factor R, in the case of a line according to example I: without recesses;
• curve B represents this factor R in the case of a line according to example X: with recesses 5.

From these curves, it very clearly appears that the effect due to the recesses is very significant and beneficial. Nothing suggested to a person skilled in the art that the realization of such recesses under the bridges would produce the effect of additional increase in the deceleration factor R and this in addition without leading to disadvantages greater than the advantages which one expected to increase this parameter R, such as additional losses or unwanted disturbances of the wave. In fact, those skilled in the art know very well that as soon as one parameter is varied in a system which includes a fairly large number of parameters, it becomes difficult to predict the exact effect obtained, even in the case where one can perform simulations using programs computer. Indeed in the latter case, we are always led to assume certain parameters which are negligible in theory, which prove in practice to be precisely non-negligible.

The recesses 5 indeed produce the desired favorable effect of additional deceleration, by acting both on the characteristic impedance of the line, on the thickness of dielectric e'₁ under the bridges, on the value of the permissiveness ε _r1 since the only most favorable dielectric can be found under the bridges, and all this while benefiting from a technology which is easy to implement, the recesses 5 being produced during conventional stages of integrated circuit technology.

Thus a clear improvement is still obtained compared to examples I and II, this in a simple and elegant manner, without running the risk of envisaging unacceptable values for the value of the height e ₁ of the bridges.

en fonction de la période ℓ des lignes, pour une valeur donnée de la fréquence F (dans cet exemple de figure, F = 12 GHz et ε_r1 = 1, pour différentes valeurs du paramètre $\text{k₂ = ℓ₁/ℓ₂}$

où ℓ₁ est la longueur des ponts et ℓ₂ la longueur des piliers dans cet exemple X, avec ℓ₃ ≃ ℓ₁.The curves in Figure 15c represent the deceleration factor ${\text{R = λ₀ / λ}}_{\text{g}}$

as a function of the period ℓ of the lines, for a given value of the frequency F (in this example of a figure, F = 12 GHz and ε _r1 = 1, for different values of the parameter $\text{k₂ = ℓ₁ / ℓ₂}$

where ℓ₁ is the length of the bridges and ℓ₂ the length of the pillars in this example X, with ℓ₃ ≃ ℓ₁.

The curves in FIG. 15c teach that there is an optimal value where R passes through a maximum, which of course depends on the other parameters of the lines, and that the person skilled in the art can therefore act on these parameters to optimize the system.

The reduction in the dimensions of the line is such that a person skilled in the art can then think of incorporating complex devices, using these lines, in circuits with high integration density. This was previously impossible. The components using the lines were produced on substrates juxtaposed with integrated circuits microwave and connected by thin wires, which limited the cutoff frequency. On the contrary, by having the possibility of producing the lines on the same substrate as the microwave transistors and other components of the integrated circuits, the connections are technologically identical to those of the rest of the circuit and they no longer limit the frequency.

To meet the conditions established by these calculations, in this example X, a slow wave line was produced having the same technological characteristics as those of example I, for example the same materials.

However, it is obvious that other choices can be made for the materials.

Table II below collates the preferred values of the parameters for implementing the line in this example X.

FIG. 14a shows that the other characteristics of the line of example X are very comparable to those of the line of examples I and II shown in FIGS. 1a and 1b.

FIG. 5e is also valid for representing the real and imaginary parts, respectively Re (Z _c ) and Im (Z _c ) of the characteristic impedance Z _c of this line.

Figure 6 is also valid for showing the losses α in the line, expressed in dB / cm, as a function of the frequency F in GHz. The curve α ′ in this figure 6 represents the losses in dB per wavelength.

Because the phase speed is lower, the slow wave line has a total length Δ reduced compared to the line of Example I. The reduction in lengths is inversely proportional to the deceleration factor R. Now in the case of a conventional microstrip line at around 12 GHz, R was of the order of 2.5, while R was of the order of 4 in the line described in Example I. In Example X, as shown FIG. 15a, at this frequency, R is of the order of 4.5. As was said in Example I, the performance of the slow wave line according to the invention is not deteriorated, while it is notably shorter.

For example, for a 180 ° phase shift, in the frequency band KU, the present slow wave line structure produces losses evaluated at around 1dB.

EXAMPLE XI

This example is illustrated by FIG. 14c seen from above and by FIG. 14d which is a section along the axis BB 'of FIG. 14c.

In the previous example X, we studied the case where we only make 1 recess 5 under each bridge. In this example, several recesses 5a, 5b, etc. are made under each bridge, thus creating a period in the period ℓ.

The advantages are that an additional increase in the deceleration factor R is obtained due to the discontinuities thus produced.

A variant to this embodiment XI which proceeds from the same principle, is to provide for the capacities 13, capacities of different values, distributed alternately along the line. There is thus also obtained a period in the line period, and a consequent improvement in the line slowing factor.

On the other hand, the production of a line having both the characteristic of two or more recesses 5a, 5b etc. under the bridges, and capacities of alternating values for the pillars 13 is also possible. By varying these different factors, a person skilled in the art will easily obtain the results most suitable for each application envisaged.

EXAMPLE XII

In this example, one of the slow wave lines described above is applied to the production of a Lange coupler.

The coupler known from the publication IEEE, MTT, Dec. 1969, p.1150-1151 cited is constituted by at least 3 parallel lines connected 2 to 2 alternately to form an interdigitated structure. The cited publication shows a 3 dB coupler with 5 transmission lines. An electromagnetic field coupling appears between the adjacent parallel lines.

Figure 16a below shows schematically this coupler. FIG. 16b represents the same coupler seen from above, in a simplified manner, produced by means of layers specific to integrated circuits.

As shown in this figure 16a, the coupler comprises two so-called input poles N₁ and N₂, and two so-called output poles N₃ and N₄. According to FIG. 16a, the Lange coupler consists of 5 parallel microstrip lines including a so-called main line 110, electrically connected to lines 111 and 114, and two lines 112 and 113 electrically connected together, and forming a structure interdigitated by the fact that line 112 is arranged between lines 110 and 111 and line 113 between lines 110 and 114. The coupler is symmetrical: that is to say that if N₃ and N₄ are inputs, then N₁ and N₂ are outputs .

Lines 110 and 111 are electrically connected directly to pole N₁ by a simple conductor 101. The lines 110 and 114 are electrically connected directly to pole N₄ by a single conductor 104. Line 112 and line113 are electrically connected to poles N₂ and N₃ respectively by single conductors 102 and 103.

• le point milieu de la ligne principale 110 est relié d'une part à l'extrémité ouverte de la ramification 111 et d'autre part à l'extrémité ouverte de la ramification 114 ;
• l'extrémité ouverte de la ligne 112 est reliée au point commun de la ligne 113 et du conducteur 103 ;
• l'extrémité ouverte de la ligne 113 est reliée au point commun de la ligne 112 et du conducteur 102.

So :

• the midpoint of the main line 110 is connected on the one hand to the open end of the branch 111 and on the other hand to the open end of the branch 114;
• the open end of line 112 is connected to the common point of line 113 and conductor 103;
• the open end of line 113 is connected to the common point of line 112 and conductor 102.

The poles N₂ and N₃ are electrically connected, by this assembly, crosswise with respect to the poles N₁ and N₄, as shown in Figure 16a and in Figure 16b.

On the other hand, the adjacent lines 110 and 112, the 110 and 113, are respectively parallel over a length L, while, in the interdigitated structure 110, 111, 112, the line 111 is parallel to the line 112 over a length equal to L / 2. It is the same in the interdigitated structure 110, 114, 113, where the line 114 is parallel to the line 113 over a length also L / 2.

The length L can be of the order of a quarter of the wavelength λ of the signal transported according to the prior art.

Lines 111, 112, 110, 113, 114 of the Lange coupler can be produced using slow wave lines according to the invention. In FIG. 16b, the connections 115, 116, 117 and 118 are formed by means of a conductive layer arranged at a level different from the layers 11 and 12, with openings on the layer 12 at the appropriate places to form the electrical connection with layer 12 according to a technique known as VIA well known to those skilled in the art, and with portions of insulating layers in the parts where on the contrary the electrical connection is not desired with layers 11 or 12. The other simple connections can be formed by means of parts of the conductive layer 12.

FIG. 16c represents an enlarged part of the coupler of FIG. 16b, in which it appears that the lines used by way of nonlimiting example to produce the coupler of example XII, are those described in example X.

In the case of FIG. 16c, the recesses 5 of the lines are produced individually under each bridge 4.

FIG. 16d represents an enlarged part of the coupler of FIG. 16b, in which it appears that the recesses 5 of the parallel lines, for example 112, 110, 113, 114 can be grouped together, to form a single recess 5, the bridges 4 being respectively opposite for all the lines, and the pillars 13 also. This device has a technological advantage over the previous one, due to its simplicity of construction; in fact the mask relating to the recesses 5 is less critical to position. This coupler then accepts the same operating principle as the known coupler. By making the lines necessary for the formation of such a Lange coupler, by means of the slow wave lines according to the invention, we also obtain the advantages that this device is very efficient and much more compact, compatible with circuit projects. integrated with high density, and of a low cost for the applications general public, in the field of television or the automobile for example.

FIG. 17 shows on the curve M, the adaptation of the coupler in dB as a function of the frequency F, and on the curve K the coupling in dB, as a function of the frequency F. These curves show that, by producing the coupler at using the lines of Example X, this coupler can be favorably used in a wide frequency band, around 12 GHz.

EXAMPLE XIII

As shown in Figure 18, a device conventional transceiver, known to any person skilled in the art, includes an input Q₁ for a first signal V₁, at the frequency F₁, propagating through an amplifier Δ₁ then through a duplexer 50, to an antenna A, then to the outside environment . This signal is applied to the N₁ pole of the duplexer 50 and exits to the N₃ pole of this duplexer 50.

This device further comprises an output Q₂ for a second signal V₂ at the frequency F₂. This signal is first picked up by the same antenna A then it propagates through the duplexer 50, in which it enters the pole N₃ and through an amplifier Δ₂ towards the output Q₂.

• a) une seule antenne doit être utilisée pour des raisons économiques ;
• b) le signal émis V₁ a généralement une amplitude très supérieure à celle du signal reçu V₂ ;
• c) il ne doit cependant pas se produire d'intermodulation ;
• d) le dispositif doit montrer une très bonne adaptation ;
• e) les pertes doivent être faibles ;
• f) la fréquence d'utilisation est éventuellement élevée, par exemple 60 GHz ;
• g) le dispositif doit être intégrable. et éventuellement :
• h) la bande de fréquence doit être large ;

The problems which arise in microwave transceivers, that is to say operating up to frequencies around 60 GHz, reside in the facts that:

• a) only one antenna should be used for economic reasons;
• b) the transmitted signal V₁ generally has an amplitude much greater than that of the received signal V₂;
• c) however, there should be no intermodulation;
• d) the device must show very good adaptation;
• e) losses must be low;
• f) the frequency of use is possibly high, for example 60 GHz;
• g) the device must be integrable. and eventually :
• h) the frequency band must be wide;

In this example XIII, these problems are solved by using as duplexer 50, a "Lange" coupler according to example XII connected to the other elements of the circuit in a special way to the invention.

According to the invention, there are circulated in this coupler two signals V₁ and V₂ at two different frequencies F₁ and F₂. As the Lange coupler is broadband, greater than 1 octave, the difference between the frequencies F₁ and F₂ is not a drawback if it is less than this passband, for example less than 1 octave. However, the length L of the main line will be chosen as a function of the wavelength λ of the weakest signal, generally V₂.

In variants of the invention, to increase the coupling factor, it is possible to provide a Lange coupler structure having several interdigitated structures similar to those formed by lines 111 and 112 on the one hand and 113 and 114 on the other hand . The coupler must show a center of symmetry.

The increase in the number of fingers makes it possible to increase the coupling factor and to decrease the losses in the coupler. Thus with 4 fingers (or 5 lines), the losses are 3 dB; with 6 fingers (or 7 lines), the losses are 2 dB, etc ...

On the other hand, increasing the number of fingers also makes it possible to increase the bandwidth of the device.

According to the invention, to make the transmitter-receiver device, the first signal V₁ at the frequency F₁ is applied to the pole N₁ of the Lange coupler as shown in FIG. 16b, and exits through the pole N₃, to be emitted then by an antenna A to the outside environment.

The second signal V₂, at the frequency F₂, picked up by the antenna is applied to the Lange coupler on the same pole N₃ (so as to solve the problem of using a single antenna), and leaves the coupler by the pole N₂.

The fourth pole N₄ of the Lange coupler is connected to ground through an impedance Z _C.

According to the invention, the conductor 101 (or pole N₁) is an input, the conductor 102 (or pole N₂) is an output, the conductor 104 (or pole N₄) is isolated, and the conductor 103 (or pole N₃) is both an input and an output.

Whereas, according to a use known to those skilled in the art, the conductor 103 is for example only an input and the conductors 101 and 102 are then only phase-shifted outputs, the conductor 104 being insulated.

The coupler is connected as shown in FIG. 19 on the one hand to the antenna A and on the other hand to the amplifiers Δ₁ and Δ₂.

Thus, to achieve the aims of the invention, as shown in FIG. 19, the signal V₁ at the frequency F₁ to be transmitted is processed by an amplifier Δ₁ with high gain and high insulation, and the signal V₂ of frequency F₂ received is processed by a low noise Δ₂ amplifier. In this case, the operation of the transceiver device is as follows:
The signal V₁ to be transmitted at the frequency F₁ first enters the node Q₁ of the transmitter-receiver device, then is processed by the amplifier Δ₁. It then passes by coupling from pole N₁ to pole N₃;
The signal V₁ to be transmitted at the frequency F₁ also passes directly by conduction into the characteristic impedance Z _c connected to the output pole N₄;
The signal to be transmitted V₁ at the frequency F₁ then propagates from the pole N₃ of the coupler to the outside environment by means of the antenna A.

The latter receives the second signal V₂ at another frequency F₂, of amplitude generally much lower than the first signal V₁ of frequency F₁. This second signal V₂ passes by conduction, directly from the input-output pole N₃ to the output pole N₂. Then the second signal V₂ is processed as already said by the low noise amplifier Δ₂ and leaves the device at the node Q₂.

• d'une part, il est de faible amplitude ;
• d'autre part, il se trouve, après le pôle N₂, devant la sortie de l'amplificateur Δ₁ à forts gain et isolation.

Il ne peut donc pas être retrouvé au noeud Q₁ d'entrée.The second signal V₂ or signal received at the frequency F₂, however also passes by coupling of the pole N₃ to the pole N₁, but:

• on the one hand, it is of low amplitude;
• on the other hand, it is, after the pole N₂, in front of the output of the amplifier Δ₁ with high gain and insulation.

It cannot therefore be found at the input node Q₁.

Therefore the object of the invention which is to successfully propagate the V₁ and V₂ signals without intermodulation is achieved.

If we consider the new use of a Lange coupler proposed by the invention compared to that known from the prior art, in fact, only the signal V₂ is processed in a substantially traditional manner. Indeed, for this signal, N₃ is an input and N₁ and N₂ are coupled and phase-shifted outputs. The application of the signal V₁ on the system according to the invention is then a completely original use.

Indeed, on the one hand, in the use, according to the invention the signal V₁ is not processed at all in the known traditional way. And on the other hand, the state of the art does not teach either to apply two different signals simultaneously, such as V₁ and V₂ on the same coupler.

An originality of this use therefore lies in the fact of applying to the coupler two signals V₁ and V₂, of different frequency and amplitude and of obtaining the propagation of these signals without intermodulation.

• on s'affranchit par ce montage, du fait que le coupleur de Lange est un élément passif, des effets des non-linéarités du dispositif actif (amplificateur distribué) connu de l'état de la technique ;
• le coupleur de Lange est intégrable du fait de ses dimensions, au contraire d'autres dispositifs passifs connus de l'homme du métier sous le nom de circulateurs, qui permettaient aussi une séparation des signaux, mais qui, du fait qu'ils ne sont pas intégrables, sont exclus des technologies du futur ;
• l'isolation du pôle N₁ vis-à-vis du pôle N₂ est très bonne (20 à 35 dB) ;
• les pertes sont faibles (1 à 3 dB) ;
• l'adaptation est très bonne (meilleure que 25 dB) ;
• les "traces" de V₂ au pôle N₁ ne peuvent se retrouver au noeud d'entrée Q₁ ;
• il n'y a donc pas d'intermodulation entre les signaux V₁ et V₂ ;
• les pertes peuvent être minimisées au besoin comme il a été dit, et la bande de fréquence peut être choisie plus ou moins large, en agissant sur les facteurs w, s, L du coupleur de Lange, où w = w₂ des exemples précédents, et où s est l'espacement entre les lignes du coupleur ;
• la structure du coupleur de Lange selon l'invention est aisée à mettre en oeuvre et d'un coût de réalisation faible ;
• cette technologie est tout à fait compatible avec la technologie des circuits intégrés MMICs (Monolithic Microwave Integrated circuits) ;

   Dans un exemple de réalisation on choisira, pour l'application aux hyperfréquences :
Z_c = 50 ohms ; w = 9 microns ; s = 7 microns.
L ≃ 200 µm pour 60 GHz , ou bien 1,5 mm pour 10 GHz ;
   Un coupleur de Lange en technologie microruban connue peut aussi être utilisé de la même manière que décrit ci-dessus, mais ses dimensions sont plus grandes.The advantages obtained by using a Lange coupler mounted according to the invention are numerous:

• by this arrangement, the fact that the Lange coupler is a passive element is freed from the effects of the non-linearities of the active device (distributed amplifier) known from the state of the art;
• the Lange coupler can be integrated because of its dimensions, unlike other passive devices known to those skilled in the art under the name of circulators, which also allow separation of the signals, but which, because they are not not integrable, are excluded from future technologies;
• the insulation of the N₁ pole from the N₂ pole is very good (20 to 35 dB);
• losses are low (1 to 3 dB);
• the adaptation is very good (better than 25 dB);
• the "traces" of V₂ at the pole N₁ cannot be found at the input node Q₁;
• there is therefore no intermodulation between the signals V₁ and V₂;
• the losses can be minimized if necessary as has been said, and the frequency band can be chosen to be more or less wide, by acting on the factors w, s, L of the Lange coupler, where w = w₂ from the previous examples, and where s is the spacing between the lines of the coupler;
• the structure of the Lange coupler according to the invention is easy to implement and of a low production cost;
• this technology is fully compatible with MMICs (Monolithic Microwave Integrated circuits) technology;

In an exemplary embodiment, for the application to microwave frequencies, we will choose:
Z _c = 50 ohms; w = 9 microns; s = 7 microns.
L ≃ 200 µm for 60 GHz, or 1.5 mm for 10 GHz;
A Lange coupler in known microstrip technology can also be used in the same manner as described above, but its dimensions are larger.

EXAMPLE XIV

In this example, the objects of the invention are achieved by using as duplexer 50, a coupler with branches, as described for example in the publication "Millimeter wave engineering and applications" by P. BHARTIA and IJ BAHL at John Wiley and Sons, New-York (A Wiley-Interscience Publication) p.355, or even in the publication de Microwave Journal, July 1988 p.119 and pp.122-123, entitled "Microstrip Power Dividers at mm-wave frequencies" by Mazen Hamadallah (p.115).

As described in these publications, and illustrated below by FIG. 20, a branch coupler comprises two sections of line 201 and 202 of length L and of impedance Z _c √2, connected at each of their ends by two sections of line 203 and 204 of impedance Z _c and of length L.

In series with the first line sections 201 and 202, there are line sections to form the poles N₁ and N₂ on the one hand, and N₃ and N₄ on the other hand, each of impedance Z _c .

où λ serait la longueur d'onde du seul signal d'entrée appliqué sur un pôle par exemple N₃. Le pôle N₄ serait isolé. Un signal direct serait recueilli sur le pôle N₁ et un signal couplé sur le pôle N₂.According to the documents cited, $\text{L = λ / 4}$

where λ would be the wavelength of the only input signal applied to a pole, for example N₃. The N₄ pole would be isolated. A direct signal would be collected on the N₁ pole and a coupled signal on the N₂ pole.

According to the invention, on the other hand, on the one hand, a type of slow wave lines is chosen chosen from those described above, and on the other hand, as shown in FIG. 19, two input signals, one V₁, are applied as before. on pole N₁, the other V₂ on pole N₃ (via the single antenna A). The pole N₄ is the isolated pole, the pole N₂ is the output pole for the signal V₂ and the pole N₃ is the output pole for the signal V₁.

As in Example XIII, and as illustrated by FIGS. 18 and 19, amplifiers Δ₁ and Δ₂ are added to the coupler to optimize the results.

The technology used is the same as in Example XIII, and the results are identical except that which concerns the bandwidth which is less wide.

However, to increase the bandwidth, the branch coupler can be provided with several branches parallel to the branches 201 and 202.

The surface occupied by the device according to Example XIV is also slightly greater than that occupied by the device according to Example XIII, but this device is nevertheless perfectly integrable.

EXAMPLE XI

.In an example of using the circuits according to examples XIII or XIV, to produce a microwave head of a radar transceiver module, as shown in FIG. 21, there is a generator 58 of signal V₁ to the frequency F₁ said local oscillator OL whose signal is applied to the amplifier Δ₁ possibly formed by two medium power amplifiers Δ'₁, Δ''₁, then the signal is applied to the pole N₁ of the coupler 50. The pole N₃ is applied to the antenna A, the pole N₄ is connected to the ground via the impedance Z _c , for example 50 Ω, the pole N₂ is connected to the input of the amplifier Δ₂ possibly formed by two low noise amplifiers Δ'₂, Δ''₂. The output of the amplifier Δ₂ is applied to a mixer 59 which also also receives the signal at the frequency F₁ coming from the local oscillator 58 and whose output provides the frequency
intermediate $\text{IF = | F₂-F₁ |}$

.

• . Communication sans fil (en anglais : MOBILE COMMUNICATION),
• . Radar doppler,
• . Application aux faisceaux hertziens, à l'électronique automobile (radar anti-collision, mesure des vitesses) etc... En particulier dans le domaine de l'automobile, on doit disposer d'une part de circuits l'automobile, on doit disposer d'une part de circuits intégrables pour des raisons de coûts de fabrication et d'autre part de circuits travaillant dans la bande 60 à 80 GHz pour des raisons d'encombrement spectral.

The applications of such a circuit are numerous:

• . Wireless communication (in English: MOBILE COMMUNICATION),
• . Doppler radar,
• . Application to radio-relay systems, to automobile electronics (anti-collision radar, speed measurement) etc ... In particular in the automobile field, one must have circuits on the one hand automotive, one must have on the one hand integrated circuits for reasons of manufacturing costs and on the other hand circuits working in the 60 to 80 GHz band for reasons of spectral congestion.

The circuit according to the invention is both integratable and perfectly capable of working at such high frequencies. It therefore fully meets these conditions, however severe they may be.

Claims (25)

Wave transmission line, in slow wave mode, of the so-called microstrip type, including a first so-called lower conductive layer acting as ground plane, a second so-called upper conductive layer in the form of a ribbon of specific transverse and longitudinal dimensions, and a third non-conductive material disposed between these two conductive layers, characterized in that the transmission line has, longitudinally, a periodic structure, each period, of length ℓ, being formed of a said bridge followed by a said pillar, in that each bridge consists of a section of the upper conductive tape, of length ℓ₁ <ℓ, disposed on the surface of said first part of the third material, which is of dielectric nature, and in that each pillar is a capacitor. Transmission line according to claim 1, characterized in that the first conductive layer serving as ground plane has at least one recess respectively under each bridge. Line according to claim 2, characterized in that the recesses in the ground plane are respectively in number greater than 1 under each bridge. Line according to one of claims 1 to 3, characterized in that the capacity forming the pillars is of the MIM (metal-insulator-metal) type consisting of the stack of the lower conductive layer, of said second part of the third material , which is of a dielectric nature, and of a section of the upper conductive strip of length ℓ₂, the sum of the respective lengths ℓ₁ of the bridges and ℓ₂ of the pillars being equal to the value ℓ of the period. Line according to one of Claims 1 to 3, characterized in that the capacity of the pillars is produced by the capacity of a diode or by the gate-source capacity of a field effect transistor. Line according to claim 4, characterized in that said second dielectric part included in the pillars of MIM structure has a thickness e₂ small compared to that e₁ of said first dielectric part under the bridges and forms a continuous layer disposed on the first conductive layer acting ground plane, this continuous dielectric layer having sufficient dimensions to avoid short circuits between the upper strip and the lower conductive layer acting as ground plane. Line according to Claim 4, characterized in that the said second dielectric part is limited to the regions of MIM structure, having dimensions sufficient to avoid short-circuits between the upper strip and the lower layer acting as ground plane. Line according to one of claims 6 or 7, characterized in that the first dielectric part of the third material is air, the second dielectric part of the third material is chosen between silica (SiO₂) and silicon nitride (Si₃N₄) . Line according to claim 8, characterized in that W₁ and W₂ being the respective transverse dimensions of the lower conductive layer and of the tape, ε _r1 and ε _r2 being the relative permitivities of the first and second parts of third material respectively under the bridges and in MIM structures, the characteristics of the line are given by ε _r1 = 1 (air), ε _r2 ≃ to 6 to 7, (silica or silicon nitride) e₁ ≃ 1.5 µm to 2.5 µm, e₂ ≃ e₁ / 10, ℓ₁ (bridge) ≃ 100 µm, ℓ₂ (pillar) ≃ ℓ₁ / 10, W₂ (tape) ≃ 20 µm, W₁ ≃ 100 µm (lower conductive layer), and possibly ℓ₃ (length of the recess of the plane of mass under bridges) equivalent to ℓ₁. Line according to one of Claims 1 to 9, characterized in that the capacities have alternating values along the line. Line according to one of the claims 6 to 8, characterized in that e₁ = e₂, e₁ being the thickness of the first dielectric part, and e₂ that of the second. Line according to one of claims 6 to 8, characterized in that the relative permissivities of the first and second dielectric parts are equal. Line according to one of Claims 1 to 12, characterized in that the length ℓ of the line period is constant to obtain a deceleration factor λ₀ / λ _g (defined by the ratio between the wavelength in a vacuum λ₀ and the wavelength propagating in the line λ _g ) constant at the same time as a non-constant phase shift β, as a function of the frequency in the line. Line according to one of claims 1 to 12, characterized in that the length ℓ of the period is increasing to obtain a deceleration factor λ₀ / λ _g (defined by the ratio between the wavelength λ₀ in vacuum and the wavelength λ _g in the line, not constant, at the same time as a constant phase shift β as a function of the frequency in the line. Line according to claim 14, characterized in that the length ℓ of the period increases geometrically. Line according to claim 5, characterized in that in the region of the pillars an active area suitable for receiving a transistor is formed in the line support, in that the lower conductive layer is narrowed in this region to have transverse dimensions and longitudinal characteristics of the Schottky contact or gate of a field effect transistor, in that this gate is arranged parallel to the longitudinal axis of the line, and on the surface of the active area, between two ohmic pads called source and drain of the transistor, having no electrical contact with the lower conductive layer, in that, longitudinally on either side of the active area, the upper ribbon is divided into two parts of ribbon which respectively establish electrical contact with the pads source and drain of the transistor, while avoiding short circuits between the lower conductive layer and the upper strip in the regions where these elements are superimposed with a short distance. Use of a line according to claim 16, characterized in that the ribbon and the lower conductive layer are each connected to different continuous potentials allowing the operation of the transistor in an area capable of resulting in a gate-source capacitance desired for the operation slow waves of the line. Integrated circuit including a line according to one of claims 1 to 16. Integrated circuit according to claim 18, further including a line of the so-called coplanar type, according to which a conductive tape is arranged on the surface of a support between two ground lines, characterized in that the tape of the coplanar line is arranged in continuity of the upper ribbon of the slow wave line, in that the two ground lines are connected to the lower conductive layer of the slow wave line in a single layer, and in that a portion of electrically insulating layer is disposed between the upper tape and the lower conductive layer in the connection region between the two types of line to avoid short circuits. Integrated circuit according to claim 18, including a directional coupler, characterized in that the transmission lines necessary to form this coupler are chosen from the lines according to one of claims 1 to 4, or 6 to 15. Circuit according to claim 20, characterized in that this coupler is of the so-called Lange type comprising an odd number of these interdigitated transmission lines. Circuit according to Claim 20, characterized in that this coupler is of the so-called branch type. Integrated circuit according to claim 20 for producing a transceiver device comprising a frequency duplexer for transmitting a first signal and receiving a second signal on a single pole, characterized in that the integrated frequency duplexer is a directional coupler according to one of claims 21 or 22, having two said first poles connected by electromagnetic coupling to two said second poles, in that one of said first poles constitutes an input for the first signal from a first amplifier, and the other says first pole an output for the second signal, which propagates towards the input of a second amplifier and in that one of said second poles constitutes an output for the first signal and an input for the second signal, and the other of said second pole is isolated. Circuit according to claim 23, characterized in that the pole of the duplexer which constitutes both the output for the first signal and the input for the second is connected to a single transmit-receive antenna for the first and second signals. Use of a circuit according to one of claims 23 or 24, to produce a radar.

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