DE29924130U1 - Device and system for downconverting an electromagnetic signal - Google Patents

Description

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PARKERVISIONJNC. New description pages

o.Z.:F1507GM-DE/D

Device and system for down-converting an electromagnetic signal

Background of the invention

Scope of the invention

The present invention relates to the down-conversion of electromagnetic (EM) signals. In particular, the present invention relates to the down-conversion of EM signals to intermediate frequency signals, the direct down-conversion of modulated EM signals to demodulated baseband signals, and the conversion of FM signals to non-FM signals. The present invention also relates to under-sampling and energy transfer at aliasing rates.

Related technology

Electromagnetic (EM) information signals (baseband signals) include, for example, video or picture baseband signals, voice baseband signals, computer baseband signals, etc. Baseband signals include analog baseband signals and digital baseband signals.

It is often advantageous to transmit EM signals at higher frequencies. This is generally true whether the transmission medium is a wire, an optical fiber, space, air, a liquid, etc. To increase efficiency and improve practical applicability, e.g. to obtain an improved radiating capability and an additional possibility for multiple channels of baseband signals, upconversion to a higher frequency is used. Conventional upconversion techniques modulate higher frequency carrier

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signals with baseband signals. Modulation is one of various techniques for impressing information from the baseband signals onto the higher frequency carrier signals. The resulting signals are referred to herein as modulated carrier signals. For example, the amplitude of an AM carrier signal changes in relation to changes in the baseband signal, the frequency of an FM carrier signal changes in relation to changes in the baseband signal, and the phase of a PM carrier signal changes in relation to changes in the baseband signal.

In order to process the information present in the baseband signal, the information must be extracted, or demodulated, from the modulated carrier signal. However, because conventional signal processing techniques are limited in terms of processing speed, conventional signal processing techniques cannot easily demodulate a baseband signal directly from a higher frequency modulated carrier signal. Instead, higher frequency modulated carrier signals must be converted to an intermediate frequency (IF) signal.
signal from which a conventional demodulator can demodulate the baseband signal.

Conventional buck converters have electrical components whose characteristics are frequency dependent. Therefore, conventional buck converters are designed for specific frequencies or frequency ranges and are not suitable for operation outside their design frequency range.

Conventional downconverters generate unwanted image signals and therefore must have filters to filter out the unwanted image signals. However, such filters reduce the power level of the modulated carrier signals. Therefore, conventional downconverters have power amplifiers that require external power sources.

When a received modulated carrier signal is relatively weak, such as in a radio receiver, conventional downconverters have additional power amplifiers that require additional external power sources.

The following are required without restriction:

an improved method and system for down-converting EM signals;

a method and system for directly downconverting modulated carrier signals into demodulated baseband signals;

a method and system for transferring energy and amplifying that energy transfer in the down-conversion of EM/signals;

a controlled impedance method and system for down-converting an EM signal;

a controlled pulse width or time window undersampling method and system for down-converting an EM signal;

a method and system for downconverting EM signals using a universal downconverter design that is easily configurable for different frequencies;

a method and system for down-converting EM signals using a local oscillator frequency substantially lower than the carrier frequency;

a method and system for down-converting EM signals using only a local oscillator;

a method and system for down-converting EM signals using fewer filters than in conventional down-converters;

a method and system for down-converting EM signals while requiring less power than conventional down-converters;

a method and system for down-converting EM signals while requiring less space than conventional down-converters;

a method and system for down-converting EM signals using fewer components than conventional down-converters;

a method and system for downconverting EM signals that can be implemented on an integrated circuit (IC); and

a method and system for down-converting EM signals, which can also be used as a method and system for up-converting a baseband signal.

Brief description of the invention

The present invention relates to methods, systems and apparatus for down-converting an electromagnetic (EM) signal by aliasing the EM signal and applications therefor.

In general, the invention relates to receiving an EM signal. The invention also relates to receiving an aliasing signal at an aliasing rate. According to the invention, the EM signal is aliased according to the aliasing signal in order to down-convert the EM signal. The term aliasing as used herein refers to both down-converting an EM signal by undersampling the EM signal at an aliasing rate and down-converting an EM signal by transferring energy from the EM signal at the aliasing rate.

In one embodiment, the EM signal is down-converted into an intermediate frequency (IF) signal according to the invention.

In another embodiment, the EM signal is downconverted to a demodulated baseband information signal.

In another embodiment, the EM signal is a frequency modulated (FM) signal that is down-converted to a non-FM signal, e.g., a phase modulated (PM) signal or an amplitude modulated (AM) signal.

The invention is applicable to any type of EM signal, including (but not limited to) modulated carrier signals (the invention is applicable to any modulation scheme or combination of modulation schemes) and unmodulated carrier signals.

Other features and advantages of the invention, as well as the structure and operation of various embodiments of the invention, are described in detail below with reference to the accompanying drawings. However, the present invention is not limited to the specific embodiments described herein. These embodiments are presented herein for illustration purposes only. Other embodiments will be apparent to those skilled in the art based on the present description.

The drawing in which an element first appears is typically indicated in the leftmost position(s) of the corresponding reference number.

Short description of the drawings

The present invention will now be described with reference to the accompanying drawings, in which:

Fig. 1 is a structural block diagram of an exemplary modulator;

Fig. 2 shows an exemplary analog modulating baseband signal;

Fig. 3 shows an exemplary digital modulating baseband signal;

Fig. 4 shows an exemplary carrier signal;

Figures 5A-5C show exemplary signal diagrams related to amplitude modulation;

Figures 6A-6C show exemplary signal diagrams related to amplitude shift keying (ASK) modulation;

Figures 7A-7C show exemplary signal diagrams related to frequency modulation;

Figures 8A-8C show exemplary signal diagrams related to frequency shift keying (FSK) modulation;

Figures 9A-9C show exemplary signal diagrams related to phase modulation;

Figures 10A-10C show exemplary signal diagrams related to phase shift keying (PSK) modulation;

Fig. 11 is a structural block diagram of a conventional receiver;

Figures 12A-D show various flow diagrams for down-converting an EM signal according to embodiments of the invention;

Fig. 13 is a structural block diagram of an aliasing system according to an embodiment of the invention;

Figures 14A-D show various flow diagrams for down-converting an EM signal by undersampling the EM signal according to embodiments of the invention;

Figures 15A-E show exemplary signal diagrams associated with the flow diagrams of Figures 14A-D for embodiments of the invention;

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Fig. 16 is a structural block diagram of an undersampling system according to an embodiment of the invention;

17 is a flowchart of an exemplary method for determining an aliasing rate according to an embodiment of the invention;

Figures 18A-E show exemplary signal diagrams associated with the downconversion of a digital AM signal by undersampling to an intermediate frequency signal, for embodiments of the invention;

Figures 19A-E show exemplary signal diagrams associated with the downconversion of an analog AM signal by undersampling into an intermediate frequency signal, for embodiments of the invention;

Figures 20A-E show exemplary signal diagrams associated with the downconversion of an analog FM signal by undersampling into an intermediate frequency signal, for embodiments of the invention;

Figures 2IA-E show exemplary signal diagrams associated with the downconversion of a digital FM signal by undersampling into an intermediate frequency signal, for embodiments of the invention;

Figures 22A-E show exemplary signal diagrams associated with the downconversion of a digital PM signal by undersampling to an intermediate frequency signal, for embodiments of the invention;

Figures 23A-E show exemplary signal diagrams associated with the downconversion of an analog PM signal by undersampling into an intermediate frequency signal, for embodiments of the invention;

24A is a structural block diagram of a sequence switching undersampling system according to an embodiment of the invention;

Fig. 24B is an exemplary timing diagram of an undersampling signal for an embodiment of the invention;

Fig. 24C is an exemplary timing diagram of a decoupling or isolation signal for an embodiment of the invention;

Figures 25A-H illustrate exemplary aliasing signals with different aliasing rates for embodiments of the invention;

Fig. 26A is a structural block diagram of an exemplary sample and hold system according to an embodiment of the invention;

26B is a structural block diagram of an exemplary inverted sample and hold system according to an embodiment of the invention;

Fig. 27 is a structural block diagram of a sample and hold module according to an embodiment of the invention;

Figures 28A-28D illustrate exemplary implementations of a switching module in accordance with embodiments of the invention;

Figures 29A-F illustrate exemplary implementations of a holding module according to embodiments of the invention;

Fig. 29G illustrates an integrated undersampling system according to embodiments of the invention;

Figures 29H-K illustrate exemplary implementations of pulse generators according to embodiments of the invention;

Fig. 29L shows an exemplary oscillator;

30 is a structural block diagram of an undersampling system including an undersampling signal optimizer according to embodiments of the invention;

Fig. 31 is a structural block diagram of an undersampling signal optimizer according to embodiments of the invention;

32A shows an example of an undersampling signal module according to an embodiment of the invention;

32B is a flow diagram of a state machine operation associated with an undersampling module for embodiments of the invention;

32C illustrates an exemplary undersampling module having an analog circuit with automatic gain control in accordance with embodiments of the invention;

Figures 33A-D are exemplary signal diagrams associated with the direct downconversion of an EM signal by undersampling to a baseband signal for embodiments of the present invention;

Figures 34A-F illustrate exemplary signal diagrams associated with an inverted sample and hold module for embodiments of the invention;

Figures 35A-E show exemplary signal diagrams associated with a direct downconversion of an analog AM signal by undersampling into a demodulated baseband signal, for embodiments of the invention;

36A-E are exemplary signal diagrams associated with downconversion of a digital AM signal by undersampling to a demodulated baseband signal, for embodiments of the invention;

Figures 37A-E show exemplary signal diagrams associated with a direct downconversion of an analog PM signal by undersampling into a demodulated baseband signal, for embodiments of the invention;

Figures 38A-E show exemplary signal diagrams associated with downconversion of a digital PM signal by undersampling into a demodulated baseband signal, for embodiments of the invention;

Figures 39A-E illustrate downconversion of an FM signal by undersampling into a non-FM signal according to embodiments of the invention;

Figures 40A-E illustrate downconversion of an FSK signal by undersampling into a PSK signal according to embodiments of the invention;

Figures 4IA-E show a downconversion of an FSK signal by undersampling into an ASK signal according to embodiments of the invention;

Figure 42 is a structural block diagram of an inverted sample and hold module according to an embodiment of the present invention;

Figures 43A and B show exemplary waveforms present in the circuit of Figure 31;

44A is a structural block diagram of a differential system according to embodiments of the invention;

Fig. 44B is a structural block diagram of a differential system having a differential input and a differential output according to embodiments of the invention;

Fig. 44C is a structural block diagram of a single-input, differential-output differential system according to embodiments of the invention;

Fig. 44D shows a differential input and single output system according to embodiments of the invention;

44E illustrates an exemplary differential input to single output system in accordance with embodiments of the invention;

Figures 45A-B illustrate a conceptual illustration of an aliasing process including undersampling and energy transfer, in accordance with embodiments of the invention;

Figures 46A-D illustrate various flow diagrams for down-converting an EM signal by transferring energy from the EM signal at the aliasing rate in accordance with embodiments of the invention;

Figures 47A-E show exemplary signal diagrams for embodiments of the invention associated with the flow diagrams of Figures 46A-D;

48 is a flowchart illustrating exemplary processing for determining an aliasing rate associated with an aliasing signal according to an embodiment of the invention;

Figures 49A-H illustrate exemplary energy transfer signals in accordance with embodiments of the invention;

Figures 50A-G are exemplary signal diagrams associated with downconverting an analog AM signal to an intermediate frequency signal by transferring energy at an aliasing rate, for embodiments of the invention;

Figures 51A-G are exemplary signal diagrams associated with downconversion of a digital AM signal by energy transfer at an aliasing rate to an intermediate frequency signal, for embodiments of the invention;

Figures 52A-G are exemplary signal diagrams associated with downconverting an analog FM signal to an intermediate frequency signal by energy transfer at an aliasing rate, for embodiments of the invention;

Figures 53A-G are exemplary signal diagrams associated with downconversion of a digital FM signal by energy transfer at an aliasing rate to an intermediate frequency signal, for embodiments of the invention;

Figures 54A-G are exemplary signal diagrams associated with downconversion of an analog PM signal by energy transfer at an aliasing rate to an intermediate frequency signal, for embodiments of the invention;

Figures 55A-G illustrate exemplary signal diagrams associated with downconverting a digital PM signal to an intermediate frequency signal by transferring energy at an aliasing rate, for embodiments of the invention;

Figures 56A-D show exemplary signal diagrams associated with direct down conversion for embodiments of the invention;

Figures 57A-F illustrate a direct downconversion of an analog AM signal into a demodulated baseband signal according to embodiments of the invention;

Figures 58A-F illustrate a direct downconversion of a digital AM signal into a demodulated baseband signal according to embodiments of the invention;

Figures 59A-F illustrate a direct downconversion of an analog PM signal into a demodulated baseband signal according to embodiments of the invention;

Figures 60A-F illustrate a direct downconversion of a digital PM signal into a demodulated baseband signal according to embodiments of the invention;

Figures 61A-F illustrate downconversion of an FM signal to a PM signal according to embodiments of the invention;

Figures 62A-F illustrate downconversion of an FM signal to an AM signal according to embodiments of the invention;

Fig. 63 is a block diagram of a power transmission system according to an embodiment of the invention;

64A illustrates an exemplary gated transmission system according to an embodiment of the invention;

64B illustrates an exemplary inverted gated transmission system according to an embodiment of the invention;

Fig. 65 shows an exemplary embodiment of the gated transmission module according to an embodiment of the invention;

Figures 66A-D illustrate exemplary implementations of a switching module according to embodiments of the invention;

67A illustrates an exemplary embodiment of the gated transmission module with a sequence switching module according to an embodiment of the invention;

67B is an exemplary timing diagram for a power transfer signal according to an embodiment of the invention;

67C is an exemplary timing diagram for a decoupling or separation signal according to an embodiment of the invention;

Figures 68A-F illustrate exemplary memory modules according to embodiments of the invention;

Fig. 68G illustrates an integrated gated transmission system according to an embodiment of the invention;

Figures 68H-K exemplary pulse width generators;

Fig. 68L shows an oscillator according to an embodiment of the present invention;

Fig. 69 shows an energy transfer system with an optional energy transfer signal module according to an embodiment of the invention;

Fig. 70 an aliasing module with input/output impedance matching according to an embodiment of the invention;

Fig. 71 an exemplary pulse generator;

Figures 72A and B show exemplary waveforms associated with the pulse generator of Figure 71;

Fig. 73 shows an exemplary energy transfer module with a switching module and a reactive storage module according to an embodiment of the invention;

Fig. 74 shows an exemplary inverted gated transmission module with a switching module and a memory module according to an embodiment of the invention;

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Figures 75A-F illustrate exemplary signal diagrams associated with an inverted gated energy transfer module for embodiments of the invention;

Figures 76A-E illustrate energy transfer modules in various differential configurations according to embodiments of the invention;

77A-C illustrate exemplary impedance matching circuits in accordance with embodiments of the invention;

Figures 78A-B illustrate exemplary undersampling systems in accordance with embodiments of the invention;

Figures 79A-F illustrate exemplary timing diagrams for undersampling systems according to embodiments of the invention;

Figures 80A-F are exemplary timing diagrams for an undersampling system in which a load has a relatively low impedance, according to embodiments of the invention;

Figures 8IA-F show exemplary timing diagrams for a different sampling system in which the holding capacitance has a larger value, according to embodiments of the invention;

Figures 82A-B illustrate exemplary energy transfer systems according to embodiments of the invention;

Figures 83A-F illustrate exemplary timing diagrams for energy transfer systems according to embodiments of the invention;

Figures 84A-D illustrate downconversion of an FSK signal to a PSK signal according to embodiments of the invention;

85A illustrates an exemplary power transfer signal module according to an embodiment of the invention;

85B is a flow diagram of a state machine operation according to an embodiment of the invention;

Fig. 85C shows an exemplary energy transfer signal module;

Fig. 86 is a schematic diagram of a circuit for down-converting a 915 MHz signal to a 5 MHz signal and

using a 101.1 MHz clock according to an embodiment of the invention;

Fig. 87 shows simulation waveforms for the circuit of Fig. 86 according to embodiments of the invention;

88 is a schematic diagram of a circuit for down-converting a 915 MHz signal to a 5 MHz signal using a 101 MHz clock according to an embodiment of the invention;

Fig. 89 shows simulation waveforms for the circuit of Fig. 88 according to embodiments of the invention;

90 is a schematic diagram of a circuit for down-converting a 915 MHz signal to a 5 MHz signal using a 101.1 MHz clock according to an embodiment of the invention;

Fig. 91 shows simulation waveforms for the circuit of Fig. 90 according to embodiments of the invention;

Fig. 92 is a schematic diagram of the circuit of Fig. 86 connected to an FSK source that switches between 913 and 917 MHz at a baud rate of 500 kBaud, according to an embodiment of the invention;

Fig. 93 shows the original FSK waveform 9202 and the down-converted waveform 9204 at the output of the load impedance matching circuit according to an embodiment of the invention;

94A illustrates an exemplary energy transfer system in accordance with an embodiment of the invention;

Figures 94B-C are exemplary timing diagrams for the exemplary system of Figure 94A;

95 illustrates an exemplary redirection network in accordance with an embodiment of the present invention;

96 illustrates an exemplary redirection network in accordance with an embodiment of the present invention;

Fig. 97 an exemplary embodiment of the invention;

98A illustrates an exemplary real-time pulse width control circuit according to an embodiment of the invention;

98B is a timing diagram of an exemplary clock signal for real-time pulse width control according to an embodiment of the invention;

98C is a timing diagram of an exemplary optional enable signal for real-time pulse width control according to an embodiment of the invention;

98D is a timing diagram of an inverted clock signal for real-time pulse width control according to an embodiment of the invention;

98E is a timing diagram of an exemplary delayed clock signal for real-time pulse width control according to an embodiment of the invention;

98F is a timing diagram of an exemplary energy transfer signal having pulses with real-time controlled pulse widths according to an embodiment of the invention;

Fig. 99 is a block diagram of a differential system using non-inverted gated transfer units according to an embodiment of the invention;

Fig. 100 shows an exemplary embodiment of the invention;

Fig. 101 shows an exemplary embodiment of the invention;

Fig. 102 shows an exemplary embodiment of the invention;

Fig. 103 shows an exemplary embodiment of the invention;

Fig. 104 shows an exemplary embodiment of the invention;

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Fig. 105 shows an exemplary embodiment of the invention;

Fig. 106 shows an exemplary embodiment of the invention;

Fig. 107A is a timing diagram for the exemplary embodiment of Fig. 103;

Fig. 107B is a timing diagram for the exemplary embodiment of Fig. 104;

Fig. 108A is a timing diagram for the exemplary embodiment of Fig. 105;

Fig. 108B is a timing diagram for the exemplary embodiment of Fig. 106;

Fig. 109A shows an exemplary embodiment of the invention;

Fig. 109B Equations for determining charge transfer according to the invention;

Fig. 109C shows inventive relationships between a capacitance charge and a pulse width;

Fig. 109D shows inventive relationships between a capacitance charge and a pulse width;

Fig. 109E shows equations used in the present invention to represent the relationship between power and charging;

Fig. 109F is an equation used in accordance with the invention to determine the passband or insertion loss;

Fig. HOA an aliasing module 11000 with a single FET configuration;

Fig. HOB is a diagram showing FET conductance as a function of voltage V _GS ;

Fig. 1A-C show signal waveforms associated with the aliasing module 11000;

Fig. 112 an aliasing module 11200 with a complementary FET configuration;

Figures 113A-E illustrate signal waveforms associated with aliasing module 11200;

Fig. 114 an aliasing module 11400;

Fig. 115 an aliasing module 11500;

Fig. 116 an aliasing module 11602;

Fig. 117 an aliasing module 11702;

Figures 118-120 show signal waveforms associated with the aliasing module 11602;

Figures 121-123 show signal waveforms associated with the aliasing module 11702;

124A is a block diagram of a splitter according to an embodiment of the invention;

124B is a more detailed diagram of a splitter according to an embodiment of the invention;

Figures 124C and 124D illustrate exemplary waveforms associated with the manifold of Figures 124A and 124B;

Fig. 124E is a block diagram of an I/Q circuit with a splitter according to an embodiment of the invention;

Figures 124F-124J illustrate exemplary waveforms associated with the diagram of Figure 124A;

Fig. 125 is a block diagram of a switching module according to an embodiment of the invention;

Fig. 126A is an implementation example of the block diagram of Fig. 125;

Figures 126B-126Q illustrate exemplary waveforms associated with the circuit of 126A;

Fig. 127A shows another implementation example of the block diagram of Fig. 125;

Figures 127B-Q illustrate exemplary waveforms associated with the circuit of Figure 127A;

Fig. 128A shows an exemplary MOS-FET embodiment of the invention;

Fig. 128B shows an exemplary MOS-FET embodiment of the invention;

Fig. 128C shows an exemplary MOS-FET embodiment of the invention;

129A shows another implementation example of the block diagram of Fig. 125;

129B-129Q are exemplary waveforms associated with the circuit of Fig. 127A;

130 and 131 show amplitude and pulse width modulated transmitters according to embodiments of the invention;

Figures 132A-D and 133-134 show exemplary signal diagrams associated with the amplitude and pulse width modulated transmitter according to embodiments of the invention;

Figure 135 is a block diagram of an embodiment of a receiver for recovering the amplitude or pulse width modulated information;

Fig. 136A-G exemplary waveform generator associated
Signal diagrams according to embodiments of the invention;

Figures 137-139 are exemplary schematic diagrams illustrating various circuits used in the receiver of Figure 135;

Figures 140-143 Time and frequency domain diagrams of alternative transmitter output waveforms;

Figures 144 and 145 show differential receivers according to embodiments of the invention; and

Figures 146 and 147 show time and frequency domains for a narrowband/constant carrier signal according to an embodiment of the invention.

Table of contents
I. Introduction

1. General terminology

1.1 Modulation

1.1.1 Amplitude modulation

1.1.2 Frequency modulation

1.1.3 Phase modulation

1.2 Demodulation

2. Overview of the invention

2.1 Aspects of the invention

2.2 Downconversion by undersampling

2.2.1 Downconversion to an intermediate frequency (IF) signal

2.2.2 Direct-to-data downconversion

2.2.3 Modulation implementation

2.3 Downward conversion through energy transfer

2.3.1 Down-conversion to intermediate frequency (IF) signal

2.3.2 Direct-to-data downconversion

2.3.3 Modulation implementation

2.4 Determining the aliasing rate

3. Advantages of the invention using an exemplary conventional receiver for comparison purposes

II. Undersampling

1. Downconversion of an EM carrier signal into an EM intermediate frequency signal by undersampling the EM carrier signal with the aliasing rate

1.1 High-level description

1.1.1 Description of the functionality

1.1.2 Description of the structure

1.2 Exemplary embodiments

1.2.1 First exemplary embodiment: Amplitude modulation 1.2.1.1 Description of the functionality

1.2.1.1.1 Analog AM carrier signal

1.2.1.1.2 Digital AM carrier signal 1.2.1.2 Description of the structure

1.2.2 Second exemplary embodiment: frequency modulation

1.2.2.1 Description of the functionality

1.2.2.1.1 Analog FM carrier signal

1.2.2.1.2 Digital FM carrier signal

1.2.2.2 Description of the structure

1.2.3 Third exemplary embodiment: phase modulation

1.2.3.1 Description of the functionality

1.2.3.1.1 Analog PM carrier signal

1.2.3.1.2 Digital PM carrier signal

1.2.3.2 Description of the structure

1.2.4 Other embodiments
1.3 Implementation examples

2. Direct downconversion of an EM signal into a baseband signal (direct-to-data)

2.1 High-level description

2.1.1 Description of the functionality

2.1.2 Description of the structure

2.2 Exemplary embodiments

2.2.1 First exemplary embodiment: Amplitude modulation

2.2.1.1 Description of the functionality

2.2.1.1.1 Analog AM carrier signal

2.2.1.1.2 Digital AM carrier signal

2.2.1.2 Description of the structure

2.2.2 Second exemplary embodiment: phase modulation

2.2.2.1 Description of the functionality

2.2.2.1.1 Analog PM carrier signal

2.2.2.1.2 Digital PM carrier signal

2.2.2.2 Description of the structure

2.2.3 Other embodiments

2.3 Implementation examples

3. Modulation implementation

3.1 High-level description

3.1.1 Description of the functionality

3.1.2 Description of the structure

3.2 Exemplary embodiments

3.2.1 First exemplary embodiment: Downconversion of an FM signal into a PM signal

3.2.1.1 Description of the functionality

3.2.1.2 Description of the structure

3.2.2 Second exemplary embodiment: Downconversion of an FM signal into an AM signal

3.2.2.1 Description of the functionality

3.2.2.2 Description of the structure

3.2.3 Other exemplary embodiments

3.3 Implementation examples

4. Implementation examples

4.1 The undersampling system as a sample-and-hold system

4.1.1 The sample-hold system as a switching module and holding module

4.1.2 The sample-and-hold system as a sequence switching module

4.1.3 Implementation examples of the switching module

4.1.4 Implementation examples of the holding module

4.1.5 Optional undersampling signal module

4.2 The undersampling system as an inverted sample-and-hold circuit

4.3 Other implementations

5. Optional optimizations of the undersampling process with an aliasing rate

5.1 Doubling the aliasing rate (Far) of the undersampling signal

5.2 Differential implementations

5.2.1 Differential input to differential output

5.2.2 Single input to differential output

5.2.3 Differential input to single output

5.3 Smoothing the down-converted signal

5.4 Load impedance and input/output buffering

5.5 Modification of the undersampling signal by feedback

III. Downconversion through energy transfer

0.1 Energy transfer versus undersampling 0.1.1 Overview of the undersampling process

0.1.1.1 Effects of reducing the load impedance 0.1.1.2 Effects of increasing the holding capacitance value 0.1.2 Introduction to energy transfer

1. Downconversion of an EM signal into an IF EM signal by transferring energy from the EM signal at an aliasing rate

1.1 High-level description

1.1.1 Description of the functionality

1.1.2 Description of the structure

1.2 Exemplary embodiments

1.2.1 First exemplary embodiment: Amplitude modulation

1.2.1.1 Description of the functionality

1.2.1.1.1 Analog AM carrier signal

1.2.1.1.2 Digital AM carrier signal

1.2.1.2 Description of the structure

1.2.2 Second exemplary embodiment: frequency modulation

1.2.2.1 Description of the functionality

1.2.2.1.1 Analog FM carrier signal

1.2.2.1.2 Digital FM carrier signal

1.2.2.2 Description of the structure

1.2.3 Third exemplary embodiment: phase modulation

1.2.3.1 Description of the functionality

1.2.3.1.1 Analog PM carrier signal

1.2.3.1.2 Digital PM carrier signal

1.2.3.2 Description of the structure

1.2.4 Other embodiments

1.3 Implementation examples

2. Direct downconversion of an EM signal into a demodulated baseband signal by energy transfer from the EM signal

2.1 High-level description

2.1.1 Description of the functionality

2.1.2 Description of the structure

2.2 Exemplary embodiments

2.2.1 First exemplary embodiment: Amplitude modulation 2.2.1.1 Description of the functionality

2.2.1.1.1 Analog AM carrier signal

2.2.1.1.2 Digital AM carrier signal 2.2.1.2 Description of the structure

2.2.2 Second exemplary embodiment: phase modulation

2.2.2.1 Description of the functionality

2.2.2.1.1 Analog PM carrier signal

2.2.2.1.2 Digital PM carrier signal

2.2.2.2 Description of the structure

2.2.3 Other embodiments 2.3 Implementation examples

3. Modulation implementation

3.1 High-level description

3.1.1 Description of the functionality

3.1.2 Description of the structure

3.2 Exemplary embodiments

3.2.1 First exemplary embodiment: Down-conversion of an FM signal into a PM signal

3.2.1.1 Description of the functionality

3.2.1.2 Description of the structure

3.2.2 Second exemplary embodiment: Down-conversion of an FM signal into an AM signal

3.2.2.1 Description of the functionality

3.2.2.2 Description of the structure

3.2.3 Other exemplary embodiments

3.3 Implementation examples

4. Implementation examples

4.1 The energy transmission system as a gate-controlled transmission system

4.1.1 The gate-controlled transmission system as a switching module and storage module

4.1.2 The gate-controlled transmission system as a sequential switching module

4.1.3 Implementation examples of the switching module

4.1.4 Implementation examples of the storage module

4.1.5 Optional power transfer signal module

4.2 The energy transmission system as an inverted gate-controlled transmission system

4.2.1 The inverted gate-controlled transmission system as a switching module and storage module

4.3 Rail-to-rail operation for improved dynamic range

4.3.1 Introduction

4.3.2 Complementary UFT structure for improved dynamic range

4.3.3 Preloaded configurations

4.3.4 Simulation examples

4.4 Optimized switching structures

4.4.1 Splitters in CMOS

4.4.2 I/Q circuit

4.5 Exemplary I and Q implementations

4.5.1 Switches with different sizes

4.5.2 Reduction of the total switch area

4.5.3 Charge carrier injection compensation

4.5.4 Overlapped capacity

4.6 Other implementations

5. Optional optimizations of energy transfer with an aliasing rate

5.1 Doubling the aliasing rate (Far) of the energy transfer signal

5.2 Differential implementations

5.2.1 An example to illustrate differential energy transfer

5.2.1.1 Differential input to differential output

5.2.1.2 Single input to differential output

5.2.1.3 Differential input to single output

5.2.2 Specific alternative embodiments

5.2.3 Specific examples of optimizations and configurations for inverted and non-inverted differential designs

5.3 Smoothing the down-converted signal

5.4 Impedance matching

5.5 Open resonant circuits and resonance structures

5.6 Concepts for charge and energy transfer

5.7 Optimizing and adjusting the non-minor pulse width/duration (time window)

5.7.1 Changing the input and output impedances

5.7.2 Real-time pulse width control

5.8 Adding a redirection network

,W.2

5.9 Modifying the energy transfer signal using feedback

5.10 Other implementations

6. Exemplary power transfer buck converters

IV. Further embodiments

V. Summary

1. General terminology

For explanation, the operation of the invention is often illustrated by flow charts, e.g., by flow chart 1201 in Fig. 12A. However, the use of flow charts is for illustration only and is not to be understood in a limiting sense. For example, the invention is not limited to the embodiment(s) of operation illustrated by the flow charts. Instead, alternative modes of operation will be apparent to those skilled in the art based on the present description. Furthermore, the use of flow charts should not be understood as limiting the invention to a discrete or digital mode of operation. In practice, as will be apparent to those skilled in the art based on the present description, the invention may be implemented by a discrete or continuous mode of operation, or a combination thereof. Furthermore, the control flow illustrated by the flow charts is for illustration only. As will be apparent to those skilled in the art, other control flows are possible within the scope of the present invention. Furthermore, the sequence of steps may be different in different embodiments.

This section describes various terms used in the present application. The description of this section is for illustration and understanding purposes only and is not intended to be limiting. The meaning of these terms will be apparent to those skilled in the art based on the entirety of this specification. These terms are discussed in further detail throughout the specification where appropriate.

As used herein, the term "modulated carrier signal" refers to a carrier signal that is modulated by a baseband signal.

As used herein, the term "unmodulated carrier signal" means a signal having an amplitude that oscillates at a substantially uniform frequency and phase.

As used herein, the term "baseband signal" refers to an information signal including, for example, analog information signals, digital information signals, and direct current (DC) information signals.

The term "carrier signal" as used herein refers to modulated carrier signals and unmodulated carrier signals unless otherwise specified.

As used herein, the term "electromagnetic (EM) signal" refers to a signal in the EM spectrum. The EM spectrum includes all frequencies greater than zero hertz. EM signals generally comprise waves characterized by changes in the electric and magnetic fields. Such waves may propagate in any medium, both natural and artificial, e.g., air, space, a wire, a cable, liquids, a waveguide, a microstrip, a stripline, an optical fiber, etc. Unless otherwise stated, all signals discussed herein are EM signals, even if not explicitly labeled as such.

As used herein, the term "intermediate frequency (IF) signal" means an EM signal that is substantially similar to another EM signal except that the IF signal has a lower frequency than the other signal. An IF signal frequency can be any frequency above 0 Hz. Unless otherwise specified, the terms low-

re frequency, intermediate frequency, and IF are used interchangeably herein.

As used herein, the term "analog signal" refers to a signal that is constant or continuously varying, as opposed to a signal that changes between discrete states.

As used herein, the term "baseband signal" means a frequency band occupied by any general information signal to be transmitted and/or received.

As used herein, the term "baseband signal" means any general information signal to be transmitted and/or received.

As used herein, the term "carrier frequency" refers to the frequency of a carrier signal. Typically, this is the center frequency of a transmitted signal, which is generally modulated.

As used herein, the term "carrier signal" means an EM wave having at least one characteristic that is variable by modulation so that it can carry information by modulation.

The term "demodulated baseband signal" as used herein means a signal obtained by processing a modulated signal. In some cases, the demodulated baseband signal is obtained, for example, by demodulating a modulated intermediate frequency (IF)
signal obtained by down-converting a modulated carrier signal. In another case, this is a signal obtained by a combined down-conversion and demodulation step.

As used herein, the term "digital signal" refers to a signal that changes between discrete states, as opposed to a continuous signal.

For example, the voltage of a digital signal can switch between discrete levels.

The term "electromagnetic (EM) spectrum" refers to a spectrum that includes waves characterized by changes in the electric and magnetic fields. Such waves can propagate in any natural or artificial transmission medium, such as air, space, a wire, a cable, a liquid, a waveguide, a microstrip, a stripline, an optical fiber, etc. The EM spectrum includes all frequencies greater than zero hertz.

As used herein, the term "electromagnetic (EM) signal" refers to a signal in the EM spectrum. It is also commonly referred to as an EM wave. Unless otherwise stated, all signals discussed herein are EM signals, even if they are not explicitly labeled as such.

As used herein, the term "modulating baseband signal" refers to any general information signal used to modulate an oscillating signal or a carrier signal.

1.1 Modulation

It is often advantageous to transmit electromagnetic (EM) signals at higher frequencies. These include baseband signals, e.g., digital data information signals and analog information signals. A baseband signal can be upconverted to a higher frequency EM signal using the baseband signal to modulate a higher frequency carrier signal F _c . When such a baseband signal is used in this way, it is referred to herein as a modulating baseband signal F _MB .

By modulation, changes are imposed on the carrier signal F _c which represent information in the modulating baseband signal F _MB . The changes can be amplitude changes, frequency changes, phase changes, etc. or a combination thereof. The resulting signal is referred to herein as modulated carrier signal F _MC . The modulated carrier signal F _MC comprises the carrier signal F _c modulated by the modulating baseband signal F _MB , e.g.:

F _MB combined with Fc "^ F _M c

The modulated carrier signal F _MC oscillates at or near the frequency of the carrier signal F _c and can therefore be transmitted efficiently.

Fig. 1 shows an exemplary modulator 110, wherein the carrier signal F _c is modulated by the modulating baseband signal F _MB , thereby generating the modulated carrier signal F _M c .

The modulating baseband signal F _MB can be an analog baseband signal, a digital baseband signal or a combination thereof.

Fig. 2 shows the modulating baseband signal F _MB as an exemplary analog modulating baseband signal 210. The exemplary analog modulating baseband signal 210 may represent any type of analog information, e.g., voice data, music data, image data, etc. The amplitude of the analog modulating baseband signal 210 varies over time.

Digital information has multiple discrete states. For ease of description, digital information signals are discussed below as signals having two discrete states. However, the invention is not limited to this embodiment.

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Fig. 3 shows the modulating baseband signal F _MB as an exemplary digital modulating baseband signal 310. The digital modulating baseband signal 310 may represent any type of digital data, e.g., digital computer information and digitized analog information. The digital modulating baseband signal 310 has a first state 312 and a second state 314. In a first embodiment, the first state 312 represents a binary state 0 and the second state 314 represents a binary state 1. Alternatively, the first state 312 represents the binary state 1 and the second state 314 represents the binary state 0. Throughout the remainder of the description, the former rule is followed, according to which the first state 312 represents the binary state zero and the second state 314 represents the binary state one. However, the invention is not limited to this embodiment. The first state 312 is therefore referred to herein as the low state and the second state 314 as the high state.

The digital modulating baseband signal 310 may change between the first state 312 and the second state 314 at a data or baud rate measured in bits/s.

The carrier signal F _c is modulated by the modulating baseband signal F _MB by any modulation method, e.g., amplitude modulation (AM), frequency modulation (FM), phase modulation (PM), etc., or any combination thereof. Examples of amplitude modulation, frequency modulation, and phase modulation of the analog modulating baseband signal 210 and the digital modulating baseband signal 310 with respect to the carrier signal F _c are described below. The examples are used to assist in describing the invention. The invention is not limited to or by the examples.

Fig. 4 shows the carrier signal F _c as a carrier signal 410. In the example of Fig. 4, the carrier signal 410 is shown as a 900 MHz carrier signal. Alternatively, the carrier signal 400 may have any other frequency. Example modulation schemes are shown below using signal examples shown in Figs. 2, 3 and 4.

1.1.1 Amplitude modulation

In amplitude modulation (AM), the amplitude of the modulated carrier signal F _MC is a function of the amplitude of the modulating baseband signal F _MB . Figures 5A-5C show examples of timing diagrams for amplitude modulation of the carrier signal 410 with the analog modulating baseband signal 210. Figures 6A-6C show examples of timing diagrams for amplitude modulation of the carrier signal 410 with the digital modulating baseband signal 310.

Fig. 5A shows the analog modulating baseband signal 210. Fig. 5b shows the carrier signal 410. Fig. 5C shows an analog AM carrier signal 516 that is generated when the carrier signal 410 is amplitude modulated using the analog modulating baseband signal 210. The term "analog AM carrier signal" is used herein to indicate that the modulating baseband signal is an analog signal.

The analog AM carrier signal 516 oscillates at the frequency of the carrier signal 410. The amplitude of the analog AM carrier signal 516 follows the amplitude of the analog modulating baseband signal 210, i.e., the information contained in the analog modulating baseband signal 210 is maintained in the analog AM carrier signal 516.

:34;

Fig. 6A shows the digital modulating baseband signal 310. Fig. 6B shows the carrier signal 410. Fig. 6C shows a digital AM carrier signal 616 that is generated when the carrier signal 410 is amplitude modulated using the digital modulating baseband signal 310. The term "digital AM carrier signal" is used herein to indicate that the modulating baseband signal is a digital signal.

The digital AM carrier signal 616 oscillates at the frequency of the carrier signal 4W. The amplitude of the digital AM carrier signal 616 follows the amplitude of the digital modulating baseband signal 310, i.e., the information contained in the digital modulating baseband signal 310 is maintained in the digital AM carrier signal 616. As the digital modulating baseband signal 310 changes states, the digital AM signal 616 changes amplitudes. Digital amplitude modulation is often referred to as amplitude shift keying (ASK), with the two terms being used interchangeably throughout this specification.

1.1.2 Frequency modulation

In frequency modulation (FM), the frequency of the modulated carrier signal F _MC changes as a function of the amplitude of the modulating baseband signal F _MB . Figures 8A -8C show examples of timing diagrams for the frequency modulation of the carrier signal 410 with the analog modulating baseband signal 210. Figures 8A-8C show examples of timing diagrams for the frequency modulation of the carrier signal 410 with the digital modulating baseband signal 3iO.

Fig. 7A shows the analog modulating baseband signal 210. Fig. 7B shows the carrier signal 410. Fig. 7C shows a

analog FM carrier signal 716 that is generated when the carrier signal 410 is frequency modulated using the analog modulating baseband signal 210. The term "analog FM carrier signal" is used herein to indicate that the modulating baseband signal is an analog signal.

The frequency of the analog FM carrier signal 716 changes as a function of amplitude changes of the analog baseband signal 210. In the example shown, the frequency of the analog FM carrier signal 716 changes proportionally to the amplitude of the analog modulating baseband signal 210. Therefore, at a time t1, the amplitude of the analog baseband signal 210 and the frequency of the analog FM carrier signal 716 are maximum. At a time t3, the amplitude of the analog baseband signal 210 and the frequency of the analog FM carrier signal 716 are minimum.

The frequency of the analog FM carrier signal 716 is typically centered about the frequency of the carrier signal 410. Therefore, at time t2, for example, when the amplitude of the analog baseband signal 210 is at the center, represented herein by zero volts, the frequency of the analog FM carrier signal 716 is substantially equal to the frequency of the carrier signal 410.

Fig. 8A shows the digital modulating baseband signal 310. Fig. 8B shows the carrier signal 410. Fig. 8C shows a digital FM carrier signal 816 that is generated when the carrier signal 410 is frequency modulated using the digital modulating baseband signal 310. The term "digital FM carrier signal" is used herein to indicate that the modulating baseband signal is a digital signal.

The frequency of the digital FM carrier signal 816 changes as a function of amplitude changes of the digital mo-

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modulating baseband signal 310. In the example shown, the frequency of the digital FM carrier signal 816 changes proportionally to the amplitude of the digital modulating baseband signal 310. Therefore, between times t0 and ti and between times t2 and t4, when the amplitude of the digital baseband signal 310 is in the second state with the higher amplitude value, the frequency of the digital FM carrier signal 816 is maximum. Between times ti and t2, when the amplitude of the digital baseband signal 310 is in the first state with the lower amplitude value, the frequency of the digital FM carrier signal 816 is minimum. Digital frequency modulation is often referred to as frequency shift keying (FSK), with both terms being used interchangeably throughout the description.

Typically, the frequency of the digital FM carrier signal 816 is centered about the frequency of the carrier signal 410, and the maximum and minimum frequencies are equally offset from the center frequency. Other embodiments may be used, but for ease of description, this rule is followed.

1.1.3 Phase modulation

In phase modulation (PM), the phase of the modulated carrier signal F _MC changes as a function of the amplitude of the modulating baseband signal F _MB . Figures 9A-9C show examples of timing diagrams for the phase modulation of the carrier signal 410 with the analog modulating baseband signal 210. Figures 10A-10C show examples of timing diagrams for the phase modulation of the carrier signal 410 with the digital modulating baseband signal 310.

♦··*■ ···♦ ■ .&igr;

Fig. 9A shows the analog modulating baseband signal 210. Fig. 9B shows the carrier signal 410. Fig. 9C shows an analog PM carrier signal 916 generated by phase modulating the carrier signal 410 with the analog modulating baseband signal 210. The term "analog PM carrier signal" is used herein to indicate that the modulating baseband signal is an analog signal.

In general, the frequency of the analog PM carrier signal 916 is substantially equal to the frequency of the carrier signal 410. However, the phase of the analog PM carrier signal 916 changes with amplitude changes of the analog modulating baseband signal 210. For relative comparison, the carrier signal 410 is shown by a dashed line in FIG. 9C.

The phase of the analog PM carrier signal 916 changes as a function of amplitude changes of the analog baseband signal 210. In the example shown, the phase of the analog PM carrier signal 916 is delayed by a change amount determined by the amplitude of the analog baseband signal 210. For example, when the amplitude of the analog baseband signal 210 is maximum at a time t1, the analog PM carrier signal 916 is in phase with the carrier signal 410. Between times t1 and t3, when the amplitude of the analog baseband signal 210 decreases to a minimum amplitude, the phase of the analog PM carrier signal 916 is delayed with respect to the phase of the carrier signal 410 until it reaches a maximum phase shift at a time t3. In the example shown, the phase change is shown as a phase change of approximately 180 degrees. Any suitable phase change may be used, varied in any way as a function of the baseband signal.

• · I COl > t &igr;

Fig. 10A shows the digital modulating baseband signal 310. Fig. 10B shows the carrier signal 410. Fig. 10C shows a digital PM carrier signal 1016 generated by phase modulating the carrier signal 410 with the digital baseband signal 310. The term "digital PM carrier signal" is used herein to indicate that the modulating baseband signal is a digital signal.

The frequency of the digital PM carrier signal 1016 is equal to the frequency of the carrier signal 410. The phase of the digital PM carrier signal 1016 changes as a function of amplitude changes of the digital baseband signal 310. In the example shown, when the digital baseband signal 310 is in the first state 312, the digital carrier signal 1016 is out of phase with the carrier signal 410. When the digital baseband signal 310 is in the second state 314, the digital PM carrier signal 1016 is in phase with the carrier signal 410. Therefore, between times t1 and t2, when the amplitude of the digital baseband signal 310 is in the first state 312, the digital PM carrier signal 1016 is out of phase with the carrier signal 410. Between times t0 and t1 and between times t2 and t4, when the amplitude of the digital baseband signal 310 is in the second state 314, the digital PM carrier signal 1016 is in phase with the carrier signal 410.

In the example shown, the phase shift between times ti and t3 is shown as a phase shift of about 180 degrees. However, any phase shift can be used, varied in any way as a function of the baseband signal. Digital phase modulation is often referred to as phase shift keying (PSKj), and the two terms are used interchangeably throughout the specification.

• '* MO» > r I *

1.2 Demodulation

When the modulated carrier signal F _MC is received, it can be demodulated to extract the modulating baseband signal F _MB . However, due to the typically high frequency of the modulated carrier signal F _MC, it is generally not practical to demodulate the baseband signal F _MB directly from the modulated carrier signal F _MC . Instead, the modulated carrier signal F _MC must be down-converted to a lower frequency signal containing the original modulating baseband signal.

When a modulated carrier signal is down-converted to a lower frequency signal, the lower frequency signal is referred to herein as an intermediate frequency (IF) signal F _IF . The IF signal F _IF oscillates at any frequency or in any frequency band below the frequency of the modulated carrier signal F _MC . The down-conversion of the signal F _MC to the signal F _IF is represented by:

After the signal F _MC is down-converted to the modulated IF signal F _IF , F _IF can be demodulated into a baseband signal F _DMB as given by:

FiF -^ F _DMB

F _DMB is intended to be substantially similar to the modulating baseband signal F _MB , thereby illustrating that the modulating baseband signal F _MB is substantially recoverable.

·

Throughout the description, it is assumed that the present invention can be implemented with any type of EM signal, e.g., modulated carrier signals and unmodulated carrier signals. The above examples of modulated carrier signals are for illustrative purposes only. The examples can be modified in many ways. For example, a carrier signal can be modulated with several of the modulation types described above. A carrier signal can also be modulated with several baseband signals, e.g., analog baseband signals, digital baseband signals, and combinations of analog and digital baseband signals.

2. Overview of the invention

Conventional signal processing techniques follow the Nyquist theorem, which states that to accurately reproduce or recover a sampled signal, the signal must be sampled at a rate greater than twice the frequency of the sampled signal. When a signal is sampled at a rate less than twice the frequency or at exactly twice the frequency of the signal, it is called undersampling or aliasing the signal. Therefore, in conventional signal processing, undersampling and aliasing are avoided in order to accurately reproduce a sampled signal.

2.1 Aspects of the invention

Contrary to conventional knowledge, the present invention provides a method and system for down-converting an electromagnetic (EM) signal by aliasing the EM signal. Aliasing is generally

my represented in Figures 4 5A by the reference numeral 4 502.

By using a carrier and aliasing the carrier at an aliasing rate, the carrier can be down-converted to lower frequencies according to the invention. One aspect of the present invention is that for sufficient reproduction it is not the carrier that is of interest, but the lower frequency baseband signal. The frequency component of the baseband signal satisfies the Nyquist criterion, although its carrier may be aliased, so that the baseband information is sufficiently reproducible.

12 shows a flow chart 1201 illustrating a method of aliasing a WEM signal to produce a down-converted signal. Processing begins in step 1202 where the EM signal is received. In step 1204, an aliased signal is received at an aliasing rate. In step 1206, the EM signal is aliased to down-convert the EM signal. As used herein, the term aliasing refers to both down-converting an EM signal by undersampling the EM signal at an aliasing rate and down-converting an EM signal by transferring energy from the EM signal at the aliasing rate. These concepts are described below.

Fig. 13 shows a block diagram of a general aliasing system 1302 with an aliasing module 1306. In one embodiment, the aliasing system 1302 operates according to the flow diagram 1201. For example, in step 1202, the aliasing module 1306 receives an EM signal 1304. In step 1204, the aliasing module 1306 receives an aliasing signal 1310. In step 1206, the EM signal 1304 is downconverted in the aliasing module 1306 into a downconverted signal 1308. The general aliasing system 1302 can also be used.

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used to implement any of flowcharts 1207, 1213, and 1219.

In one embodiment, the EM signal is down-converted to an intermediate frequency (IF) signal in accordance with the invention. Figure 12B shows a flow diagram 1207 illustrating a method of undersampling the EM signal at an aliasing rate to down-convert the EM signal to an IF signal. Processing begins in step 1208 where an EM signal is received. In step 1210, an aliasing signal is received at an aliasing rate F^. In step 1212, the EM signal is sub-sampled at the aliasing rate to down-convert the EM signal to an IF signal.

In another embodiment, the EM signal is downconverted to a demodulated baseband information signal in accordance with the invention. Figure 12C shows a flow chart 1213 illustrating a method for downconverting the EM signal to a demodulated baseband signal. Processing begins in step 1214 where an EM signal is received. In step 1216, an aliasing signal having an aliasing rate F^ is received. In step 1218, the EM signal is downconverted to a demodulated baseband signal. The demodulated baseband signal can be processed without further downconversion or demodulation.

In another embodiment, the EM signal is a frequency modulated (FM) signal that is down-converted to a non-FM signal, such as a phase modulated (PM) signal or an amplitude modulated (AM) signal. Figure 12D shows a flowchart 1219 illustrating a method for down-converting the FM signal to a non-FM signal. Processing begins in step 1220 where an EM signal is received. In step 1222, an aliasing signal is received at an aliasing rate. In step 1224, the FM signal is down-converted to a non-FM signal.

According to the invention, any type of EM signal is down-converted, e.g. a modulated carrier signal or an unmodulated carrier signal. For ease of description, the invention will be described below using modulated carrier signals as examples. It will be apparent to those skilled in the art from the present description and examples that the invention can also be implemented in such a way that signals other than carrier signals can be down-converted. The invention is not limited to the exemplary embodiments described above.

In one embodiment, the down-conversion is realized by undersampling an EM signal. This is described generally in Section 1.2.2 below and in detail in Section II and its subsections. In another embodiment, the down-conversion is realized by transferring non-trivial amounts of energy from an EM signal. This is described generally in Section 1.2.3 below and in detail in Section III.

2.2 Downconversion by undersampling

As used herein, the term "aliasing" refers to both down-converting an EM signal by undersampling the EM signal at an aliasing rate and down-converting an EM signal by transferring energy from the EM signal at the aliasing rate. Methods of undersampling an EM signal to down-convert the EM signal are described below in overview. Figure 14A shows a flowchart 1401 illustrating a method of undersampling the EM signal at an aliasing rate to down-convert the EM signal. Processing begins in step 1402 where an EM signal is received.

In step 1404, an undersampled signal is received at an aliasing rate. In step 1406, the EM signal is subsampled at the aliasing rate to downconvert the EM signal.

The downconversion by undersampling is illustrated in Figure 45A by reference numeral 4504 and is described in more detail in Section II.

2.2.1 Downconversion to an intermediate frequency (IF) signal

In one embodiment, the EM signal is subsampled at an aliasing rate to down-convert the EM signal to a lower frequency or intermediate frequency (IF) signal. The EM signal may be a modulated carrier signal or an unmodulated carrier signal. In an exemplary example, a modulated carrier signal F _MC is down-converted to an IF signal F _IF .

14B shows a flowchart 1407 illustrating a method of undersampling the EM signal at an aliasing rate to down-convert the EM signal to an IF signal. Processing begins at step 1408 where an EM signal is received. At step 1410, an undersampling signal at an aliasing rate is received. At step 1412, the EM signal is undersampled at the aliasing rate to down-convert the EM signal to an IF signal.

This embodiment is shown generally by reference numeral 4508 in Figure 45B and is described in Section II.1.

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I * J ·

2.2.2 Direct-to-data downconversion

In another embodiment, an EM signal is directly downconverted to a demodulated baseband signal by undersampling the EM signal at an aliasing rate (direct-to-data downconversion). The EM signal may be a modulated EM signal or an unmodulated EM signal. In an exemplary embodiment, the EM signal is a modulated carrier signal F _MC that is directly downconverted to a demodulated baseband signal F _DMB .

14C shows a flowchart 1413 illustrating a method of undersampling the EM signal at an aliasing rate to down-convert the EM signal directly to a demodulated baseband signal. Processing begins at step 1414 where an EM signal is received. At step 1416, an undersampled signal at an aliasing rate is received. At step 1418, the EM signal is undersampled at the aliasing rate to down-convert the EM signal directly to a baseband information signal.

This embodiment is shown generally by reference numeral 4510 in Figure 45B and is described in Section II.2.

2.2.3 Modulation implementation

In another embodiment, a frequency modulated (FM) carrier signal F _FMC is converted into a non-FM signal F _(NON -fm) by undersampling the FM carrier signal F _FMC .

Fig. 14D shows a flowchart 1419 illustrating a method for undersampling an FM signal to convert the

FM signal to a non-FM signal. Processing begins in step 1420 where the FM signal is received. In step 1422, the FM signal is subsampled at an aliasing rate to convert the FM signal to a non-FM signal. For example, the FM signal may be subsampled to convert it to a PM signal or an AM signal.

This embodiment is shown generally by reference numeral 4512 in Figure 45B and is described in Section II.3.

2.3 Downconversion through energy transfer

As used herein, the term "aliasing" refers to both down-converting an EM signal by under-sampling the EM signal at an aliasing rate and down-converting an EM signal by transferring non-trivial amounts of energy from the EM signal at the aliasing rate. Methods for transferring energy from an EM signal to down-convert the EM signal are described in overview below. More detailed descriptions are provided in Section III.

46A shows a flow chart 4601 illustrating a method for transferring energy from the EM signal at an aliasing rate to down-convert the EM signal. Processing begins at step 4602 where an EM signal is received. At step 4604, an energy transfer signal having an aliasing rate is received. At step 4606, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal.

The down conversion by energy transfer is illustrated in Figure 45A by reference numeral 4506 and is described in more detail in Section III.

UL. ,:

2.3.1 Downconversion to an intermediate frequency (IF) signal

In one embodiment, the EM signal is down-converted to a lower frequency or intermediate frequency (IF) signal by transferring energy from the EM signal at an aliasing rate. The EM signal may be a modulated carrier signal or an unmodulated carrier signal. In an exemplary example, a modulated carrier signal F _MC is down-converted to an IF signal Fi _F .

Fmc ■> F _1F

Figure 46B shows a flow chart 4607 illustrating a method for transferring energy from the EM signal at an aliasing rate to down-convert the EM signal to an IF signal. Processing begins at step 4608 where an EM signal is received. At step 4610, an energy transfer signal having an aliasing rate is received. At step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to an IF signal.

This embodiment is shown generally by reference numeral 4514 in Figure 45B and is described in more detail in Section III.I.

2.3.2 Direct-to-data downconversion

In another embodiment, the EM signal is down-converted to a demodulated baseband signal by transferring energy from the EM signal at an aliasing rate. This embodiment is referred to herein as direct-to-data down-conversion. The EM signal may be a modulated carrier signal or an unmodulated carrier signal.

In an exemplary example, the EM signal is the modulated carrier signal F _MC and is directly downconverted into a demodulated baseband signal F _DMB .

Fmc "^ F _DMB

Figure 46C shows a flow chart 4613 illustrating a method for transferring energy from the EM signal having an aliasing rate to directly down-convert the EM signal to a demodulated baseband signal. Processing begins at step 4614 where an EM signal is received. At step 4616, an energy transfer signal having an aliasing rate is received. At step 4618, energy is transferred from the EM signal at the aliasing rate to directly down-convert the EM signal to a baseband signal.

This embodiment is illustrated generally by reference numeral 4516 in Figure 45B and is described in more detail in Section III.2.

2.3.3 Modulation implementation

In another embodiment, a frequency modulated (FM) carrier signal Fj-mc is down-converted to a non-FM signal F _(N0N _ _FM) by transferring energy from the FM carrier signal F^c at an aliasing rate.

The FM carrier signal F _FM c may be converted, for example, into a phase modulated (PM) signal or into an amplitude modulated (AM) signal. Fig. 46D shows a flow chart 4619 illustrating a method for transferring energy from an FM signal to convert it to a non-FM signal. In step 4620, the FM signal is received. In step 4622, an energy transfer signal is received which has a

· * 4 9* *

aliasing rate. In Fig. 46D, in step 4612, energy is transferred from the FM signal to convert it to a non-FM signal. For example, energy may be transferred from an FSK signal to convert it to a PSK signal or to an ASK signal.

This embodiment is illustrated generally by reference numeral 4518 in Figure 45B and is described in more detail in Section III.3.

2.4 Determination of the aliasing rate

According to the definition of aliasing, the aliasing rate is equal to or less than twice the frequency of the EM carrier signal. Preferably, the aliasing rate is substantially lower than the frequency of the carrier signal. Preferably, the aliasing rate is more than twice the highest frequency component of the modulating baseband signal F _MB to be reproduced. The above requirements are shown in Eq. (1).

2-Fmc ^ Far > 2 (highest frequency component of Fmb) Eq. (1)

That is, by using a carrier and aliasing the carrier with an aliasing rate, the carrier can be down-converted to lower frequencies according to the invention. An important aspect of the invention is that for sufficient reproduction, it is not the carrier that is of interest, but instead the lower frequency baseband signal. The frequency component of the baseband signal satisfies the Nyquist criterion, although its carrier may be aliased, so that the baseband information is sufficiently reproducible.

·5&thetas;

either as the modulating IF carrier signal Fjp or as the demodulated direct-to-data baseband signal F _DMB .

The inventive relationships between the frequency of an EM carrier signal, the aliasing rate and the intermediate frequency of the down-converted signal are shown in Eq. (2).

Fc = II-FaRIFiF Eq. (2)

Where:

F _{c is} the frequency of the EM carrier signal to be aliased;
F _{AR is} the aliasing rate;

η is a harmonic or subharmonic of the aliasing rate (in general, η =0.5, 1, 2, 3, 4, ...); and FiF is the intermediate frequency of the down-converted signal.

As (nF^) approaches F _c , F _IF approaches zero. This is a special case where an EM signal is downconverted directly to a demodulated baseband signal. This special case is referred to herein as direct-to-data downconversion. Direct-to-data downconversion is described in later sections.

The following sections present high-level descriptions, exemplary embodiments, and exemplary implementations of the foregoing and other embodiments of the invention.

3. Advantages of the invention using an exemplary conventional receiver for comparison purposes

Fig. 11 shows an exemplary conventional receiver system 1102. The conventional system 1102 is shown

to illustrate the functional differences between conventional systems and the advantages of the present invention.

The exemplary conventional receiver system 1102 receives an electromagnetic (EM) signal 1104 via an antenna 1106. The EM signal 1104 may comprise multiple EM signals, e.g., modulated carrier signals. For example, the EM signal 1104 comprises one or more radio frequency (RF) EM signals, e.g., a modulated 900 MHz carrier signal. Higher frequency RF signals, e.g., 900 MHz signals, generally cannot be processed directly by conventional signal processors. Instead, higher frequency RF signals are typically downconverted to lower frequency intermediate frequency (IF) signals for processing. The EM signal 1104 is downconverted by the receiver system 1102 into an intermediate frequency (IF) signal 1108n, which may be provided to a signal processor 1110. If the EM signal 1104 comprises a modulated carrier signal, the signal processor 1110 typically comprises a demodulator that demodulates the IF signal 1108n into a baseband information signal (demodulated baseband signal).

The receiver system 1102 includes an RF stage 1112 and one or more IF stages 1114. The RF stage 1112 receives the EM signal 1104. The RF stage 1112 includes the antenna 1106 that receives the EM signal 1104.

The one or more IF stages 1114a-1114n stepwise downconvert the EM signal 1104 to lower intermediate frequencies. Each of the one or more IF stages 1114a-1114n includes a mixer 1118a-1118n that downconverts an EM input signal 1116 to a lower frequency IF signal. The cascade arrangement of the one or more mixers 1118a-1118n downconverts the EM

Signal 1104 is gradually down-converted into a desired IF signal 1108n.

During operation, each of the one or more mixers

1118 an EM input signal 1116 with a local oscillator (LO) signal 1119 generated by a local oscillator (LO) 1120. By mixing, sum and difference signals of the EM input signal 1116 and the LO signal

1119. For example, by mixing an EM input signal 1116a having a frequency of 900 MHz with an LO signal 1119a having a frequency of 830 MHz, a sum signal having a frequency of 900 MHz + 830 MHz = 1.73 GHz and a difference signal having a frequency of 900 MHz - 830 MHz = 70 MHz are obtained.

In the example of Figure 11, the one or more mixers 1118 generate sum and difference signals for all signal components in the EM input signal 1116. For example, if the EM signal 1116a includes a second EM signal having a frequency of 760 MHz, the mixer generates a second sum signal having a frequency of 760 MHz + 830 MHz = 1.59 MHz and a second difference signal having a frequency of 830 MHz - 760 MHz = 70 MHz. In this example, by mixing two EM input signals having frequencies of 900 MHz and 760 MHz, respectively, with an LO signal having a frequency of 830 MHz, two IF signals having a frequency of 70 MHz are obtained.

In general, it is very difficult, if not impossible, to separate the two 70 MHz signals. Instead, each mixer 1118 is preceded by one or more filters 1122 and 1123 to filter out unwanted frequencies, also known as frame frequencies. The filters 1122 and 1123 can have various filter topologies and arrangements, e.g. bandpass filters, one or more highpass

&igr; &idigr;
.· .· ·53:

filter, one or more low-pass filters, combinations thereof, etc.

Typically, the one or more mixers 1118 and the one or more filters 1122 and 1123 attenuate or reduce the strength of the EM signal 1104. For example, a typical mixer reduces the EM signal strength by 8 to 12 dB. A typical filter reduces the EM signal strength by 3 to 6 dB.

Thus, the one or more filters 1123 and 1122a-1122n are preceded by one or more low-noise amplifiers (LNA) 1121 and 1124a-1124n, respectively. The low-noise amplifiers and the filters may also be arranged in reverse order. The low-noise amplifiers compensate for losses in the mixers 1118, the filters 1122 and 1123, and other components by increasing the EM signal strength prior to filtering and mixing. Typically, a gain of 15 to 20 dB is obtained from each low-noise amplifier.

However, the low-noise amplifiers require high operating power. Higher frequency low-noise amplifiers require more power than lower frequency low-noise amplifiers. If the receiver system 1102 is to be a portable system, e.g., a cell phone receiver, the low-noise amplifiers consume a significant portion of the total power.

At higher frequencies, the strength of the EM signal 1104 is reduced by impedance mismatches between the various stages. To optimize the power transmitted through the receiver system 1102, each component should be impedance matched with respect to neighboring components. Because no two components have exactly the same impedance characteristics, even for components manufactured to high tolerances, the impedance of each receiver system 1102 must be

*54

impedance matching often has to be fine-tuned individually. As a result, impedance matching in conventional receivers tends to be labor intensive. Impedance matching requires a significant amount of additional time and effort in both the design and manufacture of conventional receivers. Because many of the components, e.g. low-noise amplifiers, filters and impedance matching circuits, are highly frequency dependent, a receiver designed for one application is generally unsuitable for other applications. Instead, a new receiver must be designed, requiring new impedance matching circuits between many of the components.

Conventional receiver components are typically arranged on multiple IC substrates rather than on a single IC substrate. This is partly because no single substrate exists that is optimal for RF, IF and baseband frequencies. Other factors may include the total number of components, their different sizes and their different impedance characteristics, etc. Often, additional signal amplification is required from one chip to the next. Therefore, multiple substrate implementations incur many additional costs in addition to the cost of the ICs themselves.

Therefore, conventional receivers require many components, are difficult and time-consuming to design and manufacture, and require substantial external power to maintain sufficient signal levels. Therefore, conventional receivers are expensive to design, manufacture, and use.

In one embodiment, the present invention is implemented such that many, if not all, of the components between the antenna 1106 and the signal processor 1110 are

·

·55

an aliasing module comprising a universal frequency conversion (UFT) module. The UFT module is capable of down-converting a wide range of EM signal frequencies using very few components. The UFT module is simple to design and manufacture and requires little external power. The UFT design is easily adaptable to different frequencies or frequency ranges. For example, the UFT design can be easily impedance matched with relatively little effort. In a direct-to-data embodiment of the invention in which an EM signal is down-converted directly to a demodulated baseband signal, the invention also eliminates the need for a demodulator in the signal processor 1110.

When the invention is implemented in a receiver system, e.g., receiver system 1102, power consumption is significantly reduced and signal-to-noise ratio is significantly increased.

In one embodiment, the invention can be easily implemented and adapted for specific applications, and the impedance matching circuits can be easily calculated and implemented. Thus, when the invention is implemented as a receiver, e.g., as receiver 1102, no special expertise is required for impedance matching.

In conventional receivers, components in the IF sections comprise approximately 80 to 90% of the total components of the receiver. The UFT design eliminates the IF section(s), thereby eliminating the approximately 80 to 90% of the total components of conventional receivers.

Other advantages of the inventions include:

The invention can be implemented as a receiver with only one local oscillator;

The invention can be implemented as a receiver with only one low-frequency local oscillator;

The invention can be implemented as a receiver using few filters;

The invention can be implemented as a receiver using uniform delay filters ;

The invention can be implemented as a receiver that can change frequencies and receive different modulation formats without requiring hardware changes;

The invention can also be implemented as a frequency upconverter in an EM signal transmitter;

The invention can also be implemented as a combination of an upconverter (transmitter) and a downconverter (receiver), also referred to herein as a transceiver;

The invention can be implemented in a differential configuration, thereby obtaining larger signal-to-noise ratios;

A receiver constructed according to the invention can be implemented on a single IC substrate, e.g. on a silicon-based IC substrate;

A receiver constructed in accordance with the invention and implemented on a single IC substrate, e.g. a silicon-based IC substrate, can down-convert EM signals from frequencies in the GHz range;

A receiver constructed according to the invention has a relatively flat response over a wide frequency range. For example, a receiver constructed according to the invention suitable for operation at about 800 MHz has, in one embodiment, a relatively flat response in a frequency range from 100 MHz to 1 GHz.

hold on (i.e. plus or minus a few dB of power). This is referred to herein as a wideband receiver; and

A receiver constructed in accordance with the invention may include a plurality of user-selectable impedance matching modules, each designed for a different wide frequency band, which may be used to scan an ultra-wide frequency band.

II. Downconversion by undersampling

1. Downconversion of an EM carrier signal into an EM intermediate frequency signal by undersampling the EM carrier signal with the aliasing rate

In one embodiment of the invention, an EM signal is down-converted to an IF signal by undersampling the EM signal. This embodiment is illustrated in Fig. 45B by reference numeral 4508.

This embodiment can be implemented with modulated and unmodulated signals. The embodiment is described herein by way of example using the modulated carrier signal F _MC in Fig. 1. In the example, the modulated carrier signal F _MC is down-converted to an IF signal F _IF . The IF signal F _iF can then be demodulated by any conventional demodulation technique to obtain a demodulated baseband signal F _DMB . Those skilled in the art will appreciate from the present description that the invention can be implemented to down-convert any EM signal, eg a modulated carrier signal and an unmodulated carrier signal.

In the following sections, exemplary methods for downconverting the modulated carrier signal F _MC into the IF signal F _IF according to embodiments of the invention are described.

Also described are exemplary structural embodiments for implementing the methods. However, the invention is not limited to the specific embodiments described below. It will be apparent to those skilled in the art based on the present description that equivalent implementations, extensions, changes, modifications, etc. are possible within the scope of the present invention.

The following sections include a high-level description, example embodiments, and implementation examples.

1.1 High-level description

Through this section (including its subsections), a high-level description of the inventive downconversion of an EM signal into an IF signal Fi _F is provided. In particular, an operational method for undersampling a modulated carrier signal F _MC to downconvert it into an IF signal Fi _F is presented in a high-level description. In addition, a structural implementation of this method is presented in a high-level description. This structural implementation is described herein for illustration and is not intended to be limiting. In particular, the method described in this section may be realized using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to those skilled in the art based on the present description.

1.1.1 Description of the functionality

14B shows a flow diagram 1407 illustrating an exemplary method for undersampling an EM signal to down-convert the EM signal to an intermediate frequency signal F _IF . The exemplary method illustrated in flow diagram 1407 is an embodiment of the flow diagram 1401 of FIG. 14A.

Any and all combinations of modulation techniques are applicable to the present invention. For ease of description, the digital AM carrier signal 616 is used to provide a high-level description of how the invention works. Detailed timing diagrams and descriptions for exemplary AM, FM and PM embodiments are provided throughout the following sections. Those skilled in the art will appreciate from this description that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and an unmodulated carrier signal.

The method illustrated in flow diagram 1407 is illustrated below in a high-level description using the digital AM carrier signal 616 of FIG. 6C. The digital AM carrier signal 6116 is shown again in FIG. 15A for clarity. FIG. 15E shows a portion 1510 of the AM carrier signal 616 between times ti and t2 in an expanded scale.

Processing begins in step 1408 where an EM signal is received. Step 1408 is represented by the digital AM carrier signal 616.

In step 1410, an undersampling signal is received having an aliasing rate F^ . Fig. 15B shows an exemplary undersampling signal 1502 having a sequence

or train of pulses 1504 with slight pulse widths tending toward zero in duration. The pulses 1504 repeat at the aliasing or pulse repetition rate. Aliasing rates are discussed below.

In step 1412, the EM signal is subsampled at the aliasing rate to down-convert the EM signal to an intermediate frequency signal F _IF . When down-converting an EM signal to an IF signal, the intermediate frequency is determined by the frequency or aliasing rate of the pulses 1504.

Fig. 15C shows a staircase or step-shaped AM intermediate frequency signal 1506 produced by the down-conversion process. The AM intermediate frequency signal 1506 is similar to the AM carrier signal 616 except that the AM intermediate frequency signal 1506 has a lower frequency than the AM carrier signal 616. The AM carrier signal 615 has thereby been down-converted to the AM intermediate frequency signal 1506. The AM intermediate frequency signal 1506 can be produced by setting the aliasing rate at any frequency below the frequency of the AM carrier signal 616.

Fig. 15D shows the AM intermediate frequency signal 1506 as a filtered output signal 1508. In an alternative embodiment, the invention provides a stepped, unfiltered, or partially filtered output signal. The choice between filtered, partially filtered, and unfiltered output signals is generally determined by design, which depends on the application of the invention.

The intermediate frequency of the down-converted signal F _IF , which in this example is the AM intermediate frequency signal 1506, can be determined according to Eq. (2), which is reproduced below for simplicity.

Eq. (2)

A suitable aliasing rate F _AR can be determined in various ways. An exemplary method for determining the F _AR is presented below. Those skilled in the art will appreciate from this description how suitable aliasing rates for EM signals can also be determined for signals other than the modulated carrier signals described herein .

Flowchart 1701 in Figure 17 shows an exemplary method for determining an aliasing rate F _n R. However, a designer or application may specify that the values be determined in a different sequence than that shown. Processing begins in step 1702 where the frequency of the EM signal is determined or selected. For example, the frequency of the FM carrier signal 616 may be 901 MHz.

In step 1704, the intermediate frequency is determined or selected. This is the frequency to which the EM signal is down-converted. The intermediate frequency may be determined or selected to match a frequency requirement of a downstream demodulator. For example, the intermediate frequency may be 1 MHz.

In step 1706, the aliasing rate (or rates) at which the EM signal is down-converted to the intermediate frequency specified in step 1704 are determined.

Eq. (2) can be rewritten as Eq. (3):

π-Far = Fc ± F _[F Eq. (3)

This equation can be rewritten as Eq. (4):

_{n =} MIk Eq. (4)

Far

or in Eq. (5):

l. (5)

± Fif) can be defined as the difference value Fqiff, as shown in Eq. (6):

(Fc±F _1F ) = F _DIFF Eq. (6)

Eq. (4) can be rewritten as Eq. (7):

η = ^EE- Eq. (7)

Far

According to equation (7), it can be seen that F _DIFF is constant for a given value η and a constant value of F _AR . For the case F _DXFF = F _c - F _IF and for a constant value of F _DIFF , when F _c increases, Fi _F necessarily increases. For the case F _DIFF = F _c + F _IF and for a constant value of F _DIFF , when F _c increases, F _IF necessarily decreases. In the latter case, F _DIFF = F _c + Fif, any phase or frequency changes of Fc correspond to reversed or inverted phase or frequency changes of F _IF . This is mentioned to clarify that when F _DIFF = F _c + F _IF is used, the phase and frequency

frequency behavior of the modulated intermediate frequency signal F _IF is influenced.

Equations (2) through (7) can be solved for any valid value of η. An appropriate value of η can be determined for any given difference frequency F _DIFF and for any desired aliasing rate F _AR ( _desired ). Equations (2) through (7) can be used to identify a particular harmonic closest to the desired aliasing rate F _{AR (desired)} that will produce the desired intermediate frequency signal F _IF .

An example of determining an appropriate value η for a given difference frequency F _DIFF and for a desired aliasing rate F _AR(desired ) is shown below. For ease of description, only the case (F _c - F _IF ) is shown in the example below.

F-F F

_n _ C ¹ IF _ ¹ DIFF

F F

AR(desired) ^l AR(desired)

For example, the desired aliasing rate F _A R( _g e _Desired ) may be 140 MHz. Using the above examples where the carrier frequency is 901 MHz and the intermediate frequency is 1 MHz, an initial value of η is determined as:

901MHz-IMHz 900 ^,

η = = = 6.4

140MHz 140

The initial value 6.4 can be rounded up or down to the next valid value n, which was defined above as (0.5, 1, 2, 3, ...). In this example, the value 6.4

rounded down to 6.0, and this value is used for the case = F _DIFF in Eq. (5).

F-F

^r AR ~

901MHz-IMHz 900MHz _{1&iacgr;&Lgr;} .”,

Far= ζ = ζ = 150 MHz

That is, by undersampling a 901 MHz EM carrier signal at 150 MHz, an intermediate frequency signal with a frequency of 1 MHz is generated. If the undersampled EM carrier signal is a modulated carrier signal, the intermediate frequency signal will also have substantially the modulation. The modulated intermediate frequency signal can be demodulated by any conventional demodulation technique.

Alternatively, instead of starting from a desired aliasing rate, a list of suitable aliasing rates can be determined from the modified form of Eq. (5) by determining a solution for various values of η. Example solutions are shown below.

_ (F _c - F _ff ) _ F _DffF _ 901 MHz -1 MHz _ 900 MHz
&eegr;&eegr;&eegr;&eegr;

Solution for &eegr; =0.5, 1, 2, 3, 4, 5 and 6:

900 MHz/0.5 = 1.8 GHz (i.e., second harmonic, as represented by 2502 in Fig. 25A);
900 MHz/1 = 900 MHz (i.e., fundamental frequency, as represented by 2504 in Fig. 25B);
900 MHz/2 = 450 MHz (i.e., second subharmonic, as represented by 2506 in Fig. 25C);
900 MHz/3 = 300 MHz (i.e., third subharmonic, as represented by 2508 in Fig. 25D);
900 MHz/4 = 225 MHz (i.e., fourth subharmonic, as represented by 2510 in Fig. 25E);

900 MHz/5 = 180 MHz (i.e., fifth subharmonic, as represented by 2512 in Fig. 25F);
900 MHz/6 = 150 MHz (i.e., sixth subharmonic, as represented by 2514 in Fig. 25G);

The steps described above can be carried out in a similar manner for the case (F _c + F _IF ). The results can be compared with the results obtained in the case of (F _c - F _IF ) to determine which gives a better result for an application.

In one embodiment, according to the invention, an EM signal is down-converted to a relatively standard intermediate frequency signal with a frequency of, for example, 100 kHz to 200 MHz. In another embodiment, referred to herein as a small offset implementation, according to the invention, an EM signal is down-converted to an IF signal with a relatively low frequency of, for example, less than 100 kHz. In another embodiment, referred to herein as a large offset implementation, according to the invention, an EM signal is down-converted to an IF signal with a relatively higher frequency of, for example, more than 200 MHz.

The different offset implementations provide choices for different applications. In general, lower data rate applications can operate at lower intermediate frequencies. However, higher intermediate frequencies can support a larger amount of information for a given modulation technique.

According to the invention, a designer selects an optimal information bandwidth for an application and an optimal intermediate frequency to support the baseband signal. The intermediate frequency should be sufficiently high to

Bandwidth of the modulating baseband signal F _MB .

In general, the frequency of the down-converted IF signal decreases as the aliasing rate approaches a harmonic or subharmonic of the EM signal frequency. Similarly, the frequency of the IF signal increases as the aliasing rate moves away from a harmonic or subharmonic of the EM signal frequency.

Aliased frequencies occur above and below each harmonic of the aliasing frequency. To avoid mixing or mapping other aliasing frequencies into the band of the aliasing frequency (IF) of interest, the intermediate frequency is preferably not near half the aliasing rate.

As described in implementation examples below, an aliasing module using a universal frequency conversion (UFT)
module, a wide range of flexibility in frequency selection is provided, making it suitable for a wide range of applications. Conventional systems do not easily provide or enable this level of flexibility in frequency selection.

1.1.2 Description of the structure

Fig. 16 shows a block diagram of an undersampling system 1602 according to an embodiment of the invention. The undersampling system 1602 is an exemplary embodiment of the general aliasing system 1302 of Fig. 13. The undersampling system 1602 includes an undersampling module 1606. The undersampling module 1606 receives the EM signal 1304 and an undersampling signal 1604 comprising undersampling pulses with slight pulse widths whose

duration tending toward zero and occurring at a frequency equal to the aliasing rate F _AR . The undersampling signal 1604 is an exemplary embodiment of the aliasing signal 1310. The EM signal 1304 is subsampled by the undersampling module 1606 at the aliasing rate F _AR of the undersampling signal 1604. The undersampling system 1602 outputs a downconverted signal 1308A.

Preferably, the EM signal 1304 is undersampled by the undersampling module 1606 to downconvert it to the intermediate frequency signal F _IF in the manner illustrated in the flowchart 1407 of FIG. 14B. Other structural embodiments for carrying out the steps of the flowchart 1407 are possible within the scope of the invention. The details of the other structural embodiments will be apparent to those skilled in the art based on the description presented herein. In one embodiment, the aliasing rate F _AR of the undersampling signal 1604 is selected in the manner described in Section II.1.1.1 so that the EM carrier signal 1304 is undersampled by the undersampling module 1606 and the intermediate frequency signal FiF is generated.

The operation of the undersampling system 1602 is described below with reference to the flowchart 1407 and timing diagrams in Figures 15A-D. In step 1408, the undersampling module 1606 receives the AM signal 616 (Figure 15A). In step 1410, the undersampling module 1606 receives the undersampling signal 1502 (Figure 15B). In step 1412, the AM signal 616 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 1502 or a multiple thereof to down-convert the AM carrier signal 616 to the intermediate frequency signal 1506 (Figure 15D).

* * &igr; ■Oe &igr; > &igr;

Example implementations of the undersampling module 1606 are presented in sections 4 and 5 below.

1.2 Exemplary embodiments

In this section (and its subsections), various embodiments related to the method(s) and structure(s) described above are presented. These embodiments are described herein for illustrative purposes and are not to be taken in a limiting sense. The invention is not limited to these embodiments. It will be apparent to those skilled in the art based on the present description that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and adapted to include these alternative embodiments.

The method illustrated in flowchart 1407 of FIG. 14B for downconverting EM signal 1304 to an intermediate frequency signal F _1F may be implemented with any type of EM signal, e.g., unmodulated EM carrier signals and modulated EM carrier signals, e.g., AM, FM, PM carrier signals, etc., or any combination thereof. The processing of flowchart 1407 of FIG. 14B is described below for AM, FM, and PM carrier signals. The following exemplary descriptions are provided to illustrate the present invention. The present invention is not limited by or to the following exemplary embodiments.

1.2.1 First exemplary embodiment: amplitude modulation
1.2.1.1 Description of the functionality

The exemplary processing of the flowchart 1407 in Fig. 14B for the analog AM carrier signal 516 shown in Fig. 5C and for the digital AM carrier signal 616 shown in Fig. 6C is described below.

1.2.1.1.1 Analog AM carrier signal

A method for down-converting the analog AM carrier signal 516 in FIG. 5C to an analog AM intermediate frequency signal is described below with reference to the flow chart 1407 in FIG. 14B. The analog AM carrier signal 516 is shown again in FIG. 19A for simplicity. In this example, the analog AM carrier signal 516 oscillates at a frequency of about 901 MHz. In FIG. 19B, an analog AM carrier signal 1904 represents a portion of the analog AM carrier signal 516 on an expanded time scale.

Processing begins in step 1408 by receiving the EM signal, which is represented by the analog AM carrier signal 516 in Figure 19A.

In step 1410, an undersampling signal is received having an aliasing rate F _AR . Figure 19C shows an undersampling signal 1906 at approximately the same time scale as in Figure 19B. The undersampling signal 1906 comprises a sequence of pulses 1907 with slight pulse widths whose duration tends toward zero. The pulses 1907 repeat at the aliasing rate or pulse repetition rate determined or selected as described above.

is chosen. In general, the aliasing rate Fp^ when downconverting to an intermediate frequency signal is essentially equal to a harmonic or, more typically, a subharmonic of the difference frequency F _DIFF . In this example, the aliasing rate is about 450 MHz.

In step 1412, the EM signal is subsampled at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _IF . Step 1412 is illustrated in Fig. 19B by undersampling points 1905.

Because a harmonic of the aliasing rate is offset from an AM carrier signal 516, the undersampling points 1905 "walk through" the analog AM carrier signal 516. In this example, the undersampling points 1905 "walk through" the analog AM carrier signal 516 at approximately a rate of 1 MHz. That is, the undersampling points 1905 appear at different positions in successive cycles of the AM carrier signal 516. Therefore, different amplitude values of the analog AM signal 516 are captured by the undersampling points 1905. For example, the undersampling point 1905A has a larger amplitude value than the undersampling point 1905B.

In Figure 19D, undersampling points 1905 are correlated with voltage points 1908. In one embodiment, voltage points 1908 form an AM intermediate frequency analog signal 1910. This can be accomplished in many ways. For example, each voltage point 1908 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 19E, an AM intermediate frequency signal 1912 represents the AM intermediate frequency signal 190 after a filtering operation on a compressed time scale. Although Fig. 19E shows the AM intermediate frequency signal 1912 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The AM intermediate frequency signal 1912 is substantially similar to the AM carrier signal 516, except that the AM intermediate frequency signal 1912 has an intermediate frequency of 1 MHz. The AM intermediate frequency signal 1912 may be demodulated by any conventional AM demodulation technique.

The drawings referred to herein illustrate a frequency down-conversion method in accordance with the present invention. For example, AM intermediate frequency signal 1910 in FIG. 19D and AM intermediate frequency signal 1912 in FIG. 19E illustrate that AM carrier signal 516 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.1.1.2 Digital AM carrier signal

A method for down-converting the digital AM carrier signal 616 in FIG. 6C to a digital AM intermediate frequency signal is described below with reference to the flow chart 1407 in FIG. 14B. The digital AM carrier signal 616 is shown again in FIG. 18A for simplicity. In this example, the digital AM carrier signal 616 oscillates at a frequency of about 901 MHz. In FIG. 18B, an AM carrier signal 1804 represents a portion of the

AM carrier signal 616 from a time tO to a time ti in an expanded time scale.

Processing begins in step 1408 where an EM signal is received. This is represented by the AM carrier signal 616 in Figure 18A.

In step 1410, an undersampling signal is received having an aliasing rate F _AR . Figure 18C shows an exemplary undersampling signal 1806 at approximately the same time scale as Figure 18B. The undersampling signal 1806 comprises a train of pulses 1807 with slight pulse widths tending toward zero in duration. The pulses 1807 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F AR is substantially equal to a harmonic or more typically a subharmonic of the difference frequency F _DIFF when downconverted to an intermediate frequency signal. In this example, the aliasing rate is approximately 450 MHz.

In step 1412, the EM signal is subsampled at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _IF . Step 1412 is illustrated in Fig. 18B by undersampling points 1805.

Because a harmonic of the aliasing rate is offset from an AM carrier signal 616, the undersampling points 1805 traverse the AM carrier signal 616. That is, the undersampling points 1805 appear at different positions in successive cycles of the AM carrier signal 616. Therefore, different amplitude values of the AM signal 616 are captured by the undersampling points 1805. In this example, the undersampling points 1805 "traverse" the AM carrier signal 616 at a rate of approximately 1 MHz. Example-

For example, the undersampling point 1805A has a larger amplitude value than the undersampling point 1805B.

In Figure 18D, undersampling points 1805 are correlated with voltage points 1808. In one embodiment, voltage points 1805 form an AM intermediate frequency signal 1810. This can be accomplished in many ways. For example, each voltage point 1808 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 18E, an AM intermediate frequency signal 1812 represents the AM intermediate frequency signal 1810 after a filtering operation on a compressed time scale. Although Fig. 18E shows the AM intermediate frequency signal 1812 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The AM intermediate frequency signal 1812 is substantially similar to the AM carrier signal 616, except that the AM intermediate frequency signal 1812 has an intermediate frequency of 1 MHz. The AM intermediate frequency signal 1812 may be demodulated by any conventional AM demodulation technique.

The drawings referred to herein illustrate a frequency downconversion method according to the present invention. For example, AM intermediate frequency signal 1810 in FIG. 18D and AM intermediate frequency signal 1812 in FIG. 18E illustrate that AM carrier signal 616 has been successfully downconverted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.1.2 Description of the structure

The operation of the undersampling system 1602 for the analog AM carrier signal 516 is described below with reference to the flowchart 1407 and the timing diagrams of Figures 19A-E. In step 1408, the undersampling module 1606 receives the AM carrier signal 516 (Figure 19A). In step 1410, the undersampling module 1606 receives the undersampling signal 1906 (Figure 19C). In step 1412, the AM carrier signal 516 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 1906 to downconvert it to the AM intermediate frequency signal 1912 (Figure 19E).

The operation of the undersampling system 1602 for the digital AM carrier signal 616 is described below with reference to the flowchart 1407 and the timing diagrams of Figures 18A-E. In step 1408, the undersampling module 1606 receives the AM carrier signal 616 (Figure 18A). In step 1410, the undersampling module 1606 receives the undersampling signal 1806 (Figure 18C). In step 1412, the AM carrier signal 616 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 1806 to downconvert it to the AM intermediate frequency signal 1812 (Figure 18E).

Implementation examples of the undersampling module 1606 are presented in sections 4 and 5 below.

1.2.2 Second exemplary embodiment: frequency modulation
1.2.2.1 Description of the functionality

The following illustrates the operation of the exemplary processing of the flow chart 1407 in Fig. 14B for the analog FM carrier signal 716 shown in Fig. 7C and for the digital FM carrier signal 816 shown in Fig. 8C.

1.2.2.1.1 Analog FM carrier signal

A method for down-converting the analog FM carrier signal 716 to an analog FM intermediate frequency signal is described below with reference to the flow chart 1407 in FIG. 14B. The analog FM carrier signal 716 is shown again in FIG. 20A for simplicity. In this example, the analog FM carrier signal 716 oscillates at a frequency of about 901 MHz. In FIG. 20B, an FM carrier signal 2004 represents a portion of the analog FM carrier signal 716 from a time ti to a time t3 on an expanded time scale.

Processing begins in step 1408 where an EM signal is received. This is represented by the FM carrier signal 716 in Fig. 20A.

In step 1410, an undersampling signal is received having an aliasing rate F _AR . Fig. 20C shows an undersampling signal 2006 at approximately the same time scale as in Fig. 20B. The undersampling signal 2006 comprises a sequence of pulses 2007 with slight pulse widths whose duration tends toward zero. The pulses 2007 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate is

F _AR when downconverted to an intermediate frequency signal of a harmonic or, more typically, a subharmonic of the difference frequency Fdiff ^is essentially equal. In this example, where the FM carrier signal 716 is centered at 901 MHz, the aliasing rate is approximately 450 MHz.

In step 1412, the EM signal is subsampled at the aliasing rate to down-convert the EM signal to the intermediate frequency signal Fj _F. Step 1412 is represented in Fig. 20B by undersampling points 2005.

Because a harmonic of the aliasing rate is offset from the FM carrier signal 716, the undersampling points 2005 appear at different positions in successive cycles of the subsampled FM carrier signal 716. That is, the undersampling points 2005 traverse the signal 716. Therefore, different amplitude values of the FM carrier signal 716 are detected by the undersampling points 2005.

In Fig. 20D, the undersampling points 2005 are correlated with voltage points 2008. In one embodiment, the voltage points 2005 form an analog FM intermediate frequency signal 2010. This can be accomplished in many ways. For example, each voltage point 2008 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 2OE, an FM intermediate frequency signal 2012 represents the FM intermediate frequency signal 2010 after a filtering process in a compressed time scale. Although Fig. 2OE shows the FM intermediate frequency signal 2012 as a filtered output signal, according to the invention the output signal does not have to

filtered or smoothed. Instead, the output signal can be adapted for different applications.

The FM intermediate frequency signal 2012 is substantially similar to the FM carrier signal 716, except that the FM intermediate frequency signal 2012 has an intermediate frequency of 1 MHz. The FM intermediate frequency signal 2012 may be demodulated by any conventional FM demodulation technique.

The drawings referred to herein illustrate a frequency down-conversion method according to the invention. For example, FM intermediate frequency signal 2010 in FIG. 20D and FM intermediate frequency signal 2012 in FIG. 20E show that FM carrier signal 716 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.2.1.2 Digital FM carrier signal

A method for down-converting the digital FM carrier signal 816 to a digital FM intermediate frequency signal will now be described with reference to the flow chart 1407 in FIG. 14B. The digital FM carrier signal 816 is shown again in FIG. 21A for simplicity. In this example, the digital FM carrier signal 816 oscillates at a frequency of about 901 MHz. In FIG. 21B, an FM carrier signal 2104 represents a portion of the FM carrier signal 816 from a time ti to a time t3 on an expanded time scale.

Processing begins in step 1408 by receiving the EM signal, which is represented by the FM carrier signal 816 in Figure 21A.

In step 1410, an undersampling signal is received having an aliasing rate F _AR . Figure 21C shows an undersampling signal 2106 at approximately the same time scale as in Figure 21B. The undersampling signal 2106 comprises a train of pulses 2107 with slight pulse widths tending toward zero in duration. The pulses 2107 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F _AR is substantially equal to a harmonic or more typically a subharmonic of the difference frequency F _DIFF when downconverted to an intermediate frequency signal. In this example, where the FM carrier signal 816 is centered at 901 MHz, an aliasing rate of approximately 450 MHz is selected, which is a subharmonic of 900 MHz offset by 1 MHz from the center frequency of the FM carrier signal 816.

In step 1412, the EM signal is subsampled at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _IF . Step 1412 is illustrated in Fig. 21B by undersampling points 2105.

Because a harmonic of the aliasing rate is offset from the FM carrier signal 816, the undersampling points 2105 appear at different positions in successive cycles of the FM carrier signal 816. That is, the undersampling points 2105 traverse the signal 816. Therefore, different amplitudes of the digital AM signal 516 are detected by the undersampling points 2105.

In Fig. 21D, the undersampling points 2105 are correlated with voltage points 2108. In one embodiment, the voltage points 2108 form a digital FM intermediate frequency signal 2110. This can be realized in many ways. For example, each voltage point

2108 at a relatively constant level until the next voltage point is received. This produces a stepped output signal which may be smoothed or filtered as described below.

In Fig. 21E, an FM intermediate frequency signal 2112 represents the FM intermediate frequency signal 2110 after a filtering operation on a compressed time scale. Although Fig. 21E shows the FM intermediate frequency signal 2112 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The FM intermediate frequency signal 2112 is substantially similar to the FM carrier signal 816, except that the FM intermediate frequency signal 2112 has an intermediate frequency of 1 MHz. The FM intermediate frequency signal 2112 may be demodulated by any conventional FM demodulation technique.

The drawings referred to herein illustrate a frequency down-conversion method according to the present invention. For example, FM intermediate frequency signal 2110 in FIG. 21D and FM intermediate frequency signal 2112 in FIG. 21E illustrate that FM carrier signal 816 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.2.2

Description of the structure

The operation of the undersampling system 1602 for the analog FM carrier signal 716 is described below with reference to the flow chart 1407 and the timing diagrams of Figures 20A-E. In step 1408, the undersampling system 1602 receives

dersampling module 1606 receives the FM carrier signal 716 (Fig. 20A). In step 1410, the undersampling module 1606 receives the undersampling signal 2006 (Fig. 20C). In step 1412, the FM carrier signal 716 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 2006 to downconvert the FM carrier signal 716 to the FM intermediate frequency signal 2012 (Fig. 20E).

The operation of the undersampling system 1602 for the digital FM carrier signal 816 is described below with reference to the flowchart 1407 and timing diagrams of Figures 21A-E. In step 1408, the undersampling module 1606 receives the FM carrier signal 816 (Figure 21A). In step 1410, the undersampling module 1606 receives the undersampling signal 2106 (Figure 21C). In step 1412, the FM carrier signal 816 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 2106 to down-convert the FM carrier signal 816 to the FM intermediate frequency signal 2112 (Figure 21E).

Implementation examples of the undersampling module 1606 are presented in sections 4 and 5 below.

1.2.3 Third exemplary embodiment: phase modulation 1.2.3.1 Description of the function

The exemplary processing of the flow chart 1407 in Fig. 14B for the analog PM carrier signal 916 shown in Fig. 9C and for the digital PM carrier signal 1016 shown in Fig. 10C is described below.

1.2.3.1.1 Analog PM carrier signal

A method for down-converting the analog PM carrier signal 916 to an analog PM intermediate frequency signal is described below with reference to the timing diagram 1407 in FIG. 14B. The analog PM carrier signal 916 is shown again in FIG. 23A for simplicity. In this example, the analog PM carrier signal 916 oscillates at a frequency of about 901 MHz. In FIG. 23B, a PM carrier signal 2304 represents a portion of the analog PM carrier signal 916 from a time ti to a time t3 on an expanded time scale.

Processing for down-converting the PM carrier signal 916 to a PM intermediate frequency signal begins in step 1408 by receiving an EM signal. This is represented in Figure 23A by the analog PM carrier signal 916.

In step 1410, an undersampling signal is received having an aliasing rate F _m . Figure 23C shows an undersampling signal 2306 at approximately the same time scale as Figure 23B. The undersampling signal 2306 comprises a train of pulses 2307 with slight pulse widths tending toward zero in duration. The pulses 2307 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of the difference frequency F _D i _FF when downconverted to an intermediate frequency signal. In this example, the aliasing rate is approximately 450 MHz.

In step 1412, the EM signal is subsampled at the aliasing rate to convert the EM signal into the intermediate frequency

•
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cross-signal F _JF . Step 1412 is illustrated in Fig. 23B by undersampling points 2305.

Because a harmonic of the aliasing rate is offset from a PM carrier signal 916, the undersampling points 2305 appear at different positions in successive cycles of the PM carrier signal 916. Therefore, different amplitude values of the PM carrier signal 916 are detected by the undersampling points.

In Figure 23D, voltage points 2308 are correlated with undersampling points 2305. In one embodiment, voltage points 2308 form an analog PM intermediate frequency signal 2310. This can be accomplished in many ways. For example, each voltage point 2308 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 23E, an analog PM IF signal 2312 represents the analog PM IF signal 2310 after a filtering operation on a compressed time scale. Although Fig. 23E shows the PM IF signal 2312 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The analog PM intermediate frequency signal 2312 is substantially similar to the analog PM carrier signal 916, except that the analog PM intermediate frequency signal 2312 has an intermediate frequency of 1 MHz. The analog PM intermediate frequency signal 2312 may be demodulated by any conventional PM demodulation technique.

The drawings referred to herein show a frequency down-conversion device according to the invention.

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The analog PM intermediate frequency signal 2310 in Fig. 23D and the analog PM intermediate frequency signal 2312 in Fig. 23E show that the analog PM carrier signal 2316 has been successfully downconverted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.3.1.2 Digital PM carrier signal

A method for down-converting the digital PM carrier signal 1016 to a digital PM intermediate frequency signal is described below with reference to the timing diagram 1407 in FIG. 14B. The digital PM carrier signal 1016 is shown again in FIG. 22A for simplicity. In this example, the digital PM carrier signal 1016 oscillates at a frequency of about 901 MHz. In FIG. 22B, a PM carrier signal 2204 represents a portion of the PM carrier signal 1016 from a time ti to a time t3 on an expanded time scale.

Processing begins in step 1408 where an EM signal is received. This is represented in Fig. 22A by the digital PM carrier signal 1016.

In step 1408, an undersampling signal is received having an aliasing rate F _AR . Fig. 22C shows an exemplary undersampling signal 2206 at approximately the same time scale as in Fig. 22B. The undersampling signal 2206 comprises a train of pulses 2207 with slight pulse widths whose duration tends toward zero. The pulses 2207 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F _AR is a harmonic or, more typically, a subharmony when downconverting to an intermediate frequency signal.

· 84 · ♦

monic of the difference frequency F _DIFF . In this example, the aliasing rate is about 450 MHz.

In step 1412, the EM signal is subsampled at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _IF . Step 1412 is illustrated in Fig. 22B by undersampling points 2205.

Because a harmonic of the aliasing rate is offset from the PM carrier signal 1016, the undersampling points 2205 appear at different positions in successive cycles of the PM carrier signal 1016.

In Fig. 22D, voltage points 2208 are correlated with undersampling points 2205. In one embodiment, voltage points 2208 form a digital PM intermediate frequency signal 2210. This can be accomplished in many ways. For example, each voltage point 2208 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 22E, a digital PM intermediate frequency signal 2212 represents the digital PM intermediate frequency signal 2210 after a filtering operation on a compressed time scale. Although Fig. 22E shows the PM intermediate frequency signal 2212 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The PM intermediate frequency signal 2212 is substantially similar to the PM carrier signal 1016, except that the PM intermediate frequency signal 2212 has an intermediate frequency of 1 MHz. The digital PM carrier signal 2212 may be

any conventional PM demodulation technique.

The drawings referred to herein illustrate a frequency down-conversion method according to the present invention. For example, digital PM intermediate frequency signal 2210 in FIG. 22D and digital PM intermediate frequency signal 2212 in FIG. 22E illustrate that digital PM carrier signal 1016 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.3.2 Description of the structure

The operation of the undersampling system 1602 for the analog PM carrier signal 916 is described below with reference to the flowchart 1407 and timing diagrams of Figures 23A-E. In step 1408, the undersampling module 1606 receives the PM carrier signal 916 (Figure 23A). In step 1410, the undersampling module 1606 receives the undersampling signal 2306 (Figure 23C). In step 1412, the PM carrier signal 916 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 2306 to down-convert the PM carrier signal 916 to the PM intermediate frequency signal 2312 (Figure 23E).

The operation of the undersampling system 1602 for the digital PM carrier signal 1016 is described below with reference to the flowchart 1407 and the timing diagrams of Figures 22A-E. In step 1408, the undersampling module 1606 receives the PM carrier signal 1016 (Figure 22A). In step 1410, the undersampling module 1606 receives the undersampling signal 2206 (Figure 22C). In step 1412, the PM carrier signal 1016 is

subsampled at the aliasing rate of the undersampling signal 2206 to downconvert the PM carrier signal 1016 into the PM intermediate frequency signal 2212 (Fig. 22E).

Implementation examples of the undersampling module 1606 are presented in sections 4 and 5 below.

1.2.4 Other embodiments

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. Those skilled in the art will appreciate from this description that other embodiments that differ slightly or substantially from those described herein may be implemented within the scope of the present invention. Examples of implementations of the undersampling module 1606 are presented in Sections 4 and 5 below.

1.3 Implementation examples

Sections 4 and 5 below describe exemplary operational and/or structural implementations related to the methods and/or embodiments described above. The implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the example implementations described herein. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention (e.g., equivalent implementations, extensions, changes and modifications of the implementations, etc.).

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2. Direct downconversion of an EM signal into a baseband signal (direct-to-data)

In one embodiment of the invention, an EM baseband signal is downconverted directly to baseband by undersampling the EM signal. This embodiment is referred to herein as direct-to-data downconversion and is illustrated in Figure 45B by reference numeral 4510.

This embodiment can be implemented with modulated and unmodulated EM signals. The embodiment is described herein by way of example using the modulated carrier signal F _MC in Fig. 1. In the example, the modulated carrier signal F _M c is directly downconverted to a demodulated baseband signal F _DMB . It will be apparent to those skilled in the art from the present description that the invention can be implemented such that any EM signal, eg a modulated carrier signal or an unmodulated carrier signal, can be downconverted.

In the following sections, exemplary methods for directly down-converting the modulated carrier signal F _MC into the demodulated baseband signal F _DMB are described. Exemplary structural embodiments for implementing the methods are also described. However, the invention is not limited to the specific embodiments described below. It will be apparent to those skilled in the art based on the present description that equivalent implementations, extensions, changes, modifications, etc. are possible within the scope of the present invention.

The following sections include a high-level description, exemplary embodiments, and implementation examples.

2.1 High-level description

Through this section (including its subsections), a high-level description of the inventive direct downconversion of the modulated carrier signal F _MC into the demodulated baseband signal F _DMB is provided. In particular, an operating method for directly downconverting the modulated carrier signal F _MC into the demodulated baseband signal F _DMB is presented in a high-level description. In addition, a structural implementation of this method is presented in a high-level description. This structural implementation is described herein for illustration and is not intended to be limiting. In particular, the method described in this section may be realized using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to those skilled in the art based on the present description.

2.1.1 Description of the functionality

Fig. 14C shows a flowchart 1413 illustrating an exemplary method for directly downconverting an EM signal into a demodulated baseband signal F _DMB . The exemplary method illustrated in the flowchart 1413

driving is an embodiment of the flowchart 1401 of Figure 14A.

Any and all combinations of modulation techniques are applicable to the present invention. For ease of description, the digital AM carrier signal 616 is used for a high-level description of the invention. In the following sections, exemplary AM and PM embodiments are described in detail. Special considerations are made with respect to frequency modulation (FM), which are discussed separately below in Section II.3. Those skilled in the art will appreciate from the present description that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and an unmodulated carrier signal.

The method illustrated in flow diagram 1413 is illustrated below in a high-level description using the digital AM carrier signal 616 of Figure 6C. The digital AM carrier signal 616 is shown again in Figure 33A for clarity.

Processing of flowchart 1413 begins in step 1414 where an EM signal is received. Step 1414 is represented by digital AM carrier signal 616.

In step 1416, an undersampling signal is received having an aliasing rate F _AR . Figure 33B shows an exemplary undersampling signal 3302 having a train of pulses 3303 with slight pulse widths tending toward zero in duration. The pulses 3303 repeat at the aliasing or pulse repetition rate. The aliasing rate is determined according to Eq. (2), which is re-illustrated below for simplicity.

Eq. (2)

If an EM signal is downconverted directly to a baseband signal (ie F _IF = O), equation 2 becomes:

Fc = n-FAR Eq. (8)

To down-convert the AM signal 616 directly to a demodulated baseband signal, the aliasing rate is substantially equal to the frequency of the AM signal 616 or a harmonic or subharmonic thereof. Although the aliasing rate is too low to allow reconstruction of higher frequency components of the AM signal 616 (i.e., the carrier frequency), it is high enough to substantially reconstruct the lower frequency modulating baseband signal 310.

In step 1418, the EM signal is subsampled at the aliasing rate to directly downconvert it to the demodulated baseband signal F _DMB . FIG. 33C shows a stepped demodulated baseband signal 3304 produced by the direct downconversion process. The demodulated baseband signal 3304 is similar to the digital modulating baseband signal 310 in FIG. 3.

Fig. 33D shows a filtered demodulated baseband signal 3306 that can be generated from the stepped baseband signal 3304. Thus, the invention can generate a filtered output signal, a partially filtered output signal, or a relatively unfiltered stepped output signal. The choice between a filtered, partially filtered, and unfiltered output signal is generally determined by the design, which depends on the application of the invention.

2.1.2 Description of the structure

16 shows a block diagram of an undersampling system 1602 according to an embodiment of the invention. The undersampling system 1602 is an exemplary embodiment of the general aliasing system 1302 of FIG. 13.

In a direct-to-data embodiment, the frequency of the undersampled signal 1604 is substantially equal to a harmonic, or more typically a subharmonic, of the EM signal 1304. Preferably, the EM signal 1304 is undersampled by the undersampling module 1606 to directly downconvert it to the demodulated baseband signal F _DMB in the manner illustrated in flowchart 1413. Other structural embodiments for carrying out the steps of flowchart 1413 are possible within the scope of the invention. The details of the other structural embodiments will be apparent to those skilled in the art based on the present description.

The operation of the aliasing system 1602 for the digital AM carrier signal 616 is described below with reference to the flowchart 1413 and timing diagrams in Figures 33A-D. In step 1414, the undersampling module 1606 receives the AM carrier signal 616 (Figure 33A). In step 1416, the undersampling module 1606 receives the undersampling signal 3302 (Figure 33B). In step 1418, the AM carrier signal 616 is undersampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 3302 to downconvert the AM carrier signal 616 directly to the demodulated baseband signal 3304 shown in FIG. 33C or to the filtered demodulated baseband signal 3306 shown in FIG. 33D.

· a ·

Implementation examples of the undersampling module 1606 are presented in sections 4 and 5 below.

2.2 Exemplary embodiments

In this section (and its subsections), various embodiments related to the method(s) and structure(s) described above are presented. These embodiments are described herein for illustrative purposes and are not to be taken in a limiting sense. The invention is not limited to these embodiments. It will be apparent to those skilled in the art based on the present description that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and adapted to include these alternative embodiments.

The method illustrated in flowchart 1413 of FIG. 14C for downconverting the EM signal 1304 to the demodulated baseband signal F _DMB may be implemented with any type of EM signal, e.g., modulated carrier signals such as AM, PM carrier signals, etc., or any combination thereof. The processing of flowchart 1413 of FIG. 14C is described below for AM and PM carrier signals. The following exemplary descriptions are provided to illustrate the present invention. The present invention is not limited by or to the following exemplary embodiments.

2.2.1 First exemplary embodiment: Amplitude modulation
2.2.1.1 Description of the functionality

The exemplary processing of the flow chart 1413 in Fig. 14C for the analog AM carrier signal 516 shown in Fig. 5C and for the digital AM carrier signal 616 shown in Fig. 6C is described below.

2.2.1.1.1 Analog AM carrier signal

A method for directly downconverting the analog AM carrier signal 516 to a demodulated baseband signal is described below with reference to the flow chart 1413 in FIG. 14C. The analog AM carrier signal 516 is shown again in FIG. 35A for simplicity. In this example, the analog AM carrier signal 516 oscillates at a frequency of about 900 MHz. In FIG. 35B, an analog AM carrier signal 3504 represents a portion of the analog AM carrier signal 516 on an expanded time scale.

Processing begins in step 1414 where an EM signal is received. This is represented by the analog AM carrier signal 516 in Figure 19A.

In step 1416, an undersampling signal is received having an aliasing rate F _AR . Fig. 35C shows an exemplary undersampling signal 3506 at approximately the same time scale as in Fig. 35B. The undersampling signal 3506 comprises a train of pulses 3507 with slight pulse widths whose duration tends toward zero. The pulses 3507 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F _AR is low when directly downconverting to a demo-

modulated baseband signal is essentially equal to a harmonic or, more typically, a subharmonic of the subsampled signal. In this example, the aliasing rate is about 450 MHz.

In step 1418, the EM signal is subsampled at the aliasing rate ? to downconvert it directly to the demodulated baseband signal F _DMB . Step 1418 is illustrated in Figure 35B by subsampling points 3505. Because a harmonic of the aliasing rate is substantially equal to the frequency of the signal 516, substantially no intermediate frequency signal is generated. The only aliased component is the baseband signal.

In Figure 35D, voltage points 3508 are correlated with undersampling points 3505. In one embodiment, voltage points 3508 form a demodulated baseband signal 3510. This can be accomplished in many ways. For example, each voltage point 3508 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 35E, a demodulated baseband signal 3512 represents the demodulated baseband signal 3510 after a filtering operation on a compressed time scale. Although Fig. 35E shows the demodulated baseband signal 3512 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be adapted for various applications.

The demodulated baseband signal 3512 is substantially similar to the modulating baseband signal 210. The demodulated baseband signal 3512 can be modulated by any

signal processing technique(s) without further downconversion or demodulation.

The aliasing rate of the undersampling signal is preferably controlled to optimize the demodulated baseband signal in terms of amplitude output values or polarity.

In the above example, the undersampling points 3505 appear at positive positions of the AM carrier signal 516. Alternatively, the undersampling points 3505 may appear at other positions, e.g., at negative points, of the analog AM carrier signal 516. When the undersampling points 3505 appear at negative points of the AM carrier signal 516, the resulting demodulated baseband signal is inverted with respect to the modulating baseband signal 210.

The drawings referred to herein illustrate a direct-to-data downconversion method according to the present invention. For example, demodulated baseband signal 3510 in FIG. 35D and demodulated baseband signal 3512 in FIG. 35E illustrate that AM carrier signal 516 has been successfully downconverted to demodulated baseband signal 3510 while maintaining sufficient baseband information for adequate reconstruction.

2.2.1.1.2 Digital AM carrier signal

A method for directly downconverting the digital AM carrier signal 616 to a demodulated baseband signal is described below with reference to the flow chart 1413 in Fig. 14C. The digital AM carrier signal 616 is shown again in Fig. 36A for simplicity. In this example, the digital AM carrier signal 616 oscillates at a frequency of about 901 MHz. In Fig. 36B, a

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digital AM carrier signal 3604 represents a portion of the digital AM carrier signal 616 in an expanded time scale.

Processing begins in step 1414 where an EM signal is received. This is represented by the digital AM carrier signal 616.

In step 1416, an undersampling signal is received having an aliasing rate F _AR . Figure 36C shows an exemplary undersampling signal 3606 at approximately the same time scale as Figure 36B. The undersampling signal 3606 comprises a train of pulses 3607 with slight pulse widths tending toward zero in duration. The pulses 3607 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F _AR when downconverted to a demodulated baseband signal is equal to a harmonic, or more typically a subharmonic, of the undersampled signal. In this example, the aliasing rate is about 450 MHz.

In step 1418, the EM signal is subsampled at the aliasing rate to downconvert it directly to the demodulated baseband signal F _DMB . Step 1418 is represented in Figure 36B by undersampling points 3605. Because the aliasing rate is substantially equal to the AM carrier signal 616 or a harmonic or subharmonic of its frequency, substantially no intermediate frequency signal is generated. The only significant aliased component is the baseband signal.

In Fig. 36D, voltage points 3608 are correlated with undersampling points 3605. In one embodiment, voltage points 3608 form a demodulated baseband signal 3610. This can be realized in many ways. For example, each voltage point 3608 can be on a re-

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relatively constant level until the next voltage point is received. This produces a stepped output signal which can be smoothed or filtered if necessary, as described below.

In Fig. 36E, a demodulated baseband signal 3612 represents the demodulated baseband signal 3610 after a filtering operation on a compressed time scale. Although Fig. 36E shows the demodulated baseband signal 3612 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The demodulated baseband signal 3612 is substantially similar to the digital modulating baseband signal 310. The demodulated analog baseband signal 3612 may be processed by any conventional signal processing technique(s) without further downconversion or demodulation.

The aliasing rate of the undersampling signal is preferably controlled to optimize the demodulated baseband signal in terms of amplitude output values or polarity.

In the above example, the undersampling points 3605 appear at positive positions of the signal portion 3604. Alternatively, the undersampling points 3605 may appear at other positions, e.g., at negative positions, of the signal portion 3604. If the undersampling points 3605 appear at negative positions, the resulting demodulated baseband signal is inverted with respect to the modulating baseband signal 310.

The drawings referred to herein show a frequency down-conversion device according to the invention.

For example, demodulated baseband signal 3610 in FIG. 36D and demodulated baseband signal 3612 in FIG. 36E show that digital AM carrier signal 616 was successfully downconverted to demodulated baseband signal 3610 while maintaining sufficient baseband information for adequate reconstruction.

2.2.1.2 Description of the structure

The operation of the undersampling module 1606 for the analog AM carrier signal 516 is described below with reference to the flowchart 1413 and the timing diagrams of Figures 35A-E. In step 1414, the undersampling module 1606 receives the analog AM carrier signal 516 (Figure 35A). In step 1416, the undersampling module 1606 receives the undersampling signal 3506 (Figure 35C). In step 1418, the analog AM carrier signal 516 is undersampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 3506 to downconvert the AM carrier signal 516 directly to the demodulated analog baseband signal 3510 shown in FIG. 35D or the filtered demodulated analog baseband signal 3512 shown in FIG. 35E.

The operation of the undersampling system 1602 for the digital AM carrier signal 616 is described below with reference to the flow chart 1413 and the timing diagrams of Figures 36A-E. In step 1414, the undersampling module 1606 receives the digital AM carrier signal 616 (Figure 36A). In step 1416, the undersampling module 1606 receives the undersampling signal 3606 (Figure 36C). In step 1418, the digital AM carrier signal 616 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 3606 to obtain the digital AM

·

·

·

carrier signal 616 into the demodulated digital baseband signal 3610 shown in Fig. 36D or into the filtered demodulated digital baseband signal 3612 shown in Fig. 36E.

Implementation examples of the undersampling module 1606 are presented in sections 4 and 5 below.

2.2.2 Second exemplary embodiment: phase modulation 2.2.2.1 Description of the function

The following illustrates the operation of the exemplary processing of the flow chart 1413 in Fig. 14C for the analog PM carrier signal 916 shown in Fig. 9C and for the digital PM carrier signal 1016 shown in Fig. 10C.

2.2.2.1.1 Analog PM carrier signal

A method for directly downconverting the analog PM carrier signal 916 to a demodulated baseband signal is described below with reference to the timing diagram 1413 in FIG. 14C. The analog PM carrier signal 916 is shown again in FIG. 37A for simplicity. In this example, the analog PM carrier signal 916 oscillates at a frequency of about 900 MHz. In FIG. 37B, an analog PM carrier signal 3704 represents a portion of the analog PM carrier signal 916 on an expanded time scale.

Processing begins in step 1414 where an EM signal is received. This is represented by the analog PM carrier signal 916.

In step 1416, an undersampling signal is received having an aliasing rate F _AR . Fig. 37C shows an exemplary undersampling signal 3706 in approximately the

same time scale as in Fig. 37B. The undersampling signal 3706 comprises a train of pulses 3707 with slight pulse widths tending to zero in duration. The pulses 3707 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, the aliasing rate F _AR for a direct downconversion to a demodulated baseband signal is equal to a harmonic, or more typically a subharmonic, of the frequency of the undersampled signal. In this example, the aliasing rate is about 4 50 MHz.

In step 1418, the analog PM carrier signal 916 is subsampled at the aliasing rate to downconvert it directly to a demodulated baseband signal. Step 1418 is illustrated in Figure 37B by undersampling points 3705.

Because a harmonic of the aliasing rate is substantially equal to the frequency of signal 916 or a harmonic or subharmonic thereof, no intermediate frequency signal is produced. The only significant aliased component is the baseband signal.

In Figure 37D, voltage points 3708 are correlated with undersampling points 3705. In one embodiment, voltage points 3708 form a demodulated baseband signal 3710. This can be accomplished in many ways. For example, each voltage point 3708 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 37E, a demodulated baseband signal 3712 represents the demodulated baseband signal 3710 after a filtering operation in a compressed time scale. Although

Fig. 37ε shows the demodulated baseband signal 3712 as a filtered output signal, according to the invention the output signal does not have to be filtered or smoothed. Instead the output signal can be adapted for different applications.

The demodulated baseband signal 3712 is substantially similar to the analog modulating baseband signal 210. The demodulated baseband signal 3712 can be processed without further downconversion or demodulation.

The aliasing rate of the undersampling signal is preferably controlled in order to optimize the demodulated baseband signal in terms of amplitude output values or polarity, if desired.

In the above example, the undersampling points 3705 appear at positive positions of the analog PM carrier signal 916. Alternatively, the undersampling points 3705 may appear at other positions, e.g., at negative positions, of the analog PM carrier signal 916. When the undersampling points 3705 appear at negative positions of the analog PM carrier signal 916, the resulting demodulated baseband signal is inverted with respect to the modulating baseband signal 210.

The drawings referred to herein illustrate a direct-to-data downconversion method according to the present invention. For example, demodulated baseband signal 3710 in FIG. 37D and demodulated baseband signal 3712 in FIG. 37E illustrate that analog PM carrier signal 916 has been successfully downconverted to demodulated baseband signal 3710 while maintaining sufficient baseband information for adequate reconstruction.

♦ ««

2.2.2.1.2 Digital PM carrier signal

A method for directly downconverting the digital PM carrier signal 1016 to a demodulated baseband signal is described below with reference to the timing diagram 1413 in FIG. 14C. The digital PM carrier signal 1016 is shown again in FIG. 38A for simplicity. In this example, the digital PM carrier signal 1016 oscillates at a frequency of about 900 MHz. In FIG. 38B, a digital PM carrier signal 3804 represents a portion of the digital PM carrier signal 1016 on an expanded time scale.

Processing begins in step 1414 where an EM signal is received. This is represented by the digital PM carrier signal 1016.

In step 1416, an undersampling signal is received having an aliasing rate Far. Figure 38C shows an exemplary undersampling signal 3806 at approximately the same time scale as Figure 38B. The undersampling signal 3806 comprises a train of pulses 3807 with slight pulse widths tending toward zero in duration. The pulses 3807 repeat at the aliasing rate or pulse repetition rate determined or selected as described above. In general, when directly downconverted to a demodulated baseband signal, the aliasing rate Far is equal to a harmonic, or more typically a subharmonic, of the frequency of the undersampled signal. In this example, the aliasing rate is about 450 MHz.

In step 1418, the digital PM carrier signal 1016 is subsampled at the aliasing rate to downconvert it directly to a demodulated baseband signal. This is illustrated in Figure 38B by undersampling points 3705.

Because a harmonic of the aliasing rate is substantially equal to the frequency of signal 1016, essentially no intermediate frequency signal is generated.

In Figure 38D, voltage points 3808 are correlated with undersampling points 3805. In one embodiment, voltage points 3808 form a demodulated baseband signal 3810. This can be accomplished in many ways. For example, each voltage point 3808 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

In Fig. 38E, a demodulated baseband signal 3812 represents the demodulated baseband signal 3810 after a filtering operation on a compressed time scale. Although Fig. 38E shows the demodulated baseband signal 3812 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for various applications.

The demodulated baseband signal 3812 is substantially similar to the digital modulating baseband signal 310. The demodulated baseband signal 3812 can be processed without further downconversion or demodulation.

The aliasing rate of the undersampling signal is preferably controlled to optimize the demodulated baseband signal in terms of amplitude output values or polarity.

In the above example, the undersampling points 3805 appear at positive positions of the digital PM carrier signal 1016. Alternatively, the undersampling points 3805 may appear at other positions, e.g. at negative positions, of the digital PM carrier signal 1016. If

the undersampling points 3805 appear at negative positions of the digital PM carrier signal 1016, the obtained demodulated baseband signal is inverted with respect to the modulating baseband signal 310.

The drawings referred to herein illustrate a frequency down-conversion method according to the present invention. The demodulated baseband signal 3810 in FIG. 38D and the demodulated baseband signal 3812 in FIG. 38E show that the digital PM carrier signal 1016 has been successfully down-converted to the demodulated baseband signal 3810 while maintaining sufficient baseband information for adequate reconstruction.

2.2.2.2 Description of the structure

The operation of the undersampling system 1602 for the analog PM carrier signal 916 is described below with reference to the flowchart 1413 and timing diagrams of Figures 37A-E. In step 1414, the undersampling module 1606 receives the analog PM carrier signal 916 (Figure 37A). In step 1416, the undersampling module 1606 receives the undersampling signal 3706 (Figure 37C). In step 1418, the analog PM carrier signal 916 is undersampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 3706 to downconvert the PM carrier signal 916 to the demodulated analog baseband signal 3710 shown in FIG. 37D or the filtered demodulated analog baseband signal 3712 shown in FIG. 37E.

The operation of the undersampling system 1602 for the digital PM carrier signal 1016 is described below with reference to the flowchart 1413 and the timing diagrams of Figures 38A-E. In step 1414, the undersampling module 1606 receives the digital PM carrier signal 1016

.· .· . :&idiagr;&dgr;5!. i

(Fig. 38A). In step 1416, the undersampling module 1606 receives the undersampling signal 3806 (Fig. 38C). In step 1418, the digital PM carrier signal 1016 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 3806 to downconvert the digital PM carrier signal 1016 to the demodulated digital baseband signal 3810 shown in Fig. 38D or the filtered demodulated digital baseband signal 3812 shown in Fig. 38E.

2.2.3 Other embodiments

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. It will be apparent to those skilled in the art from this description that other embodiments that differ slightly or substantially from those described herein can be implemented within the scope of the present invention.

2.3 Implementation examples

Sections 4 and 5 below describe exemplary functional and/or structural implementations related to the methods and/or embodiments described above. The implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the specific implementation examples described herein. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention.

(e.g. equivalent implementations, extensions, changes and modifications of the implementations, etc.).

3. Modulation implementation

In one embodiment of the invention, an FM carrier signal Ffmc is down-converted to a non-FM signal F _(NO n-fm) by undersampling the FM carrier signal Ffmc. This embodiment is illustrated in Fig. 45B by reference numeral 4512.

In an exemplary embodiment, the FM

Carrier signal F _FMC is down-converted into a phase modulated (PM) signal F _PM . In another exemplary embodiment, the FM carrier signal Ffmc is down-converted ^into an amplitude modulated (AM) signal F _m . The invention is not limited to these embodiments. The down-converted signal may be demodulated by any conventional demodulation technique to obtain a demodulated baseband signal F _DMB .

The invention can be implemented with any FM signal type. Exemplary embodiments for downconverting a frequency shift keying (FSK)
signals into a non-FSK signal. FSK signals are a subset of FM signals where an FM signal switches between two or more frequencies. Frequency shift keying (FSK) is typically used for digital modulating baseband signals, e.g., for the digital modulating baseband signal 310 in Fig. 3. For example, in Fig. 8, the digital FM signal 816 is an FSK signal that switches between an upper frequency and a lower frequency, which causes amplitude changes in the digital modulating

corresponds to the baseband signal 310. The FSK signal 816 is used in exemplary embodiments below.

In a first example, the FSK signal 816 is subsampled at an aliasing rate based on a center frequency between the upper and lower frequencies of the FSK signal 816. When the aliasing rate is based on the center frequency, the FSK signal 816 is down-converted to a phase shift keying (PSK) signal. PSK signals are a subset of phase modulated signals where a PM signal switches between two or more phases. Phase shift keying (PSK) is typically used for digital modulating baseband signals. For example, in Figure 10, the digital PM signal 1016 is a PSK signal that switches between two phases. The PSK signal 1016 may be demodulated by any conventional PSK demodulation technique.

In a second exemplary embodiment, the FSK signal 816 is subsampled at an aliasing rate based on either the upper frequency or the lower frequency of the FSK signal 816. If the aliasing rate is based on the upper frequency or the lower frequency of the FSK signal 816, the FSK signal 816 is down-converted to an amplitude shift keying (ASK) signal. ASK signals are a subset of amplitude modulated signals where an AM signal switches between two or more amplitudes. Amplitude shift keying (ASK) is typically used for digital modulating baseband signals. For example, in FIG. 6, the digital AM signal 616 is an ASK signal that switches between a first amplitude and a second amplitude. The ASK signal 616 can be demodulated by any conventional ASK demodulation technique.

The following sections describe methods for undersampling an FM carrier signal F _FM c in order to convert it into

1Γ;

down-convert a non-FM signal F _(NO n-fm). In addition, exemplary structural embodiments for implementing the methods are described. The invention is not limited to the specific embodiments described below. It will be apparent to those skilled in the art based on the present description that equivalent implementations, extensions, changes, modifications, etc. are possible within the scope of the present invention.

The following sections contain a high-level description, example embodiments, and implementation examples.

3.1 High-level description

Through this section (including its subsections) a high-level description is provided of undersampling the FM carrier signal F _FM for down-converting the FM carrier signal to a non-FM signal F(NON-FM) according to the invention. In particular, an operating method for down-converting the FM carrier signal F _MC to the non-FM signal F _(NO n-fm) is presented ⁱⁿ a high-level description. In addition, a structural implementation of this method is presented in a high-level description. This structural implementation is described herein for illustration purposes and is not to be taken in a limiting sense. In particular, the method described in this section may be realized using any number of structural implementations, one of which is described in this section. The details of such structural implementations are for

·

Those skilled in the art will appreciate the benefits of this description.

3.1.1 Description of the functionality

14D shows a flow diagram 1419 illustrating an exemplary method for down-converting an FM carrier signal F _FMC to a non-FM signal F( _NO n-fm). The exemplary method illustrated in flow diagram 1419 is an embodiment of the flow diagram 1401 of FIG. 14A.

Any and all forms of frequency modulation techniques are applicable to the present invention. For ease of description, the digital FM carrier (FSK)
signal 816 is used to provide a high-level description of how the invention works. Detailed timing diagrams and descriptions for the FSK signal 816 are provided throughout the following sections. Those skilled in the art will appreciate from this description that the invention can be implemented to down-convert any type of FM carrier signal.

The process illustrated in flow diagram 1419 is illustrated below with a high-level description for down-converting the FSK signal 816 of Figure 8C to a PSK signal. The FSK signal 816 is shown again in Figure 39A for simplicity.

Processing of flowchart 1419 begins in step 1420 where an FM signal is received. FSK signal 816 switches between an upper frequency 3910 and a lower frequency 3912. In an exemplary embodiment, upper frequency 3910 is about 901 MHz and lower frequency 3912 is about 899 MHz.

In step 1422, an undersampling signal is received having an aliasing rate F _AR . Fig. 39B shows a

Example of an undersampled signal 3902 comprising a train of pulses 3903 with slight pulse widths whose duration tends to zero. The pulses 3903 repeat at the aliasing or pulse repetition rate.

When an FM carrier signal Ffmc is down-converted to a non-FM signal F(NON-FM), the aliasing rate is substantially equal to a frequency contained in the FM signal or a harmonic or subharmonic thereof. In this overview of an exemplary embodiment in which the FSK signal 816 is down-converted to a PSK signal, the aliasing rate is based on a center frequency between the upper frequency 3110 and the lower frequency 3912. For this example, the center frequency is approximately 900 MHz. In another embodiment described below in which the FSK signal 816 is to be down-converted to an ASK signal, the aliasing rate is not based on the center position but on either the upper frequency 3910 or the lower frequency 3912.

In step 1424, the FM signal F _FMC is subsampled at the aliasing rate to down-convert the FM carrier signal F^c to the non-FM signal F _(N0N _ _FM ). Step 1424 is illustrated in Fig. 39C, which shows a stepped PSK signal 3904 generated by a modulation conversion process.

When the upper frequency 3910 is subsampled, the PSK signal 3904 has a frequency of approximately 1 MHz and is used as a phase reference. When the lower frequency 3912 is subsampled, the PSK signal 3904 has a frequency of 1 MHz and is 180 degrees out of phase with respect to the phase reference.

Fig. 39D shows a PSK signal 3906 as a filtered version of the PSK signal 3904. According to the invention, therefore, a

··

filtered or partially filtered output signal or a relatively unfiltered stepped output signal can be generated. The choice between a filtered, partially filtered and unfiltered output signal is generally determined by the design, which depends on the application of the invention.

The aliasing rate of the undersampling signal is preferably controlled to optimize the downconverted signal in terms of amplitude output values or polarity.

Detailed exemplary embodiments for down-converting an FSK signal to a PSK signal and for down-converting an FSK signal to an ASK signal are described below.

3.1.2 Description of the structure

16 shows a block diagram of the undersampling system 1602 according to an embodiment of the invention. The undersampling system 1602 includes an undersampling module 1606. The undersampling system 1602 is an exemplary embodiment of the general aliasing system 1302 in FIG. 13.

In the molecular conversion embodiment, the EM signal 1304 is an FM carrier signal, and the undersampling module 1606 subsamples the FM carrier signal at a frequency substantially equal to a harmonic, or more typically a subharmonic, of a frequency of the FM signal. Preferably, the FM carrier signal F _MC is subsampled by the undersampling module 1606 to downconvert it to a non-FM signal F _(NO n-fm) in a manner illustrated in flow diagram 1419. Other structural embodiments for

Performing the steps of flowchart 1419 is possible. The details of the other structural embodiments will be apparent to those skilled in the art based on the description presented herein.

The operation of the undersampling system 1602 is described below with reference to the flowchart 1419 and timing diagrams in Figures 39A-D. In step 1420, the undersampling module 1606 receives the FSK signal 816. In step 1422, the undersampling module 1606 receives the undersampling signal 3902. In step 1424, the FSK signal 816 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 3902 to down-convert the FSK signal 816 to the PSK signal 3904 or 3906.

Implementation examples of the undersampling module 1606 are presented in Section 4 below.

3.2 Exemplary embodiments

In this section (and its subsections), various embodiments related to the method(s) and structure(s) described above are presented. These embodiments are described herein for illustrative purposes and are not to be taken in a limiting sense. The invention is not limited to these embodiments. It will be apparent to those skilled in the art based on the present description that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and adapted to include these alternative embodiments.

The method illustrated in the flowchart 1419 of FIG. 14D for down-converting an FM carrier signal Ff ₇₁₅₀ to a non-FM signal F _(N0N _ _FM ) can be implemented with any type of FM carrier signal, such as FSK signals. The flowchart 1419 for down-converting an FSK signal to a PSK signal and for down-converting an FSK signal to an ASK signal is described in detail below. The following exemplary descriptions are provided to clarify the present invention. The present invention is not limited by or to the following exemplary embodiments.

3.2.1 First exemplary embodiment: Downconversion of an FM signal into a PM signal

3.2.1.1 Description of the functionality

The exemplary processing of the flow chart 1419 in Fig. 14D for down-converting the FSK signal 816 shown in Fig. 8C to a PSK signal is described below. The FSK signal 816 is shown again in Fig. 40A for simplicity.

The FSK signal 816 switches between a first frequency 4006 and a second frequency 4008. In the exemplary embodiment, the first frequency 4006 is lower than the second frequency 4008. In an alternate embodiment, the first frequency 4006 is higher than the second frequency 4008. In this example, the first frequency is about 899 MHz and the second frequency 4008 is about 901 MHz.

Fig. 40B shows an FSK signal portion 4004 representing a portion of the FSK signal 816 in an expanded time scale.

Processing to down-convert the FSK signal 816 to a PSK signal begins in step 1420 by receiving an FM signal, represented by the FSK signal 816.

In step 1422, an undersampling signal is received having an aliasing rate F^. Figure 40C shows an exemplary undersampling signal 4007 at approximately the same time scale as Figure 40B. The undersampling signal 4007 comprises a train of pulses 4009 with slight pulse widths tending toward zero in duration. The pulses 4009 repeat at the aliasing rate determined or selected as described above. In general, the aliasing rate of downconversion of an FM signal to a non-FM signal is substantially equal to a harmonic, or more typically a subharmonic, of a frequency contained in the FM signal.

In this example, where an FSK signal is downconverted to a PSK signal, the aliasing rate of a harmonic, or more typically a subharmonic, of the center frequency is substantially the same between frequencies 4006 and 4008. In this example, where the first frequency 4006 is 899 MHz and the second frequency 4008 is 901 MHz, the center frequency is about 900 MHz. Suitable aliasing rates are 1.8 GHz, 900 MHz, 450 MHz, etc. In this example, the aliasing rate of the undersampled signal 4008 is about 450 MHz.

In step 1424, the FM signal is subsampled at the aliasing rate to down-convert it to the non-FM signal F( _N0N _ _FM) . Step 1424 is illustrated in Fig. 40B by undersampling points 4005. The undersampling

Points 4005 appear with the aliasing rate of pulses 4009.

In Fig. 40D, voltage points 4010 are correlated with undersampling points 4005. In one embodiment, voltage points 4010 form a PSK signal 4012. This can be accomplished in many ways. For example, each voltage point 4010 can be maintained at a relatively constant level until the next voltage point is received. This provides a stepped output signal that may optionally be smoothed or filtered as described below.

When the first frequency 4006 is subsampled, the PSK signal 4012 has a frequency of approximately 1 MHz and is used as a phase reference. When the second frequency 4008 is subsampled, the PSK signal 4012 has a frequency of 1 MHz and is 180 degrees out of phase with respect to the phase reference.

In Fig. 40E, PSK signal 4014 represents PSK signal 4012 after a filtering operation on a compressed time scale. Although Fig. 40E shows PSK signal 4012 as filtered output signal 4014, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for different applications. PSK signal 4014 can be demodulated by any conventional phase demodulation technique.

The aliasing rate of the undersampling signal is preferably controlled to optimize the downconverted signal in terms of amplitude output values or polarity.

In the above example, the undersampling points 4005 appear at positive positions of the FSK signal 816. Alternatively, the undersampling points 4005 may appear at other positions.

·

·

positions, e.g., at negative positions, of the FSK signal 816. When the undersampling points 4005 appear at negative positions of the FSK signal 816, the obtained PSK signal is inverted with respect to the PSK signal 4014.

The drawings referred to herein illustrate a modulation conversion method according to the present invention. For example, PSK signal 4014 shown in Figure 40E illustrates that FSK signal 816 has been successfully downconverted to PSK signals 4012 and 4014 while maintaining sufficient baseband information for adequate reconstruction.

3.2.1.2 Description of the structure

The operation of the undersampling system 1602 for down-converting the FSK signal 816 to a PSK signal is described below with reference to the flowchart 1419 and timing diagrams of Figures 40A-E. In step 1420, the undersampling module 1606 receives the FSK signal 816 (Figure 40A). In step 1422, the undersampling module 1606 receives the undersampling signal 4007 (Figure 40C). In step 1424, the FSK signal 816 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 4007 to down-convert the FSK signal 816 into the PSK signal 4012 shown in Fig. 40D or the PSK signal 4014 shown in Fig. 40E.

3.2.2 Second exemplary embodiment: Downconversion of an FM signal into an AM signal

3.2.2.1 Description of the functionality

The exemplary processing of the flow chart of Fig. 14D for down-converting the FSK signal 816 shown in Fig. 8C to an ASK signal is described below. The FSK signal 816 is shown again in Fig. 4IA for simplicity.

The FSK signal 816 switches between a first frequency 4106 and a second frequency 4108. In the exemplary embodiment, the first frequency 4106 is lower than the second frequency 4108. In an alternate embodiment, the first frequency 4106 is higher than the second frequency 4108. In this example, the first frequency 4106 is about 899 MHz and the second frequency 4108 is about 901 MHz.

Figure 41B shows an FSK signal portion 4104 representing a portion of the FSK signal 816 in an expanded time scale.

Processing to down-convert the FSK signal 816 to an ASK signal begins in step 1420 by receiving an FM signal, represented by the FSK signal 816.

In step 1422, an undersampling signal is received having an aliasing rate F _AR . Fig. 41C shows an exemplary undersampling signal 4107 at approximately the same time scale as in Fig. 42B. The undersampling signal 4107 comprises a sequence of pulses 4109 with slight pulse widths whose duration tends toward zero. The pulses 4109 repeat at the aliasing rate or pulse

Repetition rate. The aliasing rate is determined or selected as described above.

In general, when downconverting an FM signal to a non-FM signal, the aliasing rate of a harmonic, or more typically a subharmonic, of a frequency of the FM signal is substantially equal. When downconverting an FSK signal 816 to an ASK signal, the aliasing rate of a harmonic of the first frequency 4106 or the second frequency 4108 and more typically a subharmonic of the first frequency 4106 or the second frequency 4103 is substantially equal. In this example, where the first frequency 4106 is 899 MHz and the second frequency 4108 is 901 MHz, the aliasing rate of a harmonic or subharmonic of 899 MHz or 901 MHz may be substantially equal. In this example, the aliasing rate is about 449.5 MHz, which corresponds to a subharmonic of the first frequency 4106.

In step 1424, the FM signal is subsampled at the aliasing rate to down-convert it to a non-FM signal F( _NO n-fm). Step 1424 is represented in Figure 41B by undersampling points 4105. The undersampling points appear at the aliasing rate of the pulses 4109. When the first frequency 4106 is subsampled, the aliasing pulses 4109 and the undersampling points 4105 appear at the same position in successive cycles of the FSK signal 816. This produces a relatively constant output signal level. However, when the second frequency 4108 is subsampled, the aliasing pulses 4109 and the undersampling pulses 4005 appear at different positions in successive cycles of the FSK signal 816. This produces a pattern oscillating at a frequency of approximately (901 MHz -899 MHz) = 2 MHz.

In Fig. 41D, voltage points 4110 are correlated with undersampling points 4105. In one embodiment, voltage points 4110 form an ASK signal 4112. This can be accomplished in many ways. For example, each voltage point 4110 can be maintained at a relatively constant level until the next voltage point is received. This results in a stepped output signal that can optionally be smoothed or filtered as described below.

In Fig. 41E, an ASK signal 4114 represents the PSK signal 4112 after a filtering operation on a compressed time scale. Although Fig. 41E shows the ASK signal 4114 as a filtered output signal, according to the invention, the output signal does not need to be filtered or smoothed. Instead, the output signal can be tailored for different applications. The ASK signal 4114 can be demodulated by any conventional amplitude demodulation technique.

The aliasing rate of the undersampling signal is preferably controlled during downconversion from an FM to an AM signal in order to optimize the demodulated baseband signal in terms of amplitude output values or polarity.

In an alternative embodiment, the aliasing rate is based on the second frequency and the obtained ASK signal is inverted with respect to the ASK signal 4114.

The drawings referred to herein illustrate a modulation conversion method according to the present invention. For example, the ASK signal 4114 shown in Figure 41E illustrates that the FSK signal 816 has been successfully downconverted to the ASK signal 4114 while maintaining sufficient baseband information for adequate reconstruction.

·

3.2.2.2 Description of the structure

The operation of the undersampling system 1602 for down-converting the FSK signal 816 to an ASK signal is described below with reference to the flowchart 1419 and timing diagrams of Figures 41A-E. In step 1420, the undersampling module 1606 receives the FSK signal 816 (Figure 41A). In step 1422, the undersampling module 1606 receives the undersampling signal 4107 (Figure 41C). In step 1424, the FSK signal 816 is subsampled by the undersampling module 1606 at the aliasing rate of the undersampling signal 4107 to down-convert the FSK signal 816 into the ASK signal 4112 shown in Fig. 41D or the ASK signal 4114 shown in Fig. 41E.

3.2.3 Other exemplary embodiments

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. It will be apparent to those skilled in the art from this description that other embodiments that differ slightly or substantially from those described herein can be implemented within the scope of the present invention.

3.3 Implementation examples

Sections 4 and 5 below describe exemplary functional and/or operational and/or structural implementations related to the methods and/or embodiments described above. The implementations are for illustrative purposes only.

·::«·:>. IiVOi

elucidation and are not to be taken in a limiting sense. The invention is not limited to the specific implementation examples described herein. It will be apparent to those skilled in the art based on the present description that alternative implementations are possible within the scope of the present invention (e.g., equivalent implementations, extensions, changes and modifications of the implementations, etc.).

4. Implementation examples

This section (and its subsections) describes exemplary functional and/or structural implementations related to the methods and/or embodiments described in the preceding subsections. These implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the implementation examples described herein. It will be apparent to those skilled in the art based on the present description that alternative implementations are possible within the scope of the present invention (e.g., equivalent implementations, extensions, changes and modifications of the implementations, etc.).

Fig. 13 shows a general aliasing system 1302 with an aliasing module 1306. Fig. 16 shows an undersampling system 1602 with an undersampling module 1606. The undersampling module 1606 receives an undersampling signal 1604 having an aliasing rate F _n R. The undersampling signal 1604 has a sequence of pulses with slight pulse widths whose duration tends to zero. The pulses 3903 repeat at the aliasing rate F _A r. The

Undersampling system 1602 is an exemplary implementation of general aliasing system 1303. Undersampling system 1602 outputs a downconverted signal 1308A.

26A shows an exemplary sample and hold system 2602 that is an exemplary implementation of the undersampling system 1602. The sample and hold system 2602 is described below.

Figure 26B shows an exemplary inverted sample and hold system 2606 that is an alternative exemplary implementation of the undersampling system 1602. The inverted sample and hold system 2606 is described below.

4.1 The undersampling system as a sample-and-hold system

26A shows a block diagram of the sample and hold system 2602, which is an exemplary embodiment of the undersampling module 1606 of FIG. 16, which is an exemplary embodiment of the general aliasing module 1306 of FIG. 13.

The sample and hold system 2602 includes a sample and hold module 2604 that receives the EM signal 1304 and the undersampling signal 1604. The EM signal is subsampled by the sample and hold module 2604 at the aliasing rate of the undersampling signal 1604, as described in the preceding sections with reference to the timing diagrams 1401 in FIG. 14A, 1407 in FIG. 14B, 1413 in FIG. 14C, and 1419 in FIG. 14D. The undersampling system 1602 outputs a downconverted signal 1308A.

Fig. 27 shows an undersampling system 2701 as a sample-and-hold system, which is an exemplary implementation of the undersampling system 2602. The undersampling system

2701 includes a switch module 2702 and a hold module 2706. The undersampling module 2701 is described below.

24A shows an undersampling system 2401 as a sequence-switched undersampling system, which is an alternative implementation of the undersampling system 2602. The sequence-switched undersampling system 2401 is described below.

4.1.1 The sample-hold system as a switching module and holding module

27 shows an exemplary embodiment of the sample and hold module 2604 of FIG. 26A. In the exemplary embodiment, the sample and hold module 2604 includes a switch module 2702 and a hold module 2706.

Preferably, the EM signal 1304 is subsampled by the switch module 2702 and the hold module 2706 to downconvert it in any of the manners illustrated in flow diagrams 1401, 1407, 1413, and 1419. For example, the sample and hold module 2604 may receive and subsample any of the modulated carrier signals described above, e.g., the analog AM signal 516, the digital AM signal 616, the analog FM signal 716, the digital FM signal 816, the analog PM signal 916, the digital PM signal 1016, etc., and any combinations thereof.

The EM signal 1304 is down-converted by the switch module 2702 and the hold module 2706 into an intermediate frequency signal, a demodulated baseband signal, or according to another modulation scheme depending on the aliasing rate.

The following example describes the operation of the switch module 2702 and the hold module for down-converting the EM signal 1304 into an intermediate frequency signal.

22 1001

gnal with reference to the flow chart 1407 and the exemplary timing diagrams in Figures 79A-F.

In step 1408, the switching module 2702 receives the EM signal 1304 (Fig. 79A). In step 1410, the switching module 1702 receives the undersampling signal 1604 (Fig. 79C). In step 1412, the switching module 2702 and the holding module 2706 cooperate to undersample and downconvert the EM signal 1304 to an intermediate frequency signal. That is, in step 1412, the switching module 2702 closes during each undersampling pulse to couple the EM signal 1304 to the holding module 2706. In one embodiment, the switching module 2702 closes on rising edges of the pulses. In an alternative embodiment, the switching module 2702 closes on falling edges of the pulses. When the EM signal 1304 is coupled to the hold module 2706, the amplitude of the EM signal 1304 is captured by the hold module 2706. The hold module 2706 is designed to capture and hold the amplitude of the EM signal 1304 within the short time frame specified by each of the pulses with a slight pulse width. Figure 79B shows the EM signal 1304 after the undersampling process.

The hold module 2706 essentially holds each subsampled amplitude until a subsequent undersampling pulse (Fig. 79D). The hold module 2706 outputs the subsampled amplitudes as the downconverted signal 1308A. The hold module 2706 may output the downconverted signal 1308A as an unfiltered signal, e.g., as a stepped signal (Fig. 79E), as a filtered downconverted signal (Fig. 79F), or as a partially filtered downconverted signal.

4.1.2 The sample-and-hold system as a sequence switching module

22 · lö-01

Figure 24A shows a sequence switching undersampling system 2401, which is an alternative implementation of the undersampling system 2602.

Preferably, the EM signal 1304 is subsampled by the sequence undersampling system 2401 to downconvert it in any of the manners illustrated in flow diagrams 1401, 1407, 1413, and 1419. For example, the sample and hold module 2604 may receive and subsample any of the unmodulated or modulated carrier signals described above, e.g., the analog AM signal 516, the digital AM signal 616, the analog FM signal 716, the digital FM signal 816, the analog PM signal 916, the digital PM signal 1016, etc., and any combinations thereof.

The EM signal 1304 is down-converted by the sequential switching undersampling system 2401 into an intermediate frequency signal, a demodulated baseband signal, or according to another modulation scheme depending on the aliasing rate.

24A shows a sequence switching switch 2402. The sequence switching switch 2402 includes a normally open switch 2404 and a normally closed switch 2406. The normally open switch 2404 is controlled by the undersampling signal 1604, as described above. The normally closed switch 2406 is controlled by a decoupling or separation signal 2412. In one embodiment, the decoupling or separation signal 2412 is generated from the undersampling signal 1604. Alternatively, the undersampling signal 1604 is generated from the decoupling or separation signal 2412. Alternatively, the decoupling or separation signal 2412 is generated independently of the undersampling signal 1604. The sequence switching module 2402 decouples or separates

a sample and hold input 2408 substantially from a sample and hold output 2410.

Figure 24B shows an exemplary timing diagram of the undersampling signal 1604 controlling the normally open switch 2404. Figure 24C shows an exemplary timing diagram of the decoupling or isolation signal 2412 controlling the normally closed switch 2406. The operation of the sequence switching module 2402 is described below with reference to the exemplary timing diagrams of Figures 24B and 24C.

Before a time t0, the normally open switch 2404 and the normally closed switch 2406 are set to their normal states.

At time t0, the decoupling or disconnecting signal 2412 in Figure 24C opens the normally closed switch 2406. Then, immediately after time t0, the normally open switch 2404 and the normally closed switch 2406 are open and the input 2408 is decoupled or disconnected from the output 2410.

At a time ti, the undersampling signal 1604 in Fig. 24B momentarily closes the normally open switch 2404. This couples the EM signal 1304 to the hold module 2416.

Before a time t2, the undersampling signal 1604 in Fig. 24B opens the normally open switch 2404. This decouples the EM signal 1304 from the hold module 2416.

At time t2, the decoupling or disconnect signal 2412 in Fig. 24C closes the normally closed switch 2406. This couples the holding module 2416 to the output 2410.

The sequence switching undersampling system 2401 includes a holding module 2416 which is similar to the holding module 2706 of Fig. 27

The sequence switching undersampling system 2401 downconverts the EM signal 1304 in a similar manner as described with reference to the undersampling system 2702 of FIG. 27.

4.1.3 Implementation examples of the switching module

The switching module 2702 of FIG. 27 and the switching modules 2404 and 2406 of FIG. 24A may be any type of switching element, preferably having a relatively low impedance when closed and a relatively high impedance when open. The switching modules 2702, 2404, and 2406 may be implemented by normally open or normally closed switches. The switching element need not be an ideal switching element. For example, FIG. 28B shows the switching modules 2702, 2404, and 2406 as a switching module 2810.

The switch element 2810 (e.g., switch modules 2702, 2404, and 2406) may be implemented by any suitable type of switch element, e.g., mechanical, electrical, optical switch elements, etc., and combinations thereof. Such elements include, for example, transistor switch elements, diode switch elements, relay switch elements, optical switch elements, micromachine switch elements, etc.

In one embodiment, the switching module 2810 may be implemented as a transistor, e.g., as a field effect transistor (FET), a bipolar transistor, or any other switching element.

In Fig. 28A, the switching module 2810 is shown as a FET 2802. The FET 2802 can be any type of FET, e.g. a MOSFET, a JFET, a GaAsFET, etc. The FET 2802 has a gate electrode 2804, a source electrode 2806

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and a drain electrode 2808. The gate electrode 2804 receives the undersampling signal 1604 for controlling the switching function between the source electrode 2806 and the drain electrode 2808. In general, the source electrode 2806 and the drain electrode 2808 are interchangeable.

The representation of the switching module 2810 as FET 2802 in Figure 28A is for illustrative purposes only. Any suitable element with switching characteristics could be used to implement the switching module 2810 (e.g., switching modules 2702, 2404, and 2406), as will be apparent to those skilled in the art based on the present description.

In Fig. 28C, the switching module 2810 is shown as a diode switch 2812, which acts as a two-terminal element when the undersampling signal 1604 is coupled to the output 2813.

In Fig. 28D, the switching module 2810 is shown as a diode switch 2814, which acts as a two-terminal element when the undersampling signal 1604 is coupled to the output 2815.

4.1.4 Implementation example of the holding module

The hold modules 2706 and 2416 preferably capture and hold the amplitude of the original, unaffected EM signal 1304 within the short time frame of each of the undersampled, slightly pulse width signal pulses.

In an exemplary embodiment, the holding modules 2706 and 2416 are implemented as a reactive holding module 2901 in Fig. 29A, although the invention is not limited to this embodiment. A reactive holding module is a holding module that uses one or more reactive electrical components to preferably quickly respond to the amplitude of the EM signal 1304. Reactive electrical

Components include capacitors and inductive components.

In one embodiment, the holding modules 2706 and 2416 include one or more capacitive holding elements, as shown in Figure 29B as capacitive holding module 2902. In Figure 29C, the capacitive holding module 2902 is shown as one or more capacitances, generally represented by capacitance(s) 2904. Recall that the purpose of the holding modules 2706 and 2416 is to quickly charge to the amplitude of the EM signal 1304. According to the principles of capacitances, because the slight pulse width of the undersampling pulses approaches zero over time, the capacitance value of capacitance 2904 may tend toward zero farads. Example capacitance values for capacitance 2904 may range from tens of picofarads to fractions of picofarads. A terminal 2906 serves as the output of the sample and hold module 2604. The capacitive hold module 2902 provides the undersampling pulses at terminal 2906 where they are measurable as a voltage value. Figure 29F shows the capacitive sample and hold module 2902 with a series capacitance 2912 which is usable in an inverted sample and hold system as described below.

In an alternative embodiment, the holding modules 2706 and 2416 may include one or more inductive holding elements, as shown in Figure 29D as inductive holding module 2908.

In an alternative embodiment, the holding modules 2706 and 2416 comprise a combination of one or more capacitive holding elements and one or more inductive holding elements, as shown in Figure 29E as capacitive/inductive holding module 2910.

Figure 29G shows an integrated undersampling system that can be implemented to downconvert the EM signal 1304 as shown in Figures 79A-F and described with reference to those figures.

4.1.5 Optional undersampling signal module

30 shows an undersampling system 3001 that is an exemplary embodiment of the undersampling system 1602. The undersampling system 3001 includes an optional undersampling signal module 3002 that can perform any of various functions or combinations of functions, e.g., can generate the undersampling signal 1604.

In one embodiment, the optional undersampling signal module 3002 includes a pulse width generator, an example of a pulse width generator being shown in Figure 29J as pulse width generator 2920. The pulse width generator 2920 generates pulses 2926 having slight pulse widths from an input signal 2924. The input signal 2924 may be any type of periodic signal, e.g., a sinusoidal signal, a square wave signal, a sawtooth wave signal, etc. Systems for generating the input signal 2924 are described below.

The width of the pulses 2926 is determined by the delay obtained by the tap 2922 of the pulse width generator 2920. In general, the tolerance requirements of the pulse width generator 2920 increase as the desired pulse width decreases. That is, to generate pulses with negligible pulse widths for a given EM input signal frequency, longer response times are required for the components used in the exemplary pulse width generator 2920, which are typically provided by more expensive

• · &Iacgr;&iacgr;&idiagr; · · · *

·

Elements can be obtained, e.g. by gallium arsenide (GaAs) elements, and similar elements.

The exemplary logic and implementation of pulse width generator 2920 are for illustrative purposes only and are not intended to be limiting. The actual logic used may take a variety of forms. The exemplary pulse width generator 2920 includes an optional inverter 2928 which is provided to enable polarity consistency with other examples presented herein. An example implementation of pulse width generator 2920 is shown in Figure 29K.

Further examples of pulse width generation logic are shown in Figures 29H and 29I. Figure 29H shows a rising edge pulse generator 2940 that generates pulses 2926 on rising edges of the input signal 2924. Figure 29I shows a falling edge pulse generator 2950 that generates pulses 2926 on falling edges of the input signal 2924.

In one embodiment, the input signal 2924 is generated externally from the undersampling signal module 3002, as shown in FIG. 30. Alternatively, the input signal 2924 is generated internally by the undersampling signal module 3002. The input signal 2924 may be generated by an oscillator, as shown in FIG. 29L by an oscillator 2930. The oscillator 2930 may be an internal or external device of the undersampling module 3002. The oscillator 2930 may be an external device of the undersampling module 3001.

The type of down-conversion performed by the undersampling system 3001 depends on the aliasing rate of the undersampling signal 1604, which is determined by the frequency of the pulses 2926. The frequency of the pulses 2926 is determined by the frequency of the input signal 2924. For example, if the frequency of the input signal 2924 of a

• 1«5Z « · ·

If the frequency of the input signal 2924 substantially equals a harmonic or subharmonic of the EM signal 1304, the EM signal 1304 is down-converted directly to a baseband signal (e.g., if the EM signal is an AM signal or a PM signal) or converted from an FM to a non-FM signal. If the frequency of the input signal 2924 substantially equals a harmonic or subharmonic of a difference frequency, the EM signal 1304 is down-converted to an intermediate frequency signal.

The optional 3002 undersampling signal module can be implemented in hardware, software, firmware, or any combination thereof.

4.2 The undersampling system as an inverted sample-and-hold circuit

Figure 26B shows an exemplary sample and hold system 2606, which is an alternative exemplary implementation of the undersampling module 1602.

Fig. 42 shows an inverted sample-and-hold system 4201, which is an implementation example of the inverted sample-and-hold system 2606 in Fig. 26B. The sample-and-hold system 4201 includes a sample-and-hold module 4202 with a switch module 4204 and a hold module 4206. The switch module 4204 can be implemented as described above with reference to Figs. 28A-D.

The holding module 4206 is implementable as described above with reference to Figures 29A-F for the holding modules 2706 and 2416. In the illustrated embodiment, the holding module 4206 includes one or more capacitors 4208. The capacitor(s) 4208 are selected to pass higher frequency components of the EM signal 1304 through a terminal 4210 regardless of the state of the switching module 4202. The capacitor 4202 stores

saves charge from EM signal 1304 during the aliasing pulses of undersampling signal 1604, and the signal at terminal 4210 is subsequently shifted by an amount related to the charge stored in capacitor 4208.

The operation of the inverted sample and hold system 4201 is illustrated in Figures 34A-F. Figure 34A shows an exemplary EM signal 1304. Figure 34B shows the EM signal 1304 after the undersampling process. Figure 34C shows the undersampling signal 1606 which comprises a train of aliased pulses with slight pulse widths.

Figure 34D shows an exemplary down-converted signal 1308. Figure 34E shows the down-converted signal 3308A in a compressed time scale. Because the hold module 4206 is a series element, the higher frequencies (e.g., RF frequencies) of the EM signal 1304 are visible on the down-converted signal. These can be filtered out as shown in Figure 34F.

The 4201 inverted sample and hold system can be used to down-convert any type of EM signal, such as modulated carrier signals and unmodulated carrier signals, into IF signals and demodulated baseband signals.

4.3 Other implementations

The implementations described above are illustrative. These implementations are not intended to limit the invention. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention that differ slightly or substantially from the implementations described herein.

5. Optional optimizations of the undersampling process with an aliasing rate

The methods and systems described in the preceding sections may optionally be optimized by one or more methods and systems as described below.

5.1 Doubling the aliasing rate (Far) of the undersampling signal

In one embodiment, the optional undersampling signal module 3002 of FIG. 30 includes a pulse generator module that generates aliasing pulses at a multiple of the frequency of the oscillating source, e.g., twice the frequency of the oscillating source. The input signal 2926 may be any suitable oscillating source.

31 shows an exemplary circuit 3102 that generates a doubler output signal 3104 (FIGS. 31 and 43B) that is usable as undersampling signal 1604. The exemplary circuit 3102 generates pulses at rising and falling edges of the oscillating input signal 3106 of FIG. 31B. The oscillating input signal 3106 is one embodiment of an optional input signal 2926. The circuit 3102 may be implemented as a pulse generator and aliasing rate ( ^f ar) doubler that provides the undersampling signal 1604 to the undersampling module 1606 of FIG. 30.

The aliasing rate corresponds to twice the frequency of the oscillating input signal F _O sc 3106, as shown by the following equation (9).

Eq. (9)

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The pulse width of the aliasing pulses is determined by the delay obtained by a first inverter 3108 of Figure 31. As the delay increases, the pulse width increases. A second inverter 3112 is shown which serves to maintain polarity consistency with respect to examples described elsewhere. In an alternative embodiment, inverter 3112 is omitted. The pulses preferably have marginal pulse widths whose duration tends toward zero. Doubler output signal 3104 may be further adapted to drive a switching module by pulses having marginal pulse widths. Circuit 3102 may be implemented by an integrated circuit, discretely, by an equivalent logic circuit, or by any suitable manufacturing technique.

5.2 Differential implementations

The invention is implementable by various differential configurations. Differential configurations are useful for reducing in-phase noise. This may be useful in receiver systems where in-phase interference is caused by intentional or accidental radiators, e.g., cellular phones, CB radios, electrical equipment, etc. Differential configurations are also useful for reducing any in-phase noise caused by carrier injection of the switch in the switching module or due to the design and construction of the system in which the invention is used. Any trace signals induced in equal magnitude and phase into both input terminals of the invention are substantially reduced or eliminated. Some differential configurations, e.g., some

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the configurations shown below are also suitable for increasing the voltage and/or increasing the power of the down-converted signal 1308A. The example of a differential undersampling module shown below is merely illustrative of the invention and is not intended to limit the invention. Those skilled in the art will appreciate, based on the present description, that alternative embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc.) of the embodiments described herein. The invention is intended and adapted to include these alternative embodiments.

44A shows an exemplary differential system 4402 that may be disposed in the undersampling module 1606. The differential system 4402 has an inverted undersampling configuration similar to the configuration described with reference to FIG. 42. The differential system 4402 has inputs 4404 and 4406 and outputs 4408 and 4410. The differential system 4402 has a first inverted sample and hold module 4412 with a hold module 4414 and a switch module 4416. The differential system 4402 also has a second inverted sample and hold module 4418 with a hold module 4420 and the switch module 4416 that it shares with the sample and hold module 4412.

One or both of the inputs 4404 and 4406 are connected to an EM signal source. The inputs may be connected to an EM signal source, with the input voltages at inputs 4404 and 4406 having substantially the same amplitude but 180 degrees out of phase with each other. Alternatively, if duplicate inputs are not available, one of the inputs 4404 and 4406 may be connected to ground.

• * I
• I

In operation, when the switch module 4416 is closed, the latch modules 4414 and 4420 are connected in series, and assuming they have similar capacitance values, the latch modules are charged to equal amplitude values but with opposite polarities. When the switch module 4416 is open, the voltage at the output 4408 is referenced to the input 4404, and the voltage at the output 4410 is referenced to the voltage at the input 4406.

The voltages at outputs 4408 and 4410 comprise a partial voltage obtained from the charge stored in holding modules 4414 and 4420, respectively, when switching module 4416 was closed. The partial voltages at outputs 4408 and 4410 obtained from the stored charge have substantially the same amplitude, but they are 180 degrees out of phase with each other.

The voltages at outputs 4408 and 4410 also include voltage ripple or noise components generated by the switching function of switching module 4416. However, because the switching module is located between the two outputs, the noise generated by the switching module appears at outputs 4408 and 4410 substantially equal and in phase relative to each other. This allows the voltage ripples to be substantially filtered out by inverting the voltage at one of outputs 4408 or 4410 and adding the voltage to the voltage at the other output. In addition, any noise coupled to input terminals 4404 and 4406 by any other noise sources of substantially equal amplitude and phase will tend to be canceled in the same manner.

The 4402 differential system is effective when used with a differential front end (inputs) and a differential

»1 T»Q ·

«1 YkD ·

rential backend (outputs). It can also be used in the following configurations, e.g.:

a) single-input frontend and differential backend; and

b) Differential frontend and single output backend.
Examples of these systems are described below.

5.2.1 Differential input to differential output

Figure 44B shows the differential system 4402 with the inputs 4404 and 4406 connected to equal and opposite EM signal sources, respectively, shown herein as dipole antennas 4424 and 4426. In this embodiment, when one of the output signals 4408 or 4410 is inverted and added to the other output signal, the in-phase noise generated by the switching module 4416 and other in-phase noise present at the input terminals 4404 and 4406 tend to be substantially cancelled.

5.2.2 Single input to differential output

Fig. 44C shows the differential system 4402 with the input 4404 connected to an EM signal source, e.g., a monopole antenna 4428, and the input 4406 connected to ground.

Figure 44E shows an exemplary single input to differential output receiver/downconverter system 4436. System 4436 includes differential system 4402 with input 4406 connected to ground. Input 4404 is connected to an EM signal source 4438.

The output signals 4408 and 4410 are coupled to a differential circuit 4444, e.g., a filter, which preferably inverts one of the output signals 4408 or 4410 and adds it to the other output signal 4408 or 4410. This substantially cancels the in-phase noise generated by the switching module 4416. The differential circuit 4444 preferably filters the higher frequency components of the EM signal 1304 that pass through the holding modules 4414 and 4420. The resulting filtered signal is output as the down-converted signal 1308A.

5.2.3 Differential Input to Single Output

Figure 44D shows the differential system 4402 with inputs 4404 and 4406 connected to equal and opposite EM signal sources, respectively, shown herein as dipole antennas 4430 and 4432. The output signal is obtained at terminal 4408.

5.3 Smoothing the down-converted signal

The down-converted signal 1308A may be smoothed by filtering if desired. The differential circuit 4444 implemented as a filter in Figure 44E _is merely an example. A filter function may be implemented in any of the described embodiments by hardware, firmware, and software implementations as is known in the art.

5.4 Load impedance and input/output buffering

• t

Some of the characteristics of the down-converted signal 1303A are dependent on the characteristics of a load coupled to the down-converted signal 1308A. For example, in one embodiment, when the down-converted signal 1308A is coupled to a high impedance load, the charge applied to a holding module, the holding module 2706 of FIG. 27 or 2416 of FIG. 24A, during a pulse is generally held by the holding module until the next pulse. This results in a substantially stepped representation of the down-converted signal 1308A, such as shown in FIG. 1bC . A high impedance load enables the undersampling system 1606 to accurately represent the voltage of the original, unaffected input signal.

The down-converted signal 1308A can be buffered by a high-impedance amplifier if necessary.

Alternatively, or in addition to buffering the down-converted signal 1308A, the EM input signal may be buffered or amplified by a low-noise amplifier.

5.5 Modification of the undersampling signal by feedback

30 shows an embodiment of a system 3001 in which the down-converted signal 1308 is used as a feedback signal 3006 for controlling various characteristics of the undersampling module 1606 for modifying the down-converted signal 1308A.

In general, the amplitude of the down-converted signal 1308A varies as a function of the frequency and phase differences between the EM signal 1304 and the undersampling signal 1604. In one embodiment, the down-converted signal 1308A is used as a feedback signal 3006 for controlling the frequency and phase relationship between the

·

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putty <· · · » ·«

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EM signal 1304 and undersampling signal 1604. This may be accomplished using the exemplary block diagram shown in Figure 32A. The exemplary circuit shown in Figure 32A may be located in undersampling signal module 3002. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention. In this embodiment, a state machine is used for illustration, but the invention is not intended to be limited thereby.

In the example of Figure 32A, the state machine 3204 reads an A/D converter 3202 and controls a digital-to-analog converter (DAC) 3206. In one embodiment, the state machine 3204 has two memory locations, Previous and Current , to store and retrieve the results of the read of the A/D converter 3202. In one embodiment, the state machine 3204 uses at least one memory flag.

The digital-to-analog converter DAC 3206 controls an input of a voltage controlled oscillator VCO 3208. The VCO 3208 controls a frequency input of a pulse generator 3210, which in one embodiment is substantially similar to the pulse generator shown in Figure 29J. The pulse generator 3210 generates the undersampling signal 1604.

In one embodiment, state machine 3204 performs processing in accordance with state machine flow diagram 3220 of Figure 32B. This processing serves to modify the frequency and phase relationship between undersampling signal 1604 and EM signal 1304 to maintain the amplitude of downconverted signal 1308A at substantially an optimal level.

The amplitude of the down-converted signal 1308A can be caused to vary with the amplitude of the

■ t
» »

Undersampling signal 1604. In an embodiment where the switching module 2702 is a FET as shown in Figure 28A, with the gate electrode 2804 receiving the undersampling signal 1604, the amplitude of the undersampling signal 1604 may determine the "ON" resistance of the FET, which affects the amplitude of the down-converted signal 1308A. The undersampling signal module 3002 shown in Figure 32C may be an analog circuit that enables an automatic gain control function. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention.

III. Downconversion through energy transfer

The power transfer embodiments of the present invention provide higher signal to noise ratios and higher sensitivity to very small signals, and allow the down-converted signal to drive lower impedance loads without assistance. The power transfer aspects of the present invention are generally illustrated by reference numeral 4506 in Figures 45A and 45B. Basic descriptions of how this can be accomplished are presented step by step below, beginning with a comparison to an undersampling system.

0.1 Energy transfer compared to undersampling

In Section II above, methods and systems for down-converting an EM signal by undersampling were described. The undersampling systems use a sample-and-hold circuit controlled by an undersampling signal.

&bull;&diams; · ·

system. The undersampling signal comprises a train of pulses with small pulse widths tending to zero in duration. The small pulse widths minimize the amount of energy carried by the EM signal. This protects the undersampled EM signal from being disturbed or destroyed. The small pulse widths also cause the sample and hold system to be a high impedance system. An advantage of the undersampling process is that the high impedance input allows accurate voltage reconstruction of the undersampled EM signal. The methods and systems described in Section II are suitable for many situations, such as monitoring EM signals without disturbing or destroying them.

Because the undersampling systems described in Section II transmit only small amounts of energy, they are not suitable for all situations. For example, in radio communications, received radio frequency (RF) signals are typically very weak and must be amplified to distinguish them from noise. The small amounts of energy transmitted by the undersampling systems described in Section II may not be sufficient to distinguish received RF signals from noise.

In accordance with one aspect of the invention, methods and systems are described below for down-converting EM signals by transferring non-trivial amounts of energy from the EM signals. The resulting down-converted signals have sufficient energy to enable the down-converted signals to be distinguishable from noise. The resulting down-converted signals also have sufficient energy to operate lower impedance circuits without buffering.

· V ·

Below, downconversion by energy transfer is introduced step by step to distinguish it from undersampling. The introduction begins with further descriptions of the undersampling technique.

0.1.1 Overview of the undersampling process

Figure 78A shows an exemplary undersampling system 7802 for down-converting an EM input signal 7804. The undersampling system 7802 includes a switching module 7806 and a holding module shown as a holding capacitor 7808. An undersampling signal 7810 controls the switching module 7806. The undersampling signal 7810 includes a train of pulses with minute pulse widths that tend toward zero in time. The minute pulse width or duration may be in the range of 1-10 ps for undersampling a 900 MHz signal, for example. Any other suitable minute pulse duration may be used if an exact reproduction of the original, unaffected input signal voltage is desired without significantly affecting the original input signal voltage.

In an undersampling environment, the holding capacitance 7808 preferably has a low capacitance value. This allows the holding capacitance 7808 to charge substantially to the voltage of the EM input signal 7804 during the small pulse widths of the undersampling signal pulses. For example, in one embodiment, the holding capacitance 7808 has a value in the range of 1 pF. Other suitable capacitance values may be used to substantially maintain the voltage of the original unaffected input signal. Different capacitances may be used for particular effects, as described below. The undersampling system is connected to a load

· ■

7812. In Fig. 78B, load 7812 of Fig. 78A is shown as high impedance load 7818. A high impedance load is a load that is relatively insignificant to an output drive impedance of the system for a given output frequency. High impedance load 7818 allows holding capacitance 7808 to substantially hold the charge accumulated during the undersampling pulses.

Figures 79A-F show exemplary timing diagrams for the undersampling system 7802. Figure 79A shows an exemplary EM input signal 7804.

Fig. 79C shows an exemplary undersampling signal 7810 having pulses 7904 with slight pulse widths whose duration tends toward zero.

Figure 79B shows the minor effects of the undersampled EM input signal 7804 measured at a terminal 7814 of the undersampling system 7802. In Figure 79B, minor perturbations 7902 are correlated with the pulses of the undersampling signal 7810. In this embodiment, the minor perturbations 7902 appear at different positions in successive cycles of the EM input signal 7804. This downconverts the EM input signal. The minor perturbations 7902 represent minor amounts of energy in the form of charge transferred to the holding capacitance 7808.

When the load 7812 is a high impedance load, the holding capacitance 7808 does not discharge significantly between pulses 7904. As a result, the sampled voltage value is "held" substantially constant at terminal 7816 until the next pulse 7904 by the charge transferred to the holding capacitance 7808 during a pulse 7904. When the voltage of the EM input signal 7804 changes between pulses 7904, the holding capacitance 7808 essentially reaches the new voltage value and the voltage obtained at terminal 7816

Voltage values form a step-like pattern as shown in Fig. 79D.

Fig. 79E shows the stepped voltage of Fig. 79D on a compressed time scale. The stepped voltage shown in Fig. 79E can be filtered to produce the signal shown in Fig. 79F. The signals shown in Figs. 79D, E and F have substantially all of the baseband characteristics of the EM input signal 7804 of Fig. 79A, except that the signals shown in Figs. 79D, E and F have been successfully downconverted.

The voltage level of the signals shown in Figures 79E and 79F is substantially the same as the voltage level of the EM input signal 7804. The EM input signal 7804 is therefore down-converted by the undersampling system 7802, which reproduces the voltage appropriately and does not significantly affect the EM input signal 7804. The power available at the output is relatively small (e.g., V ² /R; ~5 mV and 1 μΩ), given that the EM input signal 7804 typically has a drive impedance of 50 Ω (e.g., V ² /R; ~5 mV and 50 Ω) in an RF environment.

0.1.1.1 Effects of reducing the load impedance

Effects of reducing the impedance of the load 7812 are described below. Figures 80A-E show exemplary timing diagrams for the undersampling system 7802 when the load 7812 is a relatively low impedance load, i.e., having an impedance that is relatively significant for a given output frequency with respect to the output driver impedance of the system.

·

Fig. 8OA shows an exemplary EM input signal 7804 that is substantially similar to the signal shown in Fig. 79A.

80C shows an exemplary undersampling signal 7810 having pulses 8004 with slight pulse widths tending toward zero in duration. The exemplary undersampling signal 7810 shown in FIG. 80C is substantially similar to the signal shown in FIG. 79C.

Fig. 80B shows minor effects on the undersampled EM input signal 7804 measured at a terminal 7814 of the undersampling system 7802. In Fig. 80B, minor perturbations 8002 are correlated with the pulses 8004 of the undersampling signal 7810 of Fig. 80C. In this example, the minor perturbations 8002 appear at different positions in successive cycles of the EM input signal 7804. This downconverts the EM input signal 7804. The minor perturbations 8002 represent minor amounts of energy in the form of charge transferred to the holding capacitance 7808.

If the load 7812 is a low impedance load, the holding capacitance 7808 is substantially discharged by the load between pulses 8004 (Fig. 80C). As a result, the holding capacitance 7808 cannot adequately reach or "hold" the voltage of the original EM input signal 7804, unlike in the case of Fig. 79D. Instead, the charge appears as the output signal shown in Fig. 80D.

Fig. 80E shows the output signal of Fig. 80D on a compressed time scale. The output signal of Fig. 80E can be filtered to produce the signal shown in Fig. 80F. The down-converted signal shown in Fig. 80F is substantially similar to the down-converted signal shown in Fig. 79F, except that the signal shown in Fig. 80F has a much smaller amplitude.

than the signal shown in Fig. 79F. This is because the low impedance of the load 7812 prevents the holding capacitance 7808 from substantially reaching or "holding" the voltage of the original EM input signal 7804. Therefore, optimal voltage reproduction cannot be obtained by the down-converted signal shown in Fig. 80F and the signal has relatively low available power at the output (e.g.: V /R; -200 µV and 2 kΩ), given that the EM input signal 7804 typically has a drive impedance of 50 Ω (e.g.: V ² /R; ~5 mV and 50 Ω) in an RF environment.

0.1.1.2 Effects of increasing the holding capacity

worth

Described below are effects of increasing the values of the holding capacitances 7808 while driving a low impedance load 7812. Figures 8IA-F show exemplary timing diagrams for the undersampling system 7802 when the holding capacitance 7808 has a larger value in the range of, for example, 18 pF.

Figure 81A shows an exemplary EM input signal 7804 that is substantially similar to the signal shown in Figures 79A and 80A.

81C shows an exemplary undersampling signal 7810 having pulses 8104 with slight pulse widths tending toward zero in duration. The exemplary undersampling signal 7810 shown in FIG. 81C is substantially similar to the signal shown in FIGS. 79C and 80C.

Fig. 81B -shows the minor effects on the undersampled EM input signal 7804 measured at a terminal 7814 of the undersampling system 7802. In Fig. 81B,

minor perturbations 8102 are correlated with the pulses 8104 of the undersampling signal 7810 of Fig. 81C. Upon close inspection, the minor perturbations 8102 appear at different positions in successive cycles of the EM input signal 7804. This downconverts the EM input signal 7804. The minor perturbations 8102 represent minor amounts of energy in the form of charge transferred to the holding capacitance 7808.

Figure 81D shows the voltage measured at terminal 7816 as the holding capacitance 7808 attempts to reach and "hold" the voltage of the original EM input signal during the time windows defined by the minute pulse widths of the pulses 8104 shown in Figure 81C, but fails to do so.

As mentioned, when the load 7812 is a low impedance load, the holding capacitance 7808 is substantially discharged by the load between pulses 8104 (Fig. 81C), which case is illustrated in Figs. 81D and E. As a result, the holding capacitance 7808 cannot properly reach or "hold" the voltage of the original EM input signal 7804, unlike in the case of Fig. 79D. Instead, the charge appears as the output signal illustrated in Fig. 81D.

Fig. 81E shows the down-converted signal 8106 on a compressed time scale. The amplitude of the down-converted signal 8106 is much smaller than the amplitude of the down-converted signal shown in Figs. 80D and 80E. This is due to the higher capacitance value of the holding capacitance 7808. In general, as the capacitance value increases, more charge is required to increase the voltage for a time window defined by a given pulse width. Due to the small pulse width of the pulses 8104 of Fig. 81C, there is not enough

C* ·

Time is available to transfer significant amounts of energy or charge from the EM input signal 7804 to the holding capacitance 7808. As a result, the amplitudes produced by the holding capacitance 7808 are substantially smaller than the amplitudes of the down-converted signal shown in Figures 80D and 80E.

In Figures 80E and 80F, if the EM input signal 7804 typically has a driver impedance of 50 Ω (eg: V ² /R; ~5 mV and 50 Ω) in an RF environment, optimal voltage reproduction cannot be obtained from the unfiltered or filtered output signal and the signal has relatively negligible power available at the output (eg: V ² /R; -150 μV and 2 kΩ).

That is, undersampling systems, such as the 7802 undersampling system shown in Fig. 78, are suitable for down-converting EM signals with relatively accurate voltage reproduction. In addition, they have little effect on the original EM input signal. As shown above, the undersampling systems, such as the 7802 undersampling system shown in Fig. 78, are not well suited for power transfer or for driving loads with lower impedances.

0.1.2 Introduction to energy transfer

One embodiment of the invention relates to energy transfer from an EM signal using an energy transfer signal instead of an undersampling signal. Unlike undersampling signals, which comprise pulses with marginal pulse widths, the energy transfer signal comprises a sequence of pulses with non-marginal

&bull; ·· &bull;&bull;IDT.·« ,

Pulse widths tending away from zero in duration, allowing more time for energy to be transferred from an EM input signal. A direct benefit is that the input impedance of the system is reduced, allowing practical impedance matching circuits to be implemented to further improve energy transfer and thereby overall efficiency. The non-de minimis energy transferred significantly improves signal-to-noise ratio and response to very weak signals, and allows the down-converted signal to drive lower impedance loads without assistance. Examples of signals for which this is particularly advantageous include low-power signals, typically RF signals. One benefit of a non-de minimis pulse width is that phase noise in the energy transfer signal does not have as significant an effect on the down-converted output signal as the phase noise of the undersampled signal or the phase noise of a conventional sampling signal have on their respective output signals.

82A shows an exemplary energy transfer system 8202 for down-converting an EM input signal 8204. The energy transfer system 8202 includes a switching module 8206 and a storage module shown as storage capacitance 8208. As used herein, the terms "storage module" and "storage capacitance" are to be distinguished from the terms "holding module" and "holding capacitance," respectively. The holding modules and holding capacitances shown above indicate systems that store small amounts of energy from a subsampled EM input signal with the intent of "holding" a voltage value. In contrast, storage modules and storage capacitances indicate systems that store non-small amounts of energy from an EM input signal.

The energy transfer system 8202 receives an energy transfer signal 8210 that controls the switching module 8206. The energy transfer signal 8210 comprises a train of energy transfer pulses with non-minor pulse widths that trend away from . The non-minor pulse widths may have any duration. For example, the non-minor pulse widths may correspond to half the period of the EM input signal. Alternatively, the non-minor pulse widths may correspond to any other fraction of a period of the EM input signal, or a multiple of a period plus a fraction. In an exemplary embodiment, the EM input signal has a frequency of about 900 MHz and the non-minor pulse width is about 550 ps. Any other suitable non-minor pulse duration may be used.

In a power transfer environment, the storage module, shown in Figure 82 as a storage capacitance 8208, preferably has a capacitance value that can handle the power transferred and that can transfer a non-trivial amount of energy or power during the time window defined by a non-trivial pulse width. This allows the storage capacitance 8208 to store energy transferred by the EM input signal 8204 without significant consideration of exactly reproducing the original, unaffected voltage level of the EM input signal 8204. For example, in one embodiment, the storage capacitance 8208 has a capacitance value in the range of 18 pF. Other suitable capacitance values and storage modules may be used.

An advantage of the energy transfer system 8202 is that, even if the EM input signal 8204 is a very

&bull; -P c -j · · .

weak signal, the energy transfer system 8202 transfers sufficient energy from the EM input signal 8204 so that the EM input signal can be efficiently down-converted.

The energy transfer system 8202 is coupled to a load 8212. As shown above in an overview of the undersampling system, these loads can be classified as high impedance loads or low impedance loads. A high impedance load is a load that is relatively insignificant to an output driver impedance of the system for a given output frequency. A low impedance load is a load that is relatively significant. Another advantage of the energy transfer system 8202 is that the non-trivial amounts of energy transferred enable the energy transfer system 8202 to drive loads that would otherwise be classified as low impedance loads in undersampling systems and conventional sampling systems. That is, the non-trivial amounts of energy transferred ensure that the storage capacitance 8208 will accept and hold sufficient energy or charge to drive the load 8202 even for lower impedance loads. This is illustrated below in the timing diagrams of Figures 83A-F.

Figures 83A-F show exemplary timing diagrams for the energy transfer system 8202 in Figure 82. Figure 83A shows an exemplary EM input signal 8302.

Fig. 83C shows an exemplary undersampling signal 8304 with energy transfer pulses 8306 with non-negligible pulse widths whose duration tends away from zero.

Fig. 83B shows the effects on the EM input signal 8302 measured at a terminal 8214 in Fig. 82A when non-trivial amounts of energy are transferred from it. In Fig. 83B, non-trivial disturbances 8308 are associated with the

Energy transfer pulses 8306 in Fig. 83C. In this example, the non-minor perturbations 8303 appear at different positions in successive cycles of the EM input signal 8302. The non-minor perturbations 8308 represent non-minor amounts of transferred energy in the form of charge transferred to the storage capacitance 8208 of Fig. 82.

Fig. 83D shows a down-converted signal 8310 formed by energy transferred from the EM input signal 8302.

83E shows the down-converted signal 8310 in a compressed scale. The down-converted signal 8310 can be filtered to produce the down-converted signal 8312 shown in FIG. 83F. The down-converted signal 8312 is similar to the down-converted signal shown in FIG. 79F, except that the down-converted signal 8312 has significantly more power (eg: V ² /R; about (~) 2 mV and 2 kΩ) than the down-converted signal shown in FIG. 79F (eg: V ² /R; -5 mV and 1 μΩ). This allows the down-converted signals 8310 and 8312 to efficiently drive lower impedance loads, provided that the EM input signal 8204 in an RF environment typically has a driver impedance of 50 Ω (eg: V ² /R; ~5

mV and 50 Ω).

The energy transfer aspects of the invention are generally illustrated by reference numeral 4506 in Figures 45A and 45B.

1. Downconversion of an EM signal into an IF EM intermediate frequency signal by transferring energy from the EM signal at an aliasing rate

In one embodiment of the invention, an EM signal is down-converted to an IF signal by transferring energy from the EM signal at an aliasing rate. This embodiment is illustrated in Figure 45B by reference numeral 4514.

This embodiment is implementable with any type of EM signal, e.g., modulated carrier signal or unmodulated carrier signal. This embodiment is described herein using the modulated carrier signal F _MC of Fig. 1 as an example. In the example, a modulated carrier signal F _MC is down-converted to an intermediate frequency (IF) signal F _{x F.} The intermediate frequency signal F _{x F} can be demodulated to a baseband signal F _DMB using conventional demodulation techniques. Those skilled in the art will appreciate from this description that the invention is implementable to down-convert any EM signal, e.g., a modulated carrier signal and an unmodulated carrier signal.

In the following sections, methods for down-converting an EM signal to an IF signal F _IF by transferring energy from the EM signal at an aliasing rate are described. Exemplary structural embodiments for implementing the methods are also described. However, the invention is not limited to the specific embodiments described below. It will be apparent to those skilled in the art based on the present description that equivalent implementations, extensions, changes, modifications, etc. are possible within the scope of the present invention.

The following sections include a high-level description, exemplary embodiments, and implementation examples.

1.1 High-level description

Through this section (including its subsections), a high-level description of the inventive downconversion of an EM signal to an IF signal Fjf by energy transfer is provided. In particular, an operating method for downconverting the modulated carrier signal F _MC to the modulated IF carrier signal F _1F by energy transfer is presented in a high-level description. In addition, a structural implementation of this method is presented in a high-level description. This structural implementation is described herein for illustration and is not intended to be limiting. In particular, the method described in this section may be realized using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to those skilled in the art based on the present description.

1.1.1 Description of the functionality

46B shows a flow diagram 4607 illustrating an exemplary method for down-converting an EM signal to an intermediate frequency signal F _XF by transferring energy from the EM signal at the aliasing rate. The exemplary method illustrated in flow diagram 4607 is an embodiment of the flow diagram 4601 of FIG. 46A.

Any and all combinations of modulation techniques are applicable to the present invention. For ease of description, the digital AM carrier signal 616 is used for a high-level functional description of the invention. Detailed timing diagrams and descriptions for examples of AM, FM and PM embodiments are provided throughout the following sections. Those skilled in the art will appreciate from this description that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and an unmodulated carrier signal.

The method illustrated in flow diagram 4607 is illustrated below with a high-level description using the digital AM carrier signal 616 of FIG. 6C. Detailed flow diagrams and descriptions are provided throughout the following sections for example AM, FM, and PM embodiments. Those skilled in the art will appreciate from this description that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and an unmodulated carrier signal.

Processing begins in step 4608 where an EM signal is received. Step 4608 is represented by the digital AM carrier signal 616. The digital AM carrier signal 616 of FIG. 6C is re-illustrated in FIG. 47A for simplicity. FIG. 47E shows a portion of the digital AM carrier signal 616 on an expanded time scale.

In step 4610, an energy transfer signal is received that has an aliasing rate F _AR . Fig. 47B shows an exemplary energy transfer signal 4702. The energy

Energy transfer signal 4702 comprises a train of energy transfer pulses 4704 having non-minor pulse widths 4701 whose duration tends away from zero. Pulse widths 4701 may generally have any duration other than the period of the EM signal. For example, pulse widths 4701 may be greater or less than a period of the EM signal. Therefore, pulse widths 4701 may be approximately 1/10, 1/4, 1/2, 3/4, etc., or any other fraction of the period of the EM signal. Alternatively, pulse widths 4701 may be one or more periods of the EM signal plus 1/10, 1/4, 1/2, 3/4, etc., or any other fraction of a period of the EM signal. The 4701 pulse widths can be optimized based on one or more of various criteria, as described in the following sections.

The energy transfer pulses 4704 repeat at the aliasing rate. A suitable aliasing rate can be determined or selected as described below. In general, when an EM signal is down-converted to an intermediate frequency signal, the aliasing rate is substantially equal to a difference frequency, described below, or a harmonic or, more typically, a subharmonic of the difference frequency.

In step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _XF . Figure 47C shows the transferred energy 4706 transferred from the EM signal during the energy transfer pulses 4704. Because a harmonic of the aliasing rate is offset from the frequency of the AM signal 616, the pulses 4704 traverse the AM signal 6116 at the offset frequency. By "traversing" the AM signal 616, the transferred energy 4706 forms an AM intermediate frequency signal 4706 that corresponds to the AM

carrier signal 616 except that the AM intermediate frequency signal has a lower frequency than the AM carrier signal 616. The AM carrier signal 616 can be down-converted to any frequency below that of the AM carrier signal 616 by adjusting the aliasing rate F _AR as described below.

Fig. 47D shows the AM intermediate frequency signal 4706 as a filtered output signal 4708. In an alternative embodiment, a stepped or unfiltered output signal is output according to the invention. The choice between a filtered, a partially filtered and an unfiltered output signal is generally determined by the design, which depends on the application of the invention.

The intermediate frequency of the down-converted signal F _XF , which in this example is the intermediate frequency signal 4706 and 4703, can be determined according to Eq. (2), which is repeated below for simplicity.

Eq. (2)

A suitable aliasing rate F^ can be determined in several ways. An exemplary method for determining the aliasing rate F _AR is shown below. Those skilled in the art will appreciate from this description how to determine suitable aliasing rates for EM signals, even for signals other than the modulated carrier signals specifically described herein.

In Fig. 48, a flow chart 4801 shows an exemplary processing for determining an aliasing rate F _AR . However, a designer may determine or

The application may specify that the values be determined in a sequence different from that shown. Processing begins in step 4802 where the frequency of the EM signal is determined or selected. For example, the frequency of the AM carrier signal 616 may be 901 MHz.

In step 4804, the intermediate frequency is determined or selected. This is the frequency to which the EM signal is down-converted. The intermediate frequency may be determined or selected to match a frequency requirement of a downstream demodulator. For example, the intermediate frequency may be 1 MHz.

In step 4806, the aliasing rate or rates by which the EM signal is down-converted to the IF signal specified in step 4804 are determined.

Eq. (2) can be rewritten as Eq. (3):

= Fc ± F _IF Eq. (3)

This equation can be rewritten as Eq. (4):

Eq. (4)

or in Eq. (5):

—Eq. (5)

± Fif) can be defined as the difference value Fqiff, as shown in Eq. (6):

(Fc±F _IF ) = F _DIFF Eq. (6)

Eq. (4) can be rewritten as Eq. (7):

^ Eq. (7)

^r AR

According to equation (7), it can be seen that F _DIFF is constant for a given value η and a constant value of F _AR . For the case F _DIFF = F _c - F _IF and for a constant value of Fdiff , as F _c increases, F _IF necessarily increases. For the case F _DIFF = F _c + Fif and for a constant value of F _D j _FF , as Fc increases, F _IF necessarily decreases. In the latter case F _DIFF = F _c + F _{IF ,} any phase or frequency changes of F _c correspond to an inverse or inverted phase or frequency change of F _IF . This is mentioned to clarify that when F _DIFF = F _c - F _IF is used, the phase and frequency behavior of the modulated intermediate frequency signal F _IF is influenced by the above-mentioned effect.

Equations (2) to (7) can be solved for any permissible value of η. An appropriate value of η can be determined for any given difference frequency F _DJFF and for any desired aliasing rate F _AR(desired) . Equations (2) to (7) can be used to identify a particular harmonic closest to the desired aliasing rate F _AR(desired ) that will produce the desired intermediate frequency signal F _IE - .

An example of determining an appropriate value η for a given difference frequency F _D i _FF and for a desired aliasing rate F _AR(desired) is shown below. For ease of description, only the case (F _c - F _IF ) is shown in the example below.

F-F F

C ¹ IF _ ^l

DIFF

F F

AR(desired) AR(desired)

For example, the desired aliasing rate F _AR(desired) may be 140 MHz. Using the above examples where the carrier frequency is 901 MHz and the intermediate frequency is 1 MHz, the initial value of η is determined as:

901MHz-IMHz 900 _CA

η = = = 6.4

140MHz 140

The initial value 6.4 can be rounded up or down to the next valid value n defined above as e.g. (0.5, 1, 2, 3, ...). In this example, the value 6.4 is rounded down to 6.0 and this value is substituted into Eq. (5) for the case (F _c - F _IF ) = F _DIFF .

F-F

_, 901MHz-IMHz 900MHz _{&igr;&sfgr;&eegr;&lgr;/&Ggr;&ugr;}
Far = Z = Z = 150µ&EEgr;&zgr;

That is, by transferring energy from a 901 MHz EM carrier signal having a frequency of 150 MHz, an intermediate frequency signal having a frequency of 1 MHz is generated. If the EM carrier signal is a modulated carrier signal, the intermediate frequency signal will also substantially exhibit the modulation. The modulated intermediate frequency signal can be demodulated by any conventional demodulation technique.

Alternatively, instead of starting from a desired aliasing rate, a list of suitable aliasing rates can be determined from the modified form of Eq. (5) by determining a solution for various values of η. Example solutions are shown below.

- ( ^f c- ^f if) _ ^F diff _ 901MHz-IMHz __ 900MHz
&eegr;&eegr;&eegr;&eegr;

Solution for &eegr; =0.5, 1, 2, 3, 4, 5 and 6:

900 MHz/0.5 = 1.8 GHz (ie, second harmonic);
900 MHz/1 = 900 MHz (ie, fundamental frequency);
900 MHz/2 = 450 MHz (ie, second subharmonic);
900 MHz/3 = 300 MHz (ie, third subharmonic);
900 MHz/4 = 225 MHz (ie, fourth subharmonic);
900 MHz/5 = 180 MHz (ie, fifth subharmonic);
900 MHz/6 = 150 MHz (i.e., sixth subharmonic).

The steps described above can be carried out in a similar manner for the case (Fc + Fi _F ). The results can be compared with the results obtained in the case of (F _c - F _IF ) to determine which result is more suitable for an application.

In one embodiment, according to the invention, an EM signal is down-converted to a relatively standard IF signal having a frequency of, for example, 100 kHz to 200 MHz. In another embodiment, referred to herein as a small offset implementation, according to the invention, an EM signal is down-converted to an IF signal having a relatively low frequency of, for example, less than 100 kHz. In another embodiment, referred to herein as a large offset implementation, according to the invention, an EM signal is down-converted to an IF signal having a relatively higher frequency of, for example, more than 200 MHz.

The different offset implementations provide choices for different applications. In general, lower data rate applications can operate at lower intermediate frequencies. However, higher intermediate frequencies can support a larger amount of information for a given modulation scheme.

According to the invention, a designer selects an optimal information bandwidth for an application and an optimal intermediate frequency to support the baseband signal. The intermediate frequency should be sufficiently high to support the bandwidth of the modulating baseband signal F _MB .

In general, the frequency of the down-converted IF signal decreases as the aliasing rate approaches a harmonic or subharmonic frequency of the EM signal. Similarly, the frequency of the down-converted IF signal decreases as the aliasing rate moves away from a harmonic or subharmonic frequency of the EM signal.

Aliased frequencies occur above and below each harmonic of the aliasing frequency. To avoid mixing or mapping other aliasing frequencies into the band of the aliasing frequency (IF) of interest, the intermediate frequency should preferably not be near half the aliasing rate.

As described in implementation examples below, an aliasing module comprising a universal frequency translation (UFT) module constructed in accordance with the invention provides a wide range of flexibility in frequency selection, making it suitable for a wide range of applications. Conventional systems do not readily provide or enable this level of flexibility in frequency selection.

1.1.2 Description of the structure

Fig. 63 shows a block diagram of an energy transfer system 6302 according to an embodiment of the invention. The energy transfer system 6302 is an exemplary embodiment of the general aliasing system 1302 of Fig. 13. The energy transfer system 6302 comprises an energy transfer module 6306. The energy transfer module 6306 receives the EM signal 1304 and an energy transfer signal 6304 comprising a train of energy transfer pulses with non-minor pulse widths whose

duration tending away from zero and occurring at a frequency equal to the aliasing rate F _m . The energy transfer signal 6306 is an exemplary embodiment of the aliasing signal 1310 of Fig. 13. The energy transfer module 6306 transfers energy from the EM signal 1304 at the aliasing rate F _AR of the energy transfer signal 6306.

Preferably, the energy transfer module 6304 transfers energy from the EM signal 1304 to down-convert it to the intermediate frequency signal Fi _F in the manner illustrated in the flow diagram 4607 of FIG. 46B. Other structural embodiments for carrying out the steps of the flow diagram 4607 are possible within the scope of the invention. The details of the other structural embodiments will be apparent to those skilled in the art based on the description presented herein.

The operation of the energy transfer system 6302 is described below with reference to the flowchart 4607 and the timing diagrams in Figures 47A-E. In step 4608, the energy transfer module 6304 receives the AM carrier signal 616. In step 4610, the energy transfer module 6304 receives the energy transfer signal 4702. In step 4612, the energy transfer module 6304 transfers energy from the AM carrier signal 616 at the aliasing rate to down-convert the AM carrier signal 616 to the intermediate frequency signal 4706 or 4708.

Implementation examples of the energy transfer module 6302 are presented in sections 4 and 5 below.

1.2 Exemplary embodiments

In this section (and its subsections), various embodiments related to the method(s) and structure(s) described above are presented. These embodiments are described herein for illustrative purposes and are not to be taken in a limiting sense. The invention is not limited to these embodiments. It will be apparent to those skilled in the art based on the present description that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and adapted to include these alternative embodiments.

The method for downconverting the EM signal 1304 by energy transfer can be implemented with any type of EM signal, such as modulated EM carrier signals and unmodulated EM carrier signals. For example, the method according to flowchart 4601 can be implemented to downconvert AM, FM, PM carrier signals, etc., or any combination thereof. The processing of flowchart 4601 of FIG. 46A for downconverting AM, FM, and PM signals is described below. The downconversion methods described include downconversion to intermediate frequency signals, direct downconversion to demodulated baseband signals, and downconversion of FM signals to non-FM signals. The exemplary descriptions below are provided to illustrate the present invention. The present invention is not limited by or to the exemplary embodiments below.

1.2.1 First exemplary embodiment: amplitude modulation
1.2.1.1 Description of the functionality

The exemplary processing of the flow chart 4607 in Fig. 46B for the analog AM carrier signal 516 shown in Fig. 5C and for the digital AM carrier signal 616 shown in Fig. 6C will be described below.

1.2.1.1.1 Analog AM carrier signal

A method for down-converting the analog AM carrier signal 516 of FIG. 5C to an analog AM intermediate frequency signal is described below with reference to the flow chart 4607 of FIG. 46B. The analog AM carrier signal 516 is shown again in FIG. 50A for simplicity. In this example, the analog AM carrier signal 516 oscillates at a frequency of about 901 MHz. In FIG. 50B, an analog AM carrier signal 5004 represents a portion of the analog AM carrier signal 516 on an expanded time scale.

Processing begins in step 4608 where the EM signal is received. This is represented by the analog AM carrier signal 516.

In step 4610, an energy transfer signal is received having an aliasing rate F _AR . Fig. 50C shows an exemplary energy transfer signal 5006 at approximately the same time scale as in Fig. 50B. The energy transfer signal 5006 comprises a train of energy transfer pulses 5007 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 5007 repeat at the aliasing rate F _AR ,

which is determined or selected as described above. In general, the aliasing rate F _AR when downconverting to an intermediate frequency signal is substantially equal to a harmonic or, more typically, a subharmonic of the difference frequency F _DIFF .

In step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _IF . In Fig. 50D, an affected analog AM carrier signal 5008 shows effects of energy transfer from the analog AM carrier signal 516 at the aliasing rate. The affected analog AM carrier signal 5008 is shown at substantially the same time scale as in Figs. 50B and 50C.

Fig. 50E shows a down-converted AM intermediate frequency signal 5012 produced by the down-conversion process. The AM intermediate frequency signal 5012 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The down-converted signal 5012 has portions 5010A correlated with the energy transfer pulses 5007 in Figure 50C and portions 5010B between the energy transfer pulses 5007. The portions 5010A represent energy transferred from the analog AM signal 516 to a storage device while simultaneously driving an output load. The portions 5010A appear when a switching module is closed by the energy transfer pulses 5007. The portions 5010B represent energy stored in a storage device to continue driving the load. The portions 5010B appear when the switching module is opened after energy transfer pulses 5007.

&bull; « i

Because a harmonic of the aliasing rate is offset from the analog AM carrier signal 516, the energy transfer pulses 5007 "traverse" the analog AM carrier signal 516 at the difference frequency F _DIFF . That is, the energy transfer pulses 5007 appear at different positions in successive cycles of the AM carrier signal 516. Therefore, different amounts of energy are captured by the energy transfer pulses 5007 from the analog AM signal 516, as represented by portions 5010A, thereby giving the AM intermediate frequency signal 5012 an oscillation frequency F _IF .

In FIG. 50F, an AM intermediate frequency signal 5014 represents the AM intermediate frequency signal 5012 in a compressed time scale. In FIG. 50G, an AM intermediate frequency signal 5016 represents a filtered version of the AM intermediate frequency signal 5014. The AM intermediate frequency signal 5016 is substantially similar to the AM carrier signal 5016 except that the AM intermediate frequency signal 5016 has an intermediate frequency. The AM intermediate frequency signal 5016 may then be demodulated by any conventional demodulation technique.

The present invention can output the unfiltered AM intermediate frequency signal 5014, the filtered AM intermediate frequency signal 5016, a partially filtered AM intermediate frequency signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The signals referred to herein represent frequency down-conversion in accordance with the invention. For example, AM intermediate frequency signals 5014 in Fig. 50F and 5016 in Fig. 50G represent that AM carrier signal 516 has been successfully down-converted to an intermediate frequency signal.

while maintaining sufficient baseband information for adequate reconstruction.

1.2.1.1.2 Digital AM carrier signal

A method for down-converting the digital AM carrier signal 616 to a digital AM intermediate frequency signal is described below with reference to the flow chart 4607 of FIG. 46B. The digital AM carrier signal 616 is shown again in FIG. 51A for simplicity. In this example, the digital AM carrier signal 616 oscillates at a frequency of about 901 MHz. In FIG. 51B, a digital AM carrier signal 5104 represents a portion of the digital AM carrier signal 616 on an expanded time scale.

Processing begins in step 4608 where an EM signal is received. This is represented by the digital AM carrier signal 616.

In step 4610, an energy transfer signal is received having an aliasing rate F _AR . Figure 51C shows an exemplary energy transfer signal 5106 at approximately the same time scale as in Figure 51B. The energy transfer signal 5106 comprises a train of energy transfer pulses 5107 with non-minor pulse widths 5109 tending away from zero in duration. The energy transfer pulses 5107 repeat at the aliasing rate F _AR determined or selected as described above. In general, the aliasing rate when downconverting to an intermediate frequency signal is substantially equal to a harmonic, or more typically a subharmonic, of the difference frequency F _DIFF .

In step 4612, energy is transferred from the EM signal at the aliasing rate to convert the EM signal into the intermediate frequency

frequency signal F _XF . In Fig. 51D, an influenced digital AM carrier signal 5108 shows effects of energy transfer from the digital AM carrier signal 616 at the aliasing rate F _A r. The influenced digital AM carrier signal 5108 is shown on substantially the same time scale as in Figs. 51B and 51C.

Figure 51E shows a down-converted AM intermediate frequency signal 5112 produced by the down-conversion process. The AM intermediate frequency signal 5112 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The down-converted signal 5112 has portions 5110A correlated with the energy transfer pulses 5107 in Figure 51C and portions 5110B between the energy transfer pulses 5107. The portions 5110A represent energy transferred from the digital AM carrier signal 616 to a storage device while simultaneously driving an output load. The portions 5110A appear when a switching module is closed by the energy transfer pulses 5107. The portions 5110B represent energy stored in a storage device to continue driving the load. The portions 5110B appear when the switching module is opened after the energy transfer pulses 5107.

Because a harmonic of the aliasing rate is offset from the AM carrier signal 616, the energy transfer pulses 5107 "travel" the digital AM carrier signal 616 at the difference frequency F _DIFF . That is, the energy transfer pulses 5107 appear at different positions in successive cycles of the digital AM carrier signal 616. Therefore, different amounts of energy are transferred from the digital AM carrier signal 616 by the energy transfer pulses 5107.

detected as represented by portions 5110A, thereby giving the AM intermediate frequency signal 5112 an oscillation frequency F _IF .

In FIG. 51F, an AM intermediate frequency signal 5114 represents the AM intermediate frequency signal 5112 in a compressed time scale. In FIG. 51G, an AM intermediate frequency signal 5116 represents a filtered version of the AM intermediate frequency signal 5114. The AM intermediate frequency signal 5116 is substantially similar to the AM carrier signal 616, except that the AM intermediate frequency signal 5116 has an intermediate frequency. The AM intermediate frequency signal 5116 may then be demodulated by any conventional demodulation technique.

The present invention can output the unfiltered AM intermediate frequency signal 5114, the filtered AM intermediate frequency signal 5116, a partially filtered AM intermediate frequency signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The signals referred to herein represent frequency down-conversion in accordance with the present invention. For example, AM intermediate frequency signals 5114 in FIG. 51F and 5116 in FIG. 51G represent that AM carrier signal 616 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.1.2

Description of the structure

The operation of the energy transfer system 6302 for the analog AM carrier signal 516 is described below with reference to the flow chart 4607 and the timing diagrams of the

Figures 50A-G. In step 4608, the energy transfer module 6304 receives the analog AM carrier signal 516. In step 4610, the energy transfer module 6304 receives the energy transfer signal 5006. In step 4612, the energy transfer module 6304 transfers energy from the analog AM carrier signal 516 at the aliasing rate of the energy transfer signal 5006 to down-convert the analog AM carrier signal 516 to the AM intermediate frequency signal 5012.

The operation of the energy transfer system 6302 for the digital AM carrier signal 616 is described below with reference to the flowchart 1401 and the timing diagrams of Figures 51A-G. In step 4608, the energy transfer module 6304 receives the digital AM carrier signal 616. In step 4610, the energy transfer module 6304 receives the energy transfer signal 5106. In step 4612, the energy transfer module 6304 transfers energy from the digital AM carrier signal 616 at the aliasing rate of the energy transfer signal 5106 to down-convert the digital AM carrier signal 616 to the AM intermediate frequency signal 5112.

Example embodiments of the energy transfer module 6304 are presented in sections 4 and 5 below.

1.2.2 Second exemplary embodiment: frequency modulation
1.2.2.1 Description of the functionality

The following illustrates the operation of the exemplary processing of the flow chart 4607 of FIG. 46B for the analog FM carrier signal 716 shown in FIG. 7C and for the digital FM carrier signal 816 shown in FIG. 8C.

1.2.2.1.1 Analog FM carrier signal

A method for down-converting the analog FM carrier signal 716 shown in FIG. 7C to an FM intermediate frequency signal will now be described with reference to the timing diagram 4607 in FIG. 46B. The analog FM carrier signal 716 is shown again in FIG. 52A for simplicity. In this example, the analog FM carrier signal 716 oscillates at a frequency of about 901 MHz. In FIG. 52B, an analog FM carrier signal 5204 represents a portion of the analog FM carrier signal 716 on an expanded time scale.

Processing begins in step 4608 where an EM signal is received. This is represented by the analog FM carrier signal 716.

In step 4610, an energy transfer signal is received having an aliasing rate F _AR . Figure 52C shows an exemplary energy transfer signal 5206 at approximately the same time scale as Figure 52B. The energy transfer signal 5206 comprises a train of energy transfer pulses 5207 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 5207 repeat at the aliasing rate F _AR determined or selected as described above. In general, the aliasing rate F AR is substantially equal to a harmonic, or typically a subharmonic, of the difference frequency F _DIFF when downconverted to an intermediate frequency signal.

In step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to an intermediate frequency signal F _IF . In Fig. 52D, an affected analog FM carrier signal 5208 shows effects of the energy

energy transfer from the analog FM carrier signal 716 at the aliasing rate F _AR . The influenced analog FM carrier signal 5208 is shown at substantially the same time scale as in Figures 52B and 52C.

Figure 52E shows a down-converted FM intermediate frequency signal 5212 produced by the down-conversion process. The FM intermediate frequency signal 5212 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The down-converted signal 5212 has portions 5210A correlated with the energy transfer pulses 5207 in Figure 52C and portions 5210B between the energy transfer pulses 5207. The portions 5210A represent energy transferred from the analog FM carrier signal 716 to a storage device while simultaneously driving an output load. The portions 5210A appear when a switching module is closed by the energy transfer pulses 5207. The portions 5210B represent energy stored in a storage device to continue driving a load. The portions 5210B appear when the switching module is opened after the energy transfer pulses 5207.

Because a harmonic of the aliasing rate is offset from an analog FM carrier signal 716, the energy transfer pulses 5207 "travel" the analog FM carrier signal 716 at the difference frequency F _D i _FF . That is, the energy transfer pulses 5207 appear at different positions in successive cycles of the analog FM carrier signal 716. Therefore, different amounts of energy are captured by the energy transfer pulses 5207 from the analog FM carrier signal 716, as represented by portions 5210, thereby

the FM intermediate frequency signal 5212 receives an oscillation frequency F _1F .

In FIG. 52F, an analog FM intermediate frequency signal 5214 represents the FM intermediate frequency signal 5212 in a compressed time scale. In FIG. 52G, an FM intermediate frequency signal 5216 represents a filtered version of the FM intermediate frequency signal 5214. The FM intermediate frequency signal 5216 is substantially similar to the analog FM carrier signal 716, except that the FM intermediate frequency signal 5216 has an intermediate frequency. The FM intermediate frequency signal 5216 may then be demodulated by any conventional demodulation technique.

The present invention can output the unfiltered FM intermediate frequency signal 5214, the filtered FM intermediate frequency signal 5216, a partially filtered FM intermediate frequency signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The signals referred to herein represent frequency down-conversion in accordance with the present invention. For example, FM intermediate frequency signals 5214 in FIG. 52F and 5216 in FIG. 52G represent that FM carrier signal 716 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.2.1.2 Digital FM carrier signal

A method for down-converting the digital FM carrier signal 816 of Fig. 8C will be described with reference to the flow chart 4607 in Fig. 46B. The digital

The FM carrier signal 816 is shown again in Fig. 53A for simplicity. In this example, the digital FM carrier signal 816 oscillates at a frequency of about 901 MHz. In Fig. 53B, a digital FM carrier signal 5304 represents a portion of the digital FM carrier signal 816 on an expanded time scale.

Processing begins in step 4 608 where an EM signal is received. This is represented by the digital FM carrier signal 816.

In step 4610, an energy transfer signal is received having an aliasing rate F^r. Figure 53C shows an exemplary energy transfer signal 5306 at approximately the same time scale as Figure 53B. The energy transfer signal 5306 comprises a train of energy transfer pulses 5307 with non-minor pulse widths 5309 tending away from zero in duration. The energy transfer pulses 5307 repeat at the aliasing rate determined or selected as described above. In general, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of the difference frequency F _D i _FF when downconverted to an intermediate frequency signal.

In step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to an intermediate frequency signal F _IF . In Fig. 53D, an influenced digital FM carrier signal 5308 shows effects of energy transfer from the digital FM carrier signal 816 at the aliasing rate F _AR . The influenced digital FM carrier signal 5308 is shown at substantially the same time scale as in Figs. 53B and 53C.

Fig. 53E shows a down-converted FM intermediate frequency signal 5312 generated by the down-conversion process.

: 179 ·: :

The down-converted signal 5312 has portions 5310A correlated with the energy transfer pulses 5307 of Figure 53C and portions 5310B between the energy transfer pulses 5307. The down-converted signal 5312 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

Sections 5310A represent energy transferred from digital FM carrier signal 816 to a storage device while simultaneously driving an output load. Sections 5310A appear when a switching module is closed by energy transfer pulses 5307.

Sections 5310B represent energy stored in a storage device to continue operating a load. Sections 5310B appear when the switching module is opened after energy transfer pulses 5307.

Because a harmonic of the aliasing rate is offset from the frequency of the digital FM carrier signal 816, the energy transfer pulses 5307 "traverse" the digital FM carrier signal 816 at the difference frequency F _DIFF . That is, the energy transfer pulses 5307 appear at different positions in successive cycles of the digital FM carrier signal 816. Therefore, different amounts of energy are captured by the energy transfer pulses 5307 from the digital FM carrier signal 816, as represented by portions 5310, thereby giving the FM intermediate frequency signal 5312 an oscillation frequency F _XF .

In Fig. 53F, a digital FM intermediate frequency signal 5314 represents the FM intermediate frequency signal 5312 in a compressed time scale. In Fig. 53G, an FM intermediate frequency signal 5316 represents a filtered version of the FM

·

intermediate frequency signal 5314. FM intermediate frequency signal 5316 is substantially similar to digital FM carrier signal 816 except that FM intermediate frequency signal 5316 has the intermediate frequency. FM intermediate frequency signal 5316 may then be demodulated by any conventional demodulation technique.

The present invention can output the unfiltered FM intermediate frequency signal 5314, the filtered FM intermediate frequency signal 5316, a partially filtered FM intermediate frequency signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The signals referred to herein represent frequency downconversion in accordance with the invention. For example, FM intermediate frequency signals 5314 in FIG. 53F and 5316 in FIG. 53G represent that FM carrier signal 816 has been successfully downconverted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.2.2

Description of the structure

The operation of the energy transfer system 6302 for the analog FM carrier signal 716 is described below with reference to the flow chart 4607 and the timing diagrams of Figures 52A-G. In step 4608, the energy transfer module 6304 receives the analog FM carrier signal 716. In step 4610, the energy transfer module 6306 receives the energy transfer signal 5206. In step 4612, the energy transfer module 6304 transfers energy from the analog FM carrier signal 716 at the aliasing rate of the energy

is: ·:

transmission signal 5206 to down-convert the analog FM carrier signal 716 into the FM intermediate frequency signal 5212.

The operation of the energy transfer system 6302 for the digital FM carrier signal 816 is described below with reference to the flow chart 4607 and the timing diagrams of Figures 53A-G. In step 4608, the energy transfer module 6304 receives the digital FM carrier signal 816. In step 4610, the energy transfer module 6304 receives the energy transfer signal 5306. In step 4612, the energy transfer module 6304 transfers energy from the digital FM carrier signal 816 at the aliasing rate of the energy transfer signal 5306 to down-convert the digital FM carrier signal 816 to the FM intermediate frequency signal 5212.

Exemplary embodiments of the energy transfer module 6304 are presented in sections 4 and 5 below.

1.2.3 Third exemplary embodiment: phase modulation
1.2.3.1 Description of the functionality

The following illustrates the operation of the exemplary processing of the flow chart 4607 in Fig. 46B for the analog PM carrier signal 916 shown in Fig. 9C and for the digital PM carrier signal 1016 shown in Fig. 10C.

1.2.3.1.1 Analog PM carrier signal

A method for down-converting the analog PM carrier signal 916 of Fig. 9C into an analog PM intermediate frequency signal will be described below with reference to the flow chart 4607 of Fig. 46B. The analog PM carrier signal 916 is shown again in Fig. 54A for simplicity. In

·

In this example, analog PM carrier signal 916 oscillates at a frequency of about 901 MHz. In Figure 54B, analog PM carrier signal 5404 represents a portion of analog PM carrier signal 916 on an expanded time scale.

Processing begins in step 4608 where an EM signal is received. This is represented by the analog PM carrier signal 916.

In step 4610, an energy transfer signal is received having an aliasing rate F _AR . Figure 54C shows an exemplary energy transfer signal 5406 at approximately the same time scale as Figure 54B. The energy transfer signal 5406 comprises a train of energy transfer pulses 5407 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 5407 repeat at the aliasing rate determined or selected as described above. In general, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of the difference frequency F _DIFF when downconverted to an intermediate frequency signal.

In step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to the intermediate frequency signal F _IF . In Figure 54D, an affected analog PM carrier signal 5408 shows effects of energy transfer from the analog PM carrier signal 916 at the aliasing rate F _AR . The affected analog PM carrier signal 5408 is shown at substantially the same time scale as in Figures 54B and 54C.

Fig. 54E shows a down-converted PM intermediate frequency signal 5412 generated by the down-conversion process. The down-converted PM intermediate frequency signal 5412 has portions 5410A associated with the energy transfer pulses

sen 5407 of Fig. 54C, and portions 5410B between the energy transfer pulses 5407. The down-converted signal 5412 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

Sections 5410A represent energy transferred from analog PM carrier signal 916 to a storage device while simultaneously driving an output load. Sections 5410A appear when a switching module is closed by energy transfer pulses 5407.

Sections 5410B represent energy stored in a storage device to continue operating the load. Sections 5410B appear when the switching module is opened after energy transfer pulses 5407.

Because a harmonic of the aliasing rate is offset from the frequency of the analog PM carrier signal 916, the energy transfer pulses 5407 "traverse" the analog PM carrier signal 916 at the difference frequency F _DIFF . That is, the energy transfer pulses 5407 appear at different positions in successive cycles of the analog PM carrier signal 916. Therefore, different amounts of energy are captured by the energy transfer pulses 5407 from the analog PM carrier signal 916, as represented by portions 5410, thereby giving the PM intermediate frequency signal 5412 an oscillation frequency F ₁ F .

In Fig. 54F, an analog PM intermediate frequency signal 5414 represents the PM intermediate frequency signal 5412 in a compressed time scale. In Fig. 54G, a PM intermediate frequency signal 5416 represents a filtered version of the PM intermediate frequency signal 5414. The PM intermediate frequency signal 5416 is substantially similar to the analog PM carrier signal 916.

except that the PM intermediate frequency signal 5416 has an intermediate frequency. The PM intermediate frequency signal 5416 can then be demodulated by any conventional demodulation technique.

The present invention can output the unfiltered PM intermediate frequency signal 5414, the filtered PM intermediate frequency signal 5416, a partially filtered PM intermediate frequency signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The signals referred to herein represent frequency down-conversion in accordance with the present invention. For example, PM intermediate frequency signals 5414 in FIG. 54F and 5416 in FIG. 54G represent that PM carrier signal 916 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.3.1.2 Digital PM carrier signal

A method for down-converting the digital PM carrier signal 1016 of FIG. 10C to a digital PM signal is described below with reference to the timing diagram 4307 in FIG. 46B. The digital PM carrier signal 1016 is shown again in FIG. 55A for simplicity. In this example, the digital PM carrier signal 1016 oscillates at a frequency of about 901 MHz. In FIG. 55B, a digital PM carrier signal 5504 represents a portion of the digital PM carrier signal 1016 on an expanded time scale.

Processing begins in step 4608 where an EM signal is received. This is represented by the PM carrier signal 1016.

In step 4610, an energy transfer signal is received having an aliasing rate F _m . Figure 55C shows an exemplary energy transfer signal 5506 at approximately the same time scale as Figure 55B. The energy transfer signal 5506 comprises a train of energy transfer pulses 5507 with non-minor pulse widths 5509 tending away from zero in duration. The energy transfer pulses 5507 repeat at the aliasing rate determined or selected as described above. In general, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of the difference frequency F _DIFF when downconverted to an intermediate frequency signal.

In step 4612, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal to an intermediate frequency signal F _IF . In Fig. 55D, an influenced digital PM carrier signal 5508 shows effects of energy transfer from the digital PM carrier signal 1016 at the aliasing rate F _AR . The influenced digital PM carrier signal 5508 is shown at substantially the same time scale as in Figs. 55B and 55C.

Figure 55E shows a down-converted PM intermediate frequency signal 5512 produced by the down-conversion process. The down-converted PM intermediate frequency signal 5512 has portions 5510A correlated with the energy transfer pulses 5507 of Figure 55C and portions 5510B between the energy transfer pulses 5507. The down-converted signal 5512 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The portions 5510A represent energy transferred from the digital PM carrier signal 1016 to a storage device

&bull;··· · m

while simultaneously operating an output load. Sections 5510A appear when a switching module is closed by the energy transfer pulses 5507.

Sections 5510B represent energy stored in a storage device to continue operating a load. Sections 5510B appear when the switching module is opened after energy transfer pulses 5507.

Because a harmonic of the aliasing rate is offset from the frequency of the digital PM carrier signal 716, the energy transfer pulses 5507 "traverse" the digital PM carrier signal 1016 at the difference frequency F _DIFF . That is, the energy transfer pulses 5507 appear at different positions in successive cycles of the digital PM carrier signal 1016. Therefore, different amounts of energy are captured by the energy transfer pulses 5507 from the digital PM carrier signal 1016, as represented by portions 5510, thereby giving the PM intermediate frequency signal 5512 an oscillation frequency F _IF .

In FIG. 55F, a digital PM intermediate frequency signal 5514 represents the PM intermediate frequency signal 5512 in a compressed time scale. In FIG. 55G, a PM intermediate frequency signal 5516 represents a filtered version of the PM intermediate frequency signal 5514. The PM intermediate frequency signal 5516 is substantially similar to the digital PM carrier signal 1016, except that the PM intermediate frequency signal 5516 has an intermediate frequency. The PM intermediate frequency signal 5516 may be demodulated by any conventional demodulation technique.

By the present invention, the unfiltered PM intermediate frequency signal 5514, the filtered PM intermediate frequency signal 5516, a partially filtered PM

intermediate frequency signal, a stepped output signal, etc. The choice between these embodiments is generally determined by the design, which depends on the application of the invention.

The signals referred to herein represent frequency down-conversion in accordance with the present invention. For example, PM intermediate frequency signals 5514 in FIG. 55F and 5516 in FIG. 55G represent that PM carrier signal 1016 has been successfully down-converted to an intermediate frequency signal while maintaining sufficient baseband information for adequate reconstruction.

1.2.3.2 Description of the structure

The operation of the power transfer system 6302 for the analog PM carrier signal 916 is described below with reference to the flowchart 4607 and the timing diagrams of Figures 54A-G. In step 4608, the power transfer module 6304 receives the analog PM carrier signal 916. In step 4610, the power transfer module 6304 receives the power transfer signal 5406. In step 4612, the power transfer module 6304 transfers power from the analog PM carrier signal 916 at the aliasing rate of the power transfer signal 5406 to down-convert the analog PM carrier signal 916 to the intermediate frequency PM signal 5412.

The operation of the energy transfer system 6302 for the digital PM carrier signal 1016 is described below with reference to the flow chart 4607 and the timing diagrams of Figures 55A-G. In step 4608, the energy transfer module 6304 receives the digital PM carrier signal 1016. In step 4610, the energy transfer module 6304 receives the energy transfer signal 5506. In step 4612, the energy transfer module 6304 transfers energy from the digital PM carrier signal 1016 to the energy transfer module 6304.

digital PM carrier signal 1016 at the aliasing rate of the power transfer signal 5506 to down-convert the digital PM carrier signal 1016 to the PM intermediate frequency signal 5512.

Example embodiments of the energy transfer module 6304 are presented in sections 4 and 5 below.

1.2.4 Other embodiments

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. Those skilled in the art will appreciate from this description that other embodiments that differ slightly or substantially from those described herein may be implemented within the scope of the present invention. Examples of implementations of the energy transfer module 6304 are presented in Sections 4 and 5 below.

1.3 Implementation examples

Sections 4 and 5 below describe exemplary operational and/or structural implementations related to the methods and/or embodiments described above. These implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the specific implementation examples described herein. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention (e.g., equivalent

implementations, extensions, changes and modifications to the implementations, etc.).

2. Direct downconversion of an EM signal into a demodulated baseband signal by energy transfer from the EM signal

In one embodiment of the invention, an EM baseband signal is downconverted directly to baseband by transferring energy from the EM signal. This embodiment is referred to herein as direct-to-data downconversion and is illustrated in Figure 45B by reference numeral 4516.

This embodiment can be implemented with modulated and unmodulated EM signals. The embodiment is described herein by way of example using the modulated carrier signal F _MC of Fig. 1. In the example, the modulated carrier signal F _MC is directly downconverted to a demodulated baseband signal F _DMB . It will be apparent to those skilled in the art from the present description that the invention can be implemented such that any EM signal, eg a modulated carrier signal and an unmodulated carrier signal, can be downconverted.

In the following sections, methods for directly downconverting the modulated carrier signal F _MC into the demodulated baseband signal F _DM b are described. Exemplary structural embodiments for implementing the methods are also described. However, the invention is not limited to the specific embodiments described below. It will be apparent to those skilled in the art based on the present description that equivalent implementations can be made within the scope of the present invention.

implementations, extensions, changes, modifications, etc. are possible.

The following sections include a high-level description, example embodiments, and implementation examples.

2.1 High-level description

Through this section (including its subsections), a high-level description is provided of the energy transfer from the modulated carrier signal F _MC for directly down-converting the modulated carrier signal F _MC into the demodulated baseband signal F _DMB according to the invention. In particular, an operating method for directly down-converting the modulated carrier signal F _MC into the demodulated baseband signal F _DM b is presented in a high-level description. In addition, a structural implementation of this method is presented in a high-level description. The structural implementation is described herein for illustration purposes only and is not intended to be taken in a limiting sense. In particular, the method described in this section may be realized using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to those skilled in the art based on the present description.

2.1.1 Description of the functionality

Fig. 46C shows a flow chart 4613 illustrating an exemplary method for transferring energy from the module

·

t ·

lated carrier signal F _MC to directly down-convert the modulated carrier signal F _MC to a demodulated baseband signal F _DMB . The exemplary method illustrated in flowchart 4613 is an embodiment of flowchart 4601 of FIG. 46A.

Any and all combinations of modulation techniques are applicable to the present invention. For ease of description, the digital AM carrier signal 616 is used for a high-level description of the invention. The following sections provide detailed timing diagrams and descriptions for examples of AM and PM embodiments. Special considerations are made with respect to frequency modulation (FM), which are discussed separately below in Section III.3. Those skilled in the art will appreciate from the present description that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and an unmodulated carrier signal.

The method illustrated in flow diagram 4613 is illustrated below with a high-level description using the digital AM carrier signal 616 of Figure 6C. The digital AM carrier signal 616 is shown again in Figure 56A for simplicity.

Processing of flowchart 4 613 begins in step 4614 where an EM signal is received. Step 4613 is represented by the digital AM carrier signal 616.

In step 4616, an energy transfer signal is received having an aliasing rate F _n R . Fig. 56B shows an exemplary energy transfer signal 5602 having a sequence of energy transfer pulses 5604 with non-minor pulse widths whose pulse widths 5606

are optimized for energy transfer. The optimized pulse widths 5606 are non-marginal and tend away from zero.

The non-minor pulse widths 5606 may have any width other than the period of the EM signal or a width that is a multiple of the period of the EM signal. For example, the non-minor pulse widths 5606 may be less than the period of the EM signal 616, e.g., their width may be 1/8, 1/4, 1/2, 3/4, etc. of the period of the signal 616. Alternatively, the non-minor pulse widths 5606 may be greater than the period of the signal 616. The width and amplitude of the pulses 5606 may be optimized based on one or more of various criteria, as described in subsequent sections.

The energy transfer pulses 5604 repeat at the aliasing rate or a pulse repetition rate. The aliasing rate is determined according to Eq. (2), which is repeated below for simplicity.

Eq. (2)

If an EM signal is directly downconverted to a baseband signal (ie F _IF = 0), Eq. (2) becomes:

Eq. (8)

To down-convert the AM signal 616 directly into a demodulated baseband signal, the aliasing rate is substantially equal to the frequency of the AM signal 616 or a harmonic or subharmonic thereof. Although the aliasing rate is too low to allow reconstruction of higher frequency grains,

&bull; · «lyj» «··

components of the AM signal 616 (i.e., the carrier frequency), it is high enough to substantially reconstruct the lower frequency modulating baseband signal 310.

In step 4618, energy is transferred from the EM signal at the aliasing rate to directly down-convert the EM signal to the demodulated baseband signal F _D mb . FIG. 56C shows a demodulated baseband signal 5610 produced by the direct down-conversion process. The demodulated baseband signal 5610 is similar to the digital modulating baseband signal 310 of FIG. 3.

56D shows a filtered demodulated baseband signal 5612 that may be generated from the demodulated baseband signal 5610. Thus, the invention may generate a filtered output signal, a partially filtered output signal, or a relatively unfiltered output signal. The choice between a filtered, partially filtered, and unfiltered output signal is generally determined by design, which depends on the application of the invention.

2.1.2 Description of the structure

In one embodiment, the energy transfer system 6302 transfers energy from any type of EM signal, e.g., modulated carrier signals and unmodulated carrier signals, to down-convert the EM signal directly to a demodulated baseband signal. Preferably, the energy transfer system 6302 transfers energy from the EM signal 1304 to down-convert it to a demodulated baseband signal in the manner illustrated in flow diagram 4613. Other structural embodiments for carrying out the steps of the flow diagram 4613 are within the scope of the invention.

diagram 4613. The details of the other structural embodiments will be apparent to those skilled in the art based on the description presented herein.

The operation of the energy transfer system 6302 is described below with reference to the flowchart 4613 and the timing diagrams in Figures 56A-D for the digital AM carrier signal 616. In step 4614, the energy transfer module 6304 receives the digital AM carrier signal 616. In step 4616, the energy transfer module 6304 receives the energy transfer signal 5602. In step 4618, the energy transfer module 6304 transfers energy from the digital AM carrier signal 616 at the aliasing rate to downconvert it directly to the demodulated baseband signal 5610.

Example implementations of the energy transfer module 6302 are presented in sections 4 and 5 below.

2.2 Exemplary embodiments

In this section (and its subsections), various embodiments related to the method(s) and structure(s) described above are presented. These embodiments are described herein for illustrative purposes and are not to be taken in a limiting sense. The invention is not limited to these embodiments. It will be apparent to those skilled in the art based on the present description that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and intended to

suitable to include these alternative embodiments.

The method illustrated in the flowchart 4613 of FIG. 46C for down-converting the EM signal to the demodulated baseband signal F _DM b can be implemented with various types of modulated carrier signals, e.g., AM and PM signals, or any combination thereof. The flowchart 4613 of FIG. 46C is described below with reference to AM and PM signals. The following exemplary descriptions serve to clarify the present invention. The present invention is not limited by or to the following exemplary embodiments.

2.2.1 First exemplary embodiment: amplitude modulation
2.2.1.1 Description of the functionality

The exemplary processing of the flow chart 4613 in Fig. 46C for the analog AM carrier signal 516 shown in Fig. 5C and for the digital AM carrier signal 616 shown in Fig. 6C will be described below.

2.2.1.1.1 Analog AM carrier signal

A method for directly down-converting the analog AM carrier signal 516 of FIG. 5C to a demodulated baseband signal is described below with reference to the flow chart 4613 of FIG. 46C. The analog AM carrier signal 516 is shown again in FIG. 57A for simplicity. In this example, the analog AM carrier signal 516 oscillates at a frequency of about 900 MHz. In FIG. 57B, an analog AM carrier signal 5704 represents a down-

section of the analog AM carrier signal 516 in an expanded time scale.

Processing begins in step 4614 where an EM signal is received. This is represented by the analog AM carrier signal 516.

In step 4616, an energy transfer signal is received having an aliasing rate F _AR . Figure 57C shows an exemplary energy transfer signal 5706 at approximately the same time scale as Figure 57B. The energy transfer signal 5706 comprises a train of energy transfer pulses 5707 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 5707 repeat at the aliasing rate determined or selected as described above. In general, when directly downconverting an EM signal to a demodulated baseband signal, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of the EM signal.

In step 4618, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal directly to the demodulated baseband signal F _DMB . In Fig. 57D, an affected analog AM carrier signal 5708 shows effects of energy transfer from the analog AM carrier signal 516 at the aliasing rate. The affected analog AM carrier signal 5708 is shown at substantially the same time scale as in Figs. 57B and 57C.

Fig. 57E shows a downconverted demodulated baseband signal 5712 produced by the downconversion process. Because a harmonic of the aliasing rate is substantially equal to the frequency of the signal 516, substantially no intermediate frequency signal is produced. The only significant aliased component is the baseband signal. The

.* .* 197:

demodulated baseband signal 5712 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The demodulated baseband signal 5712 has portions 5710A correlated with the energy transfer pulses 5707 in Figure 57C and portions 5710B between the energy transfer pulses 5707. The portions 5710A represent energy transferred from the analog AM signal 516 to a storage device while simultaneously driving an output load. The portions 5710A appear when a switching module is closed by the energy transfer pulses 5707. The portions 5710B represent energy stored in a storage device to continue driving the load. The portions 5710B appear when the switching module is opened after the energy transfer pulses 5707.

In Fig. 57F, a demodulated baseband signal 5716 represents a filtered version of the demodulated baseband signal 5712 in a compressed time scale. The demodulated baseband signal 5716 is substantially similar to the modulating baseband signal 210 and can be further processed using any signal processing technique without further downconversion or demodulation.

The present invention may output the unfiltered demodulated baseband signal 5712, the filtered demodulated baseband signal 5716, a partially filtered demodulated baseband signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The aliasing rate of the energy transfer signal is preferably controlled to ensure that the demodulated baseband signal

&bull;&diams; · · W ·

193

in terms of amplitude output values or polarity.

The drawings referred to herein illustrate a direct down-conversion method in accordance with the present invention. The demodulated baseband signals 5712 in FIG. 57E and 5716 in FIG. 57F illustrate that the analog AM carrier signal 516 has been directly down-converted to a demodulated baseband signal while maintaining sufficient baseband information for adequate reconstruction.

2.2.1.1.2 Digital AM carrier signal

A method for directly downconverting the digital AM carrier signal 616 to a demodulated baseband signal is described below with reference to the flow chart 4613 in FIG. 46C. The digital AM carrier signal 616 is shown again in FIG. 58A for simplicity. In this example, the digital AM carrier signal 616 oscillates at a frequency of about 900 MHz. In FIG. 58B, a digital AM carrier signal 5804 represents a portion of the digital AM carrier signal 616 on an expanded time scale.

Processing begins in step 4614 where an EM signal is received. This is represented by the digital AM carrier signal 616.

In step 4 616, an energy transfer signal is received having an aliasing rate F _AR . Figure 58C shows an exemplary energy transfer signal 5806 at approximately the same time scale as in Figure 58B. The energy transfer signal 5806 comprises a sequence of energy transfer pulses 5807 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 5807 repeat at the aliasing rate,

which is determined or selected as described above. In general, the aliasing rate F _AR in a direct downconversion of an EM signal to a demodulated baseband signal is equal to a harmonic or, more typically, a subharmonic of the EM signal.

In step 4618, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal directly to the demodulated baseband signal F _DMB . In Fig. 58D, an affected digital AM carrier signal 5808 shows effects of energy transfer from the digital AM carrier signal 616 at the aliasing rate Fp^. The affected digital AM carrier signal 5808 is shown at substantially the same time scale as in Figs. 58B and 58C.

Figure 58E shows a demodulated baseband signal 5812 produced by the downconversion process. Because a harmonic of the aliasing rate is substantially equal to the frequency of signal 616, essentially no intermediate frequency signal is produced. The only significant aliased component is the baseband signal. The demodulated baseband signal 5812 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The demodulated baseband signal 5812 has portions 5810A correlated with the energy transfer pulses 5807 in Figure 58C and portions 5810B between the energy transfer pulses 5807. The portions 5810A represent energy transferred from the digital AM signal 616 to a storage device while simultaneously driving an output load. The portions 5810A appear when a switching module is closed by the energy transfer pulses 5807. The portions 5810B represent energy stored in a storage device.

is stored to continue operating the load. Sections 5810B appear when the switch module is opened after the energy transfer pulses 5807.

In Figure 58F, a demodulated baseband signal 5816 represents a filtered version of the demodulated baseband signal 5812 in a compressed time scale. The demodulated baseband signal 5816 is substantially similar to the modulating baseband signal 310 and may be further processed using any signal processing technique without further down-conversion or demodulation.

The present invention may output the unfiltered demodulated baseband signal 5812, the filtered demodulated baseband signal 5816, a partially filtered demodulated baseband signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted baseband signal in terms of amplitude output values or polarity.

The drawings referred to herein illustrate a direct downconversion method in accordance with the present invention. The demodulated baseband signals 5812 in FIG. 58E and 5816 in FIG. 58F illustrate that the digital AM carrier signal 616 has been directly downconverted to a demodulated baseband signal while maintaining sufficient baseband information for adequate reconstruction.

.· .· 2&ogr;&igr;: :

2.2.1.2 Description of the structure

In one embodiment, the energy transfer system 6304 preferably transfers energy from the EM signal to down-convert it directly to a demodulated baseband signal in the manner illustrated in flowchart 4613. Other structural embodiments for carrying out the steps of flowchart 4613 are possible within the scope of the invention. The details of the other structural embodiments will be apparent to those skilled in the art based on the description presented herein.

The operation of the power transfer system 6302 is described below with reference to the flowchart 4613 and the timing diagrams in Figures 57A-F for the analog AM carrier signal 516. In step 4612, the power transfer module 6404 receives the analog AM carrier signal 516. In step 4614, the power transfer module 6404 receives the power transfer signal 5706. In step 4618, the power transfer module 6404 transfers power from the analog AM carrier signal 516 at the aliasing rate of the power transfer signal 5706 to down-convert the analog AM carrier signal 516 directly to the demodulated baseband signals 5712 or 5716.

The operation of the energy transfer system 6402 is described below with reference to the flow chart 4613 and the timing diagrams in Figures 58A-F for the digital AM carrier signal 616. In step 4614, the energy transfer module 6404 receives the digital AM carrier signal 616. In step 4616, the energy transfer module 6404 receives the energy transfer signal 5806. In step 4618, the energy transfer module 6404 transfers energy from the digital AM carrier signal 616 at the aliasing rate of the energy transfer signal 5806 to generate the digital

AM carrier signal 616 directly into the demodulated baseband signals 5812 or 5816.

Implementation examples of the energy transfer module 6302 are presented in sections 4 and 5 below.

2.2.2 Second exemplary embodiment: phase modulation

2.2.2.1

Description of how it works

The following illustrates the operation of the exemplary processing of the flow chart 4613 in Fig. 46C for the analog PM carrier signal 916 shown in Fig. 9C and for the digital PM carrier signal 1016 shown in Fig. 10C.

2.2.2.1.1 Analog PM carrier signal

A method for directly downconverting the analog PM carrier signal 916 to a demodulated baseband signal is described below with reference to the timing diagram 4613 of FIG. 46C. The analog PM carrier signal 916 is shown again in FIG. 59A for simplicity. In this example, the analog PM carrier signal 916 oscillates at a frequency of about 900 MHz. In FIG. 59B, an analog PM carrier signal 5904 represents a portion of the analog PM carrier signal 916 on an expanded time scale.

Processing begins in step 4614 where an EM signal is received. This is represented by the analog PM carrier signal 916.

In step 4616, an energy transfer signal is received having an aliasing rate F^r. Fig. 59C shows an exemplary energy transfer signal 5906 at approximately the same time scale as in Fig. 59B. The energy

· ·

·

Transfer signal 5906 comprises a train of energy transfer pulses 5907 with non-minor pulse widths tending away from zero in duration. Energy transfer pulses 5907 repeat at the aliasing rate determined or selected as described above. In general, the aliasing rate F _m is substantially equal to a harmonic, or more typically a subharmonic, of the difference frequency F _DI ff when directly downconverted to a demodulated baseband signal.

In step 4618, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal directly to the demodulated baseband signal F _DMB . In Fig. 59D, an affected analog PM carrier signal 5908 shows effects of energy transfer from the analog PM carrier signal 916 at the aliasing rate F _AR . The affected analog PM carrier signal 5908 is shown at substantially the same time scale as in Figs. 59B and 59C.

Figure 59E shows a downconverted demodulated baseband signal 5912 produced by the downconversion process. Because a harmonic of the aliasing rate is substantially equal to the frequency of signal 516, essentially no intermediate frequency signal is produced. The only significant aliased component is the baseband signal. The demodulated baseband signal 5912 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The demodulated baseband signal 5912 has portions 5910A correlated with the energy transfer pulses 5907 in Fig. 59C and portions 5910B between the energy transfer pulses 5907. The portions 5910A represent energy transferred from the analog PM carrier signal 916 to a storage device while

simultaneously driving an output load. Sections 5910A appear when a switching module is closed by energy transfer pulses 5907. Sections 5910B represent energy stored in a storage device to continue driving the load. Sections 5910B appear when the switching module is opened after energy transfer pulses 5907.

In Figure 59F, a demodulated baseband signal 5916 represents a filtered version of the demodulated baseband signal 5912 in a compressed time scale. The demodulated baseband signal 5916 is substantially similar to the modulating baseband signal 210 and may be further processed using any signal processing technique without further down-conversion or demodulation.

The present invention may output the unfiltered demodulated baseband signal 5912, the filtered demodulated baseband signal 5916, a partially filtered demodulated baseband signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted baseband signal in terms of amplitude output values or polarity.

The drawings referred to herein illustrate a direct down-conversion method according to the invention. For example, demodulated baseband signals 5912 in Fig. 59E and 5916 in Fig. 59F show that analog AM carrier signal 916 has been successfully down-converted to a demodulated baseband signal, for an

sufficient reconstruction maintains sufficient baseband information.

2.2.2.1.2 Digital PM carrier signal

A method for down-converting the digital PM carrier signal 1016 of FIG. 6C to a demodulated baseband signal is described below with reference to the timing diagram 4613 in FIG. 46C. The digital PM carrier signal 1016 is shown again in FIG. 60A for simplicity. In this example, the digital PM carrier signal 1016 oscillates at a frequency of about 900 MHz. In FIG. 60B, a digital PM carrier signal 6004 represents a portion of the digital PM carrier signal 1016 on an expanded time scale.

Processing begins in step 4614 where an EM signal is received. This is represented by the PM carrier signal 1016.

In step 4616, an energy transfer signal is received having an aliasing rate F AR . FIG. 60C shows an exemplary energy transfer signal 6006 at approximately the same time scale as FIG. 60B. The energy transfer signal 6006 comprises a train of energy transfer pulses 6007 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 6007 repeat at the aliasing rate determined or selected as described above. In general, when downconverting an EM signal to a demodulated baseband signal, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of the frequency of the EM signal.

In step 4618, energy is transferred from the EM signal at the aliasing rate to down-convert the EM signal directly to the demodulated baseband signal F _DMB . In Fig. 60D, an influenced digital PM carrier signal 6008 shows effects of energy transfer from the digital PM carrier signal 1016 at the aliasing rate F _AR . The influenced digital PM carrier signal 6008 is shown at substantially the same time scale as in Figs. 60B and 60C.

Figure 60E shows a downconverted demodulated baseband signal 6012 produced by the downconversion process. Because a harmonic of the aliasing rate is substantially equal to the frequency of signal 1016, essentially no intermediate frequency signal is produced. The only significant aliased component is the baseband signal. The demodulated baseband signal 6012 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The demodulated baseband signal 6012 has portions 6010A correlated with the energy transfer pulses 6007 in Figure 60C and portions 6010B between the energy transfer pulses 6007. The portions 6010A represent energy transferred from the digital PM carrier signal 1016 to a storage device while simultaneously driving an output load. The portions 6010A appear when a switching module is closed by the energy transfer pulses 6007. The portions 6010B represent energy stored in a storage device to continue driving the load. The portions 6010B appear when the switching module is opened after the energy transfer pulses 6007.

In Fig. 6OF, a demodulated baseband signal 6016 represents a filtered version of the demodulated baseband signal 6012 in a compressed time scale. The demo-

The modulated baseband signal 6016 is substantially similar to the modulating baseband signal 310 and can be further processed using any signal processing technique without further down-conversion or demodulation.

The present invention may output the unfiltered demodulated baseband signal 6012, the filtered demodulated baseband signal 6016, a partially filtered demodulated baseband signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention.

The aliasing rate of the power transfer signal is preferably controlled to optimize the down-converted signal in terms of amplitude output values and polarity.

The drawings referred to herein illustrate a direct down-conversion method in accordance with the present invention. For example, demodulated baseband signals 6012 in FIG. 60E and 6016 in FIG. 60F illustrate that digital AM carrier signal 1016 has been successfully down-converted to a demodulated baseband signal while maintaining sufficient baseband information for adequate reconstruction.

2.2.2.2

Description of the structure

In one embodiment, the energy transfer system 6302 preferably transfers energy from an EM signal to down-convert it directly to a demodulated baseband signal in the manner illustrated in the flow diagram 4613. Other structural embodiments for carrying out the steps of down-converting are within the scope of the invention.

.· 208

flow diagram 1413. The details of the other structural embodiments will be apparent to those skilled in the art based on the present description.

The operation of the power transfer system 6302 is described below with reference to the flowchart 4613 and the timing diagrams in Figures 59A-F for the analog PM carrier signal 916. In step 4614, the power transfer module 6304 receives the analog PM carrier signal 916. In step 4616, the power transfer module 6304 receives the power transfer signal 5906. In step 4618, the power transfer module 6304 transfers power from the analog PM carrier signal 916 at the aliasing rate of the power transfer signal 5906 to down-convert the analog PM carrier signal 916 directly to the demodulated baseband signals 5912 or 5916.

The operation of the energy transfer system 6302 is described below with reference to the flowchart 4613 and the timing diagrams in Figures 60A-F for the digital PM carrier signal 1016. In step 4614, the energy transfer module 6404 receives the digital PM carrier signal 1016. In step 4616, the energy transfer module 6404 receives the energy transfer signal 6006. In step 4618, the energy transfer module 6404 transfers energy from the digital PM carrier signal 1016 at the aliasing rate of the energy transfer signal 6006 to down-convert the digital PM carrier signal 1016 directly to the demodulated baseband signals 6012 or 5016.

Implementation examples of the energy transfer module 6302 are presented in sections 4 and 5 below.

2.2.3 Other embodiments

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. It will be apparent to those skilled in the art from this description that other embodiments that differ slightly or substantially from those described herein can be implemented within the scope of the present invention.

2.3 Implementation examples

Sections 4 and 5 below describe exemplary functional and/or structural implementations related to the methods and/or embodiments described above. These implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the implementation examples described herein. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention (e.g., equivalent implementations, extensions, changes and modifications of the implementations, etc.).

3. Modulation implementation

In one embodiment, an FM carrier signal F _FM c is down-converted to a non-FM signal F(N _0n -FM) by transferring energy from the FM carrier signal F _FMC at an aliasing rate.

.* 2-io

delt. This embodiment is shown in Fig. 45B by the reference numeral 4518.

In an exemplary embodiment, the FM carrier signal F _mc is down-converted to a phase modulated (PM) signal F _PM . In another exemplary embodiment, the FM carrier signal F _FMC is down-converted to an amplitude modulated (AM) signal Fam . The down-converted signal may be demodulated by any conventional demodulation technique to obtain a demodulated baseband signal F _DMB .

The invention is implementable with any type of FM signal. Example embodiments for down-converting a frequency shift keying (FSK) signal to a non-FSK signal are described below. FSK signals are a subset of FM signals where an FM signal switches between two or more frequencies. Frequency shift keying (FSK) is typically used for digital modulating baseband signals, e.g., for the digital modulating baseband signal 310 in FIG. 3. For example, in FIG. 8, the digital FM signal 816 is an FSK signal that switches between an upper frequency and a lower frequency, corresponding to amplitude changes in the digital modulating baseband signal 310. The FSK signal 816 is used in example embodiments below.

In a first exemplary embodiment, energy is transmitted from the FSK signal 816 at an aliasing rate based on a center frequency between the upper and lower frequencies of the FSK signal 816. When the aliasing rate is based on the center frequency, the FSK signal 816 is down-converted to a phase shift keying (PSK) signal. PSK signals are a subset of phase modulated signals where a PM signal is modulated between two or more

&diams;»

Phase shift keying (PSK) is typically used for digital modulating baseband signals. For example, in Figure 10, the digital PM signal 1016 is a PSK signal that switches between two phases. The PSK signal 1016 can be demodulated by any conventional PSK demodulation technique.

In a second exemplary embodiment, energy is transmitted from the FSK signal 816 at an aliasing rate based on either the upper frequency or the lower frequency of the FSK signal 816. If the aliasing rate is based on the upper frequency or the lower frequency of the FSK signal 816, the FSK signal 816 is down-converted to an amplitude shift keying (ASK) signal. ASK signals are a subset of amplitude modulated signals where an AM signal switches between two or more amplitudes. Amplitude shift keying (ASK) is typically used for digital modulating baseband signals. For example, in Figure 6, the digital AM signal 616 is an ASK signal that switches between a first amplitude and a second amplitude. The ASK signal 616 may be demodulated by any conventional ASK demodulation technique.

In the following sections, methods for transferring energy from an FM carrier signal F _FMC to down-convert it to a non-FM signal F _(N0N - _FM) are described. In addition, exemplary structural embodiments for implementing the methods are described. The invention is not limited to the specific embodiments described below. Those skilled in the art will appreciate, based on the present description, that equivalent implementations, extensions, changes, modifications, etc. are possible within the scope of the present invention.

The following sections include a high-level description, exemplary embodiments, and implementation examples.

3.1 High-level description

Through this section (including its subsections), a high-level description is provided of the energy transfer from the FM carrier signal F fm for down-converting the FM carrier signal to a non-FM signal F(NON-FM) according to the invention. In particular, an operating method for down-converting the FM carrier signal F _MC to the non-FM signal F _(N on-fm) is presented ⁱⁿ a high-level description. In addition, a structural implementation of this method is presented in a high-level description. This structural implementation is described herein for illustration and is not intended to be limiting. In particular, the method described in this section may be realized using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to those skilled in the art based on the present description.

3.1.1 Description of the functionality

46D shows a flow diagram 4619 illustrating an exemplary method for down-converting an FM carrier signal Ffmc to a non-FM signal F _(NO n-fm). The exemplary method illustrated in flow diagram 4619 is an embodiment of the flow diagram 4601 of FIG. 46A.

Any and all forms of frequency modulation techniques are applicable to the present invention. For ease of description, the digital FM carrier (FSK) signal 816 is used to illustrate the operation of the invention in a high-level description. Detailed timing diagrams and descriptions for the FSK signal 816 are provided in the following sections. Those skilled in the art will appreciate from this description that the invention can be implemented to down-convert any type of FM signal.

The process illustrated in flow diagram 4619 is illustrated below with a high-level description of down-converting the FSK signal 816 of Figure 8C to a PSK signal. The FSK signal 816 is shown again in Figure 84C for simplicity.

Processing of flowchart 4619 begins in step 4620 where an FM signal is received. This is represented by FSK signal 816. FSK signal 816 switches between a first frequency 8410 and a second frequency 8412. First frequency 8410 may be higher or lower than second frequency 8412. In an exemplary embodiment, first frequency 8410 is about 899 MHz and second frequency 8412 is about 901 MHz.

In step 4622, an energy transfer signal is received having an aliasing rate F _AR . Figure 84B shows an example of an energy transfer signal 8402 comprising a train of energy transfer pulses 8403 with non-minor pulse widths 8405 whose duration tends away from zero.

The energy transfer pulses 8403 repeat

with the aliasing rate F _AR determined or selected as described above. If an FM

When the carrier signal Ffmc is down-converted to a non-FM signal F( _NO n-fm), the aliasing rate generally equals a harmonic, or more typically a subharmonic, of a frequency contained in the FM signal. In this overview of an exemplary embodiment in which the FSK signal 816 is to be down-converted to a PSK signal, the aliasing rate generally equals a harmonic, or more typically a subharmonic, of the center frequency between the first frequency 8410 and the second frequency 8412. For the present example, the center frequency is approximately 900 MHz.

In step 4 624, energy is transferred from the FM signal F _FMC at the aliasing rate to down-convert the FM carrier signal F _FMC to the non-FM signal F( _N0N -fm). Fig. 84C shows a PSK signal 8404 generated by the modulation conversion process.

When the second frequency 8412 is subsampled, the PSK signal 8404 has a frequency of approximately 1 MHz and is used as a phase reference. When the first frequency 8410 is subsampled, the PSK signal 8404 has a frequency of 1 MHz and is 180 degrees out of phase with respect to the phase reference.

Fig. 84D shows a PSK signal 8406 which is a filtered version of the PSK signal 8404. Thus, the invention may produce a filtered or partially filtered output signal or a relatively unfiltered stepped output signal. The choice between a filtered, partially filtered and unfiltered output signal is generally determined by design, which depends on the application of the invention.

The aliasing rate of the energy transfer signal is preferably controlled to bias the down-converted signal

in terms of amplitude output values or polarity.

Detailed exemplary embodiments for down-converting an FSK signal to a PSK signal and for down-converting an FSK signal to an ASK signal are described below.

3.1.2 Description of the structure

63 shows the energy transfer system 6302 according to an embodiment of the invention. The energy transfer system 6302 includes the energy transfer module 6304. The energy transfer system 6302 is an exemplary embodiment of the general aliasing system 1302 of FIG. 13.

In the molecular conversion embodiment, the EM signal 1304 is an FM carrier signal F _FMC , and the energy transfer module 6304 transfers energy from the FM carrier signal at a frequency corresponding to a harmonic, or more typically a subharmonic, of a frequency within the FM frequency band. Preferably, the energy transfer module 6304 transfers energy from the FM carrier signal FpMc to downconvert it to a non-FM signal F _(NO n-fm) in a manner illustrated in flow diagram 4619. Other structural embodiments for carrying out the steps of flow diagram 4619 are possible within the scope of the invention. The details of the other structural embodiments will be apparent to those skilled in the art based on the description presented herein.

The operation or functioning of the energy transfer system 6302 is described below with reference to the flow chart 4619 and the timing diagrams in Figures 84A-D.

In step 4620, the energy transfer module 6304 receives the FSK signal 816. In step 4622, the energy transfer module 6304 receives the energy transfer signal 8402. In step 4624, the energy transfer module 6304 transfers energy from the FSK signal 816 at the aliasing rate of the energy transfer signal 8402 to down-convert the FSK signal 816 to the PSK signal 8404 or 8406.

Implementation examples of the energy transfer module 6302 are presented in sections 4 and 5 below.

3.2 Exemplary embodiments

In this section (and its subsections), various embodiments related to the method(s) and structure(s) described above are presented. These embodiments are described herein for illustrative purposes and are not to be taken in a limiting sense. The invention is not limited to these embodiments. It will be apparent to those skilled in the art based on the present description that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and adapted to include these alternative embodiments.

The method for down-converting an FM carrier signal F _FMC into a non-FM signal F _(N0N _ _FM ) shown in the flow chart 4619 of Fig. 46D can be implemented with any type of FM carrier signal, e.g. with FSK signals. The flow chart 4619 for down-converting an FSK signal into a PSK signal and for down-converting a FSK signal into a PSK signal is shown below.

Converting an FSK signal into an ASK signal is described in detail. The following exemplary descriptions serve to clarify the present invention. The present invention is not limited by or to the following exemplary embodiments.

3.2.1 First exemplary embodiment: Downconversion of an FM signal into a PM signal

3.2.1.1 Description of the functionality

A method for down-converting the FSK signal 816 shown in Fig. 8C into a PSK signal will be described below with reference to the flowchart 4619 of Fig. 46D.

The FSK signal 816 is shown again in Figure 61A for simplicity. The FSK signal 816 switches between a first frequency 6106 and a second frequency 6108. In the exemplary embodiment, the first frequency 6106 is lower than the second frequency 6108. In an alternate embodiment, the first frequency 6106 is higher than the second frequency 6108. In this example, the first frequency 6106 is about 899 MHz and the second frequency 6108 is about 901 MHz.

Figure 61B shows an FSK signal portion 6104 representing a portion of the FSK signal 816 in an expanded time scale.

Processing begins in step 4 620 where an FM signal is received. This is represented by the FSK signal 816.

In step 4622, an energy transfer signal is received having an aliasing rate F _A r. Fig. 61C shows an exemplary energy transfer signal 6107 in approx.

about the same time scale as in Fig. 61B. The energy transfer signal 6107 comprises a train of energy transfer pulses 6109 with non-minor pulse widths tending away from zero in duration. The energy transfer pulses 6109 repeat at the aliasing rate F _AR determined or selected as described above. In general, when downconverting an FM signal to a non-FM signal, the aliasing rate is substantially equal to a harmonic, or more typically a subharmonic, of a frequency contained in the FM signal.

In this example, where an FSK signal is downconverted to a PSK signal, the aliasing rate of a harmonic, or more typically a subharmonic, of the center frequency is substantially the same between frequencies 6106 and 6108. In this example, where the first frequency 6106 is 899 MHz and the second frequency 6108 is 901 MHz, the center frequency is approximately 900 MHz. Suitable aliasing rates are 1.8 GHz, 900 MHz, 450 MHz, etc.

In step 4624, energy is transferred from the FM signal at the aliasing rate to down-convert it to the non-FM signal F(NON-FM). In Fig. 61D, a biased FSK signal 6118 shows effects of energy transfer from the FSK signal 816 at the aliasing rate Far. The biased FSK signal 6118 is shown at substantially the same time scale as in Figs. 61B and 61C.

Figure 61E shows a PSK signal 6112 generated by the modulation conversion process. The PSK signal 6112 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The PSK signal 6112 has portions 6110A that correlate with the energy transfer pulses 6107 in Fig. 61C

The PSK signal 6112 also includes portions 6110B between the energy transfer pulses 6109. The portions 6110A represent energy transferred from the FSK signal 816 to a storage device while simultaneously operating an output load. The portions 6110A appear when a switching module is closed by the energy transfer pulses 6109. The portions 6110B represent energy stored in a storage device to continue operating the load. The portions 6110B appear when the switching module is opened after the energy transfer pulses 6107.

In Fig. 61F, a PSK signal 6114 represents a filtered version of the PSK signal 6112 in a compressed time scale. The present invention may output the unfiltered demodulated baseband signal 6112, the filtered demodulated baseband signal 6114, a partially filtered demodulated baseband signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention. The PSK signals 6112 and 6114 may be demodulated by a conventional demodulation technique.

The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal in terms of amplitude output values or polarity.

The drawings referred to herein illustrate a modulation conversion method according to the invention. For example, the PSK signals 6112 shown in Fig. 61E and the PSK signals 6114 shown in Fig. 61F illustrate that the FSK signal 816 has been successfully down-converted to a PSK signal, providing sufficient

t · m
·

reconstruction while maintaining sufficient baseband information.

3.2.1.2 Description of the structure

The operation of the energy transfer system 1602 for down-converting the FSK signal 816 to a PSK signal is described below with reference to the flowchart 4619 and the timing diagrams of Figures 61A-E. In step 4620, the energy transfer module 1606 receives the FSK signal 816 (Figure 61A). In step 4622, the energy transfer module 1606 receives the energy transfer signal 6107 (Figure 61C). In step 4624, the energy transfer module 1606 transfers energy from the FSK signal 816 at the aliasing rate of the energy transfer signal 6107 to down-convert the FSK signal 816 to the PSK signal 6112 of Figure 61E or the PSK signal 6114 of Figure 61F.

3.2.2 Second exemplary embodiment: Down-conversion of a
FM signal into an AM signal

3.2.2.1 Description of the functionality

A method for down-converting the FSK signal 816 of FIG. 8C into an ASK signal will be described below with reference to the flowchart 4619 of FIG. 46D.

The FSK signal 816 is shown again in Fig. 62A for simplicity. The FSK signal 816 switches between a first frequency 6206 and a second frequency 6208. In the exemplary embodiment, the first frequency 6206 is lower than the second frequency 6208. In an alternative embodiment, the first frequency 6206 is higher than the second frequency 6208. In this example, the

first frequency 6206 about 899 MHz and the second frequency 6208 about 901 MHz.

Figure 62B shows an FSK signal portion 6204 showing a portion of the FSK signal 816 in an expanded time scale.

Processing begins in step 4620 where an FM signal is received. This is represented by the FSK signal 816.

In step 4622, an energy transfer signal is received having an aliasing rate F _AR . Figure 62C shows an exemplary energy transfer signal 6207 at approximately the same time scale as Figure 62B. The energy transfer signal 6207 comprises a train of energy transfer pulses 6209 with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses 6209 repeat at the aliasing rate F _AR determined or selected as described above. In general, when downconverting an FM signal to a non-FM signal, the aliasing rate F _AR is substantially equal to a harmonic, or more typically a subharmonic, of a frequency contained in the FM signal.

In this example, where an FSK signal is down-converted to an ASK signal, the aliasing rate of a harmonic or, more typically, a subharmonic of the first frequency 6206 or the second frequency 6208 is substantially the same. In this example, if the first frequency 6206 is 899 MHz and the second frequency 6208 is 901 MHz, the aliasing rate of a harmonic or subharmonic of 899 MHz or 901 MHz may be substantially the same.

In step 4624, energy is transferred from the FM signal at the aliasing rate to convert it into the non-FM signal

· · ■

·

FM). In Fig. 62D, an affected FSK signal 6218 shows effects of energy transfer from the FSK signal 816 at the aliasing rate F _AR . The affected FSK signal 6218 is shown on substantially the same time scale as in Figs. 62B and 62C.

Figure 62E shows an ASK signal 6212 generated by the modulation conversion process. The ASK signal 6212 is shown with an arbitrary load impedance. Load impedance optimizations are discussed below in Section 5.

The ASK signal 6212 has portions 6210A that are correlated with the energy transfer pulses 6209 in Figure 62C. The ASK signal 6212 also has portions 6210B between the energy transfer pulses 6209. The portions 6210A represent energy transferred from the FSK signal 816 to a storage device while simultaneously operating an output load. The portions 6210A appear when a switching module is closed by the energy transfer pulses 6207. The portions 6210B represent energy stored in a storage device to continue operating the load. The portions 6210B appear when the switching module is opened after the energy transfer pulses 6207.

In Fig. 62F, an ASK signal 6214 represents a filtered version of the ASK signal 6212 in a compressed time scale. The present invention may output the unfiltered demodulated baseband signal 6212, the filtered demodulated baseband signal 6214, a partially filtered demodulated baseband signal, a stepped output signal, etc. The choice between these embodiments is generally determined by design, which depends on the application of the invention. The

ASK signals 6212 and 6214 can be derodulated by a conventional demodulation technique.

The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal in terms of amplitude output values or polarity.

The drawings referred to herein illustrate a modulation conversion method according to the present invention. For example, ASK signal 6212 illustrated in FIG. 62E and ASK signal 6214 illustrated in FIG. 62F illustrate that FSK signal 816 has been successfully downconverted to an ASK signal while maintaining sufficient baseband information for adequate reconstruction.

3.2.2.2 Description of the structure

The operation of the energy transfer system 1602 for down-converting the FSK signal 816 to an ASK signal is described below with reference to the flow chart 4619 and the timing diagrams of Figures 62A-F. In step 4620, the energy transfer module 6304 receives the FSK signal 816 (Figure 62A). In step 4622, the energy transfer module 6304 receives the energy transfer signal 6207 (Figure 62C). In step 4624, the energy transfer module 6304 transfers energy from the FSK signal 818 at the aliasing rate of the energy transfer signal 6207 to down-convert the FSK signal 816 to the ASK signal 6212 of Figure 62E or the ASK signal 6214 of Figure 62F.

3.2.3 Other exemplary embodiments

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. It will be apparent to those skilled in the art from this description that other embodiments that differ slightly or substantially from those described herein can be implemented within the scope of the present invention.

Sections 4 and 5 below provide implementation examples of the energy transfer module 6302.

3.3 Implementation examples

Sections 4 and 5 below describe exemplary functional and/or structural implementations related to the methods and/or embodiments described above. These implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the specific implementation examples described herein. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention (e.g., equivalent implementations, extensions, changes and modifications of the implementations, etc.).

4. Implementation examples

This section (and its subsections) describes exemplary functional and/or structural implementations related to the methods and/or embodiments described in the preceding subsections. These implementations are for illustrative purposes only and are not to be taken in a limiting sense. The invention is not limited to the implementation examples described herein. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention (e.g., equivalent implementations, extensions, changes and modifications of the implementations, etc.).

63 shows an energy transfer system 6302 that is an exemplary embodiment of the general aliasing system 1302 of FIG. 13. The energy transfer system 6302 includes an energy transfer module 6304 that receives the EM signal 1304 and an energy transfer signal 6306. The energy transfer signal 6306 includes a train of energy transfer pulses with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses repeat at an aliasing rate F _AR .

The energy transfer module 6304 transfers energy from the EM signal 1304 at the aliasing rate of the energy transfer signal 6306, as described in the previous sections with reference to the flow charts 4601 in Fig. 46A, 4607 in Fig. 46B, 4613 in Fig. 46C, and 4619 in Fig. 46D. The energy transfer module 6304 outputs a down-converted

enhanced signal 1308B having non-trivial amounts of energy carried by EM signal 1304.

64A shows an exemplary gated transmission system 6402 that is an example of the power transmission system 6302. The gated transmission system 6402 includes a gated transmission module 6404, which is described below.

64B shows an exemplary inverted gated transfer system 6406, which is an alternate example of the power transfer system 6302. The inverted gated transfer system 6406 includes an inverted gated transfer module 6408, which is described below.

4.1 The energy transmission system as a gate-controlled transmission system

64A shows the exemplary gated transfer system 6402, which is an exemplary implementation of the energy transfer system 6302. The gated transfer system 6402 includes the gated transfer module 6404 that receives the EM signal 1304 and the energy transfer signal 6306. The energy transfer signal 6306 includes a train of energy transfer pulses with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses repeat at an aliasing rate F _AR .

The gated transfer module 6404 transfers energy from the EM signal 304 at the aliasing rate of the energy transfer signal 6306 as described in the previous sections with reference to the flow charts 4601 in Fig. 46A, 4607 in Fig. 46B, 4613 in Fig. 46C and 4619 in Fig. 46D. The gated transfer module 6404 outputs the downconverted signal 308B which is non-

small amounts of energy transmitted by EM signal 1304.

4.1.1 The gate-controlled transmission system as a switching module and storage module

65 shows an exemplary embodiment of the gated transmission module 6404 including a switching module 6502 and a memory module 6506. The switching module 6502 and the memory module 6506 preferably transfer energy from the EM signal 1304 to down-convert it in any of the ways illustrated in timing diagrams 4601 of FIG. 46A, 4607 of FIG. 46B, 4613 of FIG. 46C, and 4619 of FIG. 46D.

The operation of the switching module 6502 and the memory module 6506 for down-converting the EM signal 1304 to an intermediate frequency signal is described below by way of example with reference to the flow chart 607 and the exemplary timing diagrams in Figures 83A-F.

In step 4608, the switching module 6502 receives the EM signal 1304 (FIG. 83A). In step 4610, the switching module 6502 receives the energy transfer signal 6306 (FIG. 83C). In step 4612, the switching module 6502 and the storage module 6506 cooperate to transfer energy from the EM signal 1304 and down-convert it to an intermediate frequency signal. Specifically, in step 4612, the switching module 6502 closes during each energy transfer pulse to couple the EM signal 1304 to the storage module 6506. In one embodiment, the storage module 6502 closes on rising edges of the energy transfer pulses. In an alternative embodiment, the storage module 6502 closes on falling edges of the energy transfer pulses. While the EM signal 1304 is coupled to the memory module 6506,

non-trivial amounts of energy are transferred from the EM signal 1304 to the storage module 6506. Figure 83B shows the EM signal 1304 after energy has been transferred therefrom. Figure 83D shows the transferred energy stored in the storage module 6506. The storage module 6506 outputs the transferred energy as the down-converted signal 1308B. The storage module 6506 may output the down-converted signal 1308B as an unfiltered signal, e.g., as the signal shown in Figure 83E, or as a filtered down-converted signal (Figure 83F).

4.1.2 The gate-controlled transmission system as a sequential switching module

67A shows an exemplary embodiment of the gated transmission module 6404 with a sequence switching module 6702 and a memory module 6716. The sequence switching module 6702 and the memory module 6716 preferably transfer energy from the EM signal 1304 to down-convert it in any of the manners illustrated in timing diagrams 4601 of FIG. 46A, 4607 of FIG. 46B, 4613 of FIG. 46C, and 4619 of FIG. 46D.

In Fig. 67A, the sequential switching module 6702 includes a normally open switch 6704 and a normally closed switch 6706. The normally open switch 6704 is controlled by the energy transfer signal 6306. The normally closed switch 6706 is controlled by a decoupling or disconnecting signal 6712. In one embodiment, the decoupling or disconnecting signal 6712 is generated from the energy transfer signal 6306. Alternatively, the energy transfer signal 6306 is generated from the decoupling or disconnecting signal 6712. Alternatively, the decoupling or disconnecting signal 6712 is generated independently of the energy transfer signal 6306. The sequential switching module 6702 decouples or

essentially separates an input 6708 from an output 6710.

Figure 67B shows an exemplary timing diagram of the energy transfer signal 6306 controlling the normally open switch 6704. Figure 67C shows an exemplary timing diagram of the decoupling or disconnection signal 6712 controlling the normally closed switch 6706. The operation of the sequence switching module 6702 is described below with reference to the exemplary timing diagrams of Figures 67B and 67C.

Before a time t0, the normally open switch 6704 and the normally closed switch 6706 are set to their normal states.

At time tO, the decoupling or disconnecting signal 6712 in Fig. 67C opens the normally closed switch 6706. Then, immediately after time t0, the normally open switch 6704 and the normally closed switch 6706 are open and the input 6708 is decoupled or disconnected from the output 6710.

At a time ti, the energy transfer signal 6306 in Fig. 67B closes the normally open switch 6704 for the non-trivial duration of a pulse. This couples the EM signal 1304 to the holding module 6716.

Before a time t2, the energy transfer signal 6306 in Fig. 67B opens the normally open scraper 6704. This decouples the EM signal 1304 from the holding module 6716.

At time t2, the decoupling or disconnecting signal 6712 in Fig. 67C closes the normally closed switch 6706. This couples the holding module 6716 to the output 6710.

The memory module 6716 is similar to the memory module 6506 of Fig. 65. The gated sequence switching

&bull; . i ?hriw . · · ·

f1

Transmission system 6701 downconverts EM signal 1304 in a similar manner as described with reference to gated transmission system 6501 in Figure 65.

4.1.3 Implementation examples of the switching module

The switch module 6502 of FIG. 65 and the switch modules 6704 and 6706 of FIG. 67A may be any switching element, preferably having a relatively low impedance when closed and a relatively high impedance when open. The switch modules 6502, 6704, and 6706 may be implemented with normally open or normally closed switches. The switch modules need not be ideal switch modules.

Fig. 66B shows the switching modules 6502, 6704, and 6706 as, for example, a switching module 6610. The switching module 6610 (e.g., the switching modules 6502, 6704, and 6706) may be implemented by any suitable type of switching element, e.g., mechanical switching elements and electrical switching elements, optical switching elements, etc., and combinations thereof. Such elements include, for example, transistor switching elements, diode switching elements, relay switching elements, optical switching elements, micromachine switching elements, etc.

In one embodiment, the switching module 6610 may be implemented as a transistor, e.g., as a field effect transistor (FET), bipolar transistor, or any other suitable circuit switching element.

In Fig. 66A, the switching module 6610 is shown as a FET 6602. The FET 6602 can be any type of FET, e.g., a MOSFET, a JFET, a GaAsFET, etc. The FET 6602 has a gate electrode 6604, a source electrode 6606, and a drain electrode 6608. The gate electrode 6604

·· 4

5 ··

receives the energy transfer signal 6306 to control the switching function between the source electrode 6606 and the drain electrode 6608. In one embodiment, the source electrode 6606 and the drain electrode 6608 are interchangeable.

The representation of the switching module 6610 as FET 6602 in Figure 66A is for illustrative purposes only. Any suitable element with switching characteristics could be used to implement the switching module 6610 (e.g., switching modules 6502, 6704, and 6706), as will be apparent to those skilled in the art based on the present description.

In Fig. 66C, the switching module 6610 is designed as a diode switch

6612, which acts as a two-terminal element when the energy transfer signal 6306 is applied to the output

6613 is coupled.

In Fig. 66D, the switching module 6610 is designed as a diode switch

6614, which acts as a two-terminal element when the energy transfer signal 6306 is applied to the output

6615 is coupled.

4.1.4 Implementation examples of the storage module

Storage modules 6506 and 6716 store non-trivial amounts of energy from EM signal 1304. In an exemplary embodiment, storage modules 6506 and 6716 are implemented as a reactive storage module 6801 in Figure 68A, although the invention is not limited to this embodiment. A reactive storage module is a storage module that uses one or more reactive electrical components to store energy transferred from EM signal 1304. Reactive electrical components include, for example, capacitors and inductors.

In one embodiment, memory modules 6506 and 6716 include one or more capacitive storage elements, as shown in Figure 68B as capacitive storage module 6802. In Figure 68C, capacitive storage module 6802 is shown as one or more capacitances, generally represented by capacitance(s) 6804.

It is the function of the memory modules 6506 and 6716 to store non-trivial amounts of energy transferred by the EM signal 1304. Amplitude reproduction of the original, unaffected EM input signal is not necessarily essential. In a power transfer environment, the memory module preferably has the ability to handle the transferred energy and store a non-trivial amount of energy during a time window defined by the non-trivial pulse width.

A terminal 6806 serves as the output of the capacitive storage module 6802. The capacitive storage module 6802 provides the stored energy at the terminal 6806. Figure 68F shows the capacitive storage module 6802 with a series capacitance 6812 which can be used in an inverted gated transmission system described below.

In an alternative embodiment, memory modules 6506 and 6716 include one or more inductive memory elements, as shown in Figure 68D as inductive memory module 6808.

In an alternative embodiment, memory modules 6506 and 6716 include a combination of one or more capacitive memory elements and one or more inductive memory elements, as shown in Figure 68E as capacitive/inductive memory module 6810.

Fig. 68G shows an integrated gated transmission system 6818 that can be implemented so that the EM

·

Signal 1304 is down-converted as shown in Figures 83A-F and described with reference to those figures.

4.1.5 Optional power transfer signal module

69 shows a power transfer system 6901 that is an exemplary embodiment of the power transfer system 6302. The power transfer system 6901 includes an optional power transfer signal module 6902 that can perform any of various functions or combinations of functions, for example, that can generate the power transfer signal 6306.

In one embodiment, the optional energy transfer signal module 6902 includes a time window or pulse width generator, with an example of a pulse width generator being shown in Figure 68J as pulse width generator 6820. The pulse width generator 6820 generates non-marginal pulse width pulses 6826 from an input signal 6824. The input signal 6824 may be any type of periodic signal, e.g., a sinusoidal signal, a square wave, a sawtooth wave, etc. Systems for generating the input signal 6824 are described below.

The width of the pulses 6826 is determined by the delay obtained by the tap 6822 of the pulse width generator 6820. In general, the complexity of the implementation or functional requirements for the pulse width generator 6820 decreases as the desired pulse width is larger. That is, to generate pulses with non-insignificant pulse widths for a given EM input signal frequency, the components used in the exemplary pulse width generator 6820 do not require as fast response times as in a

dersampling system operating at the same EM input signal frequency.

The exemplary logic and implementation of the pulse width generator 6820 are for illustrative purposes only and are not intended to be limiting. The actual logic used may take a variety of forms. The exemplary pulse width generator 6820 includes an optional inverter 6828 intended to provide polarity consistency with other examples presented herein.

An implementation example of pulse width generator 6820 is shown in Figure 68K. Additional examples of pulse width generation logic are shown in Figures 68H and 68I. Figure 68H shows a rising edge pulse generator 6840 that generates pulses 6826 on rising edges of input signal 6824. Figure 68I shows a falling edge pulse generator 6850 that generates pulses 6826 on falling edges of input signal 6824.

In one embodiment, input signal 6824 is generated externally by energy transfer signal module 6902, as shown in Figure 69. Alternatively, input signal 6824 is generated internally by energy transfer signal module 6902. Input signal 6824 may be generated by an oscillator, as shown in Figure 68L by oscillator 6830. Oscillator 6830 may be an internal or external device of energy transfer signal module 6902. Oscillator 6830 may be an external device of energy transfer system 6901. The output signal of oscillator 6830 may be any periodic waveform.

The type of down-conversion performed by the energy transfer system 6901 depends on the aliasing rate of the energy transfer signal 6306, which is determined by the frequency

quency of the pulses 6826. The frequency of the pulses 6826 is determined by the frequency of the input signal 6824. For example, if the frequency of the input signal 6824 is substantially equal to a harmonic or a subharmonic of the EM signal 1304, the EM signal 1304 is down-converted directly to a baseband signal (e.g., if the EM signal is an AM signal or a PM signal) or converted from FM to a non-FM signal. If the frequency of the input signal 6824 is substantially equal to a harmonic or a subharmonic of a difference frequency, the EM signal 1304 is down-converted to an intermediate frequency signal.

The optional 6902 power transfer signal module can be implemented in hardware, software, firmware, or any combination thereof.

4.2 The energy transmission system as an inverted gate-controlled transmission system

64B shows an exemplary inverted gated transfer system 6406 that is an implementation example of the energy transfer system 6302. The inverted gated transfer system 6406 includes an inverted gated transfer module 6408 that receives the EM signal 1304 and the energy transfer signal 6306. The energy transfer signal 6306 includes a train of energy transfer pulses with non-minor pulse widths whose duration tends away from zero. The energy transfer pulses repeat at an aliasing rate F _AR . The inverted gated transfer module 6408 transfers energy from the EM signal 1304 at the aliasing rate of the energy transfer signal 6306, as described in the previous sections with reference to the timing diagrams.

4601 in Fig. 46A, 4607 in Fig. 46B, 4613 in Fig. 46C, and 4619 in Fig. 46D. The inverted gated transmission module 6408 outputs a down-converted signal 1308B having non-trivial amounts of energy transmitted from the EM signal 1304.

4.2.1 The inverted gated transmission system as a switching module and a storage module

74 shows an exemplary embodiment of the inverted gated transfer module 6408 including a switching module 7404 and a memory module 7406. The switching module 7404 and the memory module 7406 preferably transfer energy from the EM signal 1304 to down-convert it in any of the manners illustrated in timing diagrams 4601 of FIG. 46A, 4607 of FIG. 46B, 4613 of FIG. 46C, and 4619 of FIG. 46D.

The switching module 7404 may be implemented as shown above with reference to Figures 66A-D. The memory module 7406 may be implemented as shown above with reference to Figures 68A-F.

In the illustrated embodiment, the storage module 7206 includes one or more capacitors 7408. The capacitor(s) 7408 are selected to pass higher frequency components of the EM signal 1304 to a terminal 7410 regardless of the state of the switching module 7404. The capacitor 7408 stores non-trivial amounts of energy from the EM signal 1304. The signal at terminal 7410 is then offset by an amount related to the energy stored in the capacitor 7408.

The operation of the inverted gated transmission system 7401 is shown in Figures 75A-F. Figure 75A shows the EM signal 1304. Figure 75B shows

the EM signal 1304 after energy has been transferred therefrom. Figure 75C shows the energy transfer signal 6306, which comprises a train of energy transfer pulses with non-minor pulse widths.

Figure 75D shows an exemplary down-converted signal 1308B. Figure 75E shows the down-converted signal 1308B in a compressed time scale. Because the memory module 7406 is a series element, the higher frequencies (e.g., RF) of the EM signal 1304 are visible on the down-converted signal. These can be filtered out as shown in Figure 75F.

The 7401 inverted gated transmission system can be used to down-convert any type of EM signal, such as modulated carrier signals and unmodulated carrier signals.

4.3 Rail-to-rail operation for improved dynamic range
4.3.1 Introduction

63. Aliasing module 11000 includes UFT module 11004 and memory module 11008. As shown in FIG. 63, UFT module 11004 is implemented as n-channel FET 11006 and memory module 11008 is implemented as capacitor 11010, but the invention is not limited to this embodiment.

The FET 11006 receives the EM signal 11002 and the aliasing signal 11014. In one embodiment, the

Aliasing signal 11014 comprises a train of pulses having non-minor pulse widths that repeat at an aliasing rate. The aliasing rate may be a harmonic or a subharmonic of the frequency of EM signal 11002. FET 11006 samples EM signal 11002 at the aliasing rate of aliasing signal 11014 to produce down-converted signal 11012. In one embodiment, aliasing signal 11014 controls the gate of FET 11006 so that FET 11006 conducts (or turns on) when the gate-source voltage (V _GS ) of the FET is greater than a threshold voltage (V _T ). When FET 11006 conducts, a channel is created from the source to drain region of FET 11006 so that charge is transferred from EM signal 11002 to capacitor 11010. In particular, the conductance (l/R) of FET 11006 as a function of V _GS is a continuous function reaching an appropriate level at V _T as shown in Fig. 10B. The charge stored by capacitor 11010 during successive samples forms down-converted signal 11012.

As described above, the n-channel FET 11006 conducts when V _GS exceeds the threshold voltage V _T . As shown in Fig. HOA, the gate voltage of the FET 11006 is determined by the aliasing signal 11014, and the source voltage is determined by the EM input signal 11002. The aliasing signal 11014 preferably consists of several pulses whose amplitude is predictable and adjustable by a system designer. However, the EM signal 11002 is typically received by a coupling device (e.g., an antenna) over a communication or transmission medium. Therefore, the amplitude of the EM signal 11102 may be variable and depend on several factors, e.g., the strength of the transmitted signal and

the attenuation of the communication or transmission medium. Therefore, the source voltage of the FET 11006 is not completely predictable and will therefore affect Vqs and the conductance of the FET 11006.

Fig. 11A shows an example of the EM signal 11102 appearing at the source of the FET 11006. The EM signal 11102 has a relatively high amplitude portion 11104 as shown. Fig. 11B shows the aliasing signal 11106 as an example of an aliasing signal 11014 controlling the gate of the FET 11006. Fig. 11C shows the voltage V _GS 11108 representing the difference between the gate and source voltages as shown in Figs. 11B and 11A, respectively. The FET 11006 has an inherent threshold voltage V _T 11112 as shown in Fig. 11C, above which the FET 11006 conducts. Preferably, during each pulse of the aliasing signal 11106, V _GS > V _T so that the FET 11006 conducts and during each pulse of the aliasing signal 11106, charge is transferred from the EM signal 11102 to the capacitor 11010. As shown in Fig. 1C, the portion 11104 of the EM signal 11102 where the signal has a high amplitude causes a V _GS pulse 11110 that does not exceed V _T 11112 so that the FET 11006 does not become fully conductive as desired. Therefore, the quality of the obtained sample of the EM signal 11102 may be compromised, possibly adversely affecting the down-converted signal 11012.

As mentioned above, the conductance of the FET 11006 as a function of V _GS is mathematically continuous and not sharply limited. That is, the FET 11006 will only be marginally conductive when controlled by the pulse 11110: even if the pulse 11110 is smaller than V _T 11112. However, the insertion loss of the FET 11006 will be significantly lower compared to

·

a V _GS pulse Hill that is greater than V _T 11112. The performance loss caused by a large amplitude input signal is often referred to as clipping or compression. Clipping causes noise in the downconverted signal 11012 that adversely affects proper downconversion of the EM input signal 11102. Dynamic range is related to the range of input signals that can be properly downconverted without noise being induced in the downconverted signal. The higher the dynamic range of a downconversion circuit, the greater the amplitudes of the input signals that can be downconverted without noise being induced in the downconverted signal.

4.3.2 Complementary UFT structure for improved dynamic range

112 shows an aliasing module 11200 according to an embodiment of the invention that downconverts an EM signal 11208 using an aliasing signal 11220 to produce a downconverted signal 11214. The aliasing module 11200 is capable of downconverting input signals over a larger range of amplitudes compared to the aliasing module 11000, so that the aliasing module 11200 has a better dynamic range compared to the aliasing module 11000. The dynamic range is improved because the aliasing module 11200 has two UFT modules implemented with complementary FET devices. That is, one FET is an n-channel FET and the other FET is a p-channel FET, so that at least one FET is always conducting during an aliasing pulse, provided that the input signal does not exceed the power supply limits. The aliasing module 11200 comprises: a delay device

11202, UFT modules 11206, 11216, nodes 11210, 11212 and an inverter 11222. The inverter 11222 is connected to power supplies V ₊ 11232 and V_ 11234. The UFT module 11206 has an n-channel FET 11204 and the UFT module 11216 has a p-channel FET 11218.

As mentioned, the aliasing module 11200 is operated with two complementary FETs to extend the dynamic range and reduce any spurious effects. This requires that two complementary aliasing signals 11224 and 11226 be generated from the aliasing signal 11220 to control sampling by the FETs 11218 and 11204, respectively. To accomplish this, the inverter 11222 receives an aliasing signal 11220 and inverts it to generate an aliasing signal 11224 that controls the p-channel FET 11218. The delay device 11202 delays the aliasing signal 11220 to generate an aliasing signal 11226, with the time delay approximately equal to the time delay associated with the inverter 11222. As a result, the aliasing signals 11224 and 11226 have approximately complementary amplitudes.

Node 11210 receives EM signal 11208 and provides EM signals 11227, 11228 to the sources of n-channel FET 11204 and p-channel FET 11218, respectively, where EM signals 11227, 11228 are substantially copies of EM signal 11208. The n-channel FET 11204 samples the EM signal 11227 controlled by the aliasing signal 11226 and produces samples 11236 at the drain of the FET 11204. Similarly, the p-channel FET 11218 samples the EM signal 11228 controlled by the aliasing signal 11224 and produces samples 11238 at the drain of the FET 11218. The node 11212 combines the obtained charge samples into charge samples 11240 which are stored by a capacitor 11230. The charge stored by the capacitor 11230 during successive samples

Charge forms a down-converted signal 11214. The aliasing module 11200 provides better dynamic range than the aliasing module 11000 because the n-channel FET 11204 and the p-channel FET 11214 are complementary elements. Therefore, when one element is turned off due to a large amplitude EM input signal 11208, the other element will be conductive and sample the input signal as long as the amplitude of the input signal is between the supply voltages V ₊ 11232 and V_ 11234. This is often referred to by those skilled in the art as rail-to-rail operation.

113A shows an exemplary EM signal 11302, which is an example of the EM signals 11227, 11228 provided to the source electrodes of the n-channel FET 11204 and the p-channel FET 11218, respectively. As shown, the EM signal 11302 has a relatively high amplitude portion 11304 comprising pulses 11303, 11305. FIG. 113B shows the aliasing signal 11306 as an example of the aliasing signal 11226 that controls the gate electrode of the n-channel FET 11204. Similarly, for the p-channel FET, Figure 113D shows the aliasing signal 11314 as an example of the aliasing signal 11224 that controls the gate of the p-channel FET 11218. The aliasing signal 11314 is amplitude complementary with respect to the aliasing signal 11306.

Fig. 113C shows the voltage V _GS 11308 corresponding to the difference between the gate and source voltages of the n-channel FET 11204 shown in Figs. 113B and 113A, respectively. Fig. 113C also shows the inherent threshold voltage V _T 11309 for the FET 11204 above which the FET 11204 conducts. Similarly, Fig. 113E shows the voltage V _GS 11316 for the p-channel FET corresponding to the difference between the gate and source voltages of the p-channel FET 11218 shown in Figs. 113D and 113A, respectively.

Fig. 113E also shows the inherent threshold voltage V _T 11317 for the FET 11218, below which the FET 11218 conducts.

As mentioned, the n-channel FET 11204 conducts when V _GS

11308 is greater than V _T 11309, and the p-channel FET 11218 conducts when V _GS 11316 is less than V _T 11317. As shown in Fig. 113C, the n-channel FET 11204 conducts over the range of the EM signal 11302 shown in Fig. 113A except during the EM signal pulse 11305, which results in a corresponding V _GS pulse 11310 (Fig. 113C) that is V _T

11309. However, the p-channel FET 11218 conducts because the same EM signal pulse 11305 produces a V _GS pulse 11320 (Fig. 113E) that is suitably smaller than that of V _T 11317 for the p-channel FET. Therefore, the sample of the EM signal 11302 is suitably taken up by the p-channel FET 11218 and no noise is induced in the down-converted signal 11214. Similarly, a V _GS pulse 11322 (Fig. 113E) is obtained by the EM signal pulse 11303 that is not sufficient to render the p-channel FET 11218 fully conductive. However, the n-channel FET 11204 does not become fully conductive because the same EM signal pulse 11303 produces a V _GS pulse 11311 (Fig. 113C) that is much larger than V _T 11309.

As shown above, a better dynamic range is obtained by the aliasing module 11200 than by the aliasing module 11000 due to the complementary FET structure. Any input signal with an amplitude within the supply voltages V ₊ 11232 and V_ 11234 will switch the FET 11204 and/or the FET 11218 to the conducting state, as shown by Figures 113A-113E. This is because any input signal that has a voltage

voltage V _GS that turns off the n-channel FET 11204 will turn the p-channel FET 11218 into a conducting state. Similarly, any input signal that turns off the p-channel FET 11218 will turn the n-channel FET 11204 into a conducting state, thus avoiding any disturbance to the down-converted signal.

4.3.3 Configuration with preload

Figure 114 shows an aliasing module 11400 which is an alternate embodiment of module 11200. Aliasing module 11400 includes positive voltage supply (V ₊ ) 11402, resistors 11404, 11406 and elements within aliasing module 11200. V ₊ 11402 and resistors 11404, 11406 produce a positive DC voltage at node 11405. This allows node 11405 to operate an attached circuit requiring a positive supply voltage and enables unipolar supply to aliasing module 11400. The positive supply voltage also has the effect of increasing the DC level of EM input signal 11208. Therefore, any input signal with an amplitude between V ₊ 11402 and ground will cause either FET 11204 and/or FET 11218 to switch to the conducting state, as will be apparent to those skilled in the art based on the present description.

Fig. 115 shows an aliasing module 11500 that includes an alternative bias configuration of the aliasing module 11200. The aliasing module 11500 includes a positive voltage supply 11502, a negative voltage supply 11508, resistors 11504, 11506, and the elements in the aliasing module 11200. By using both a positive and a negative supply voltage, the

Node 11505 may be provided with any voltage between V ₊ 11502 and V_ 11508. This allows node 11505 to operate an attached circuit that requires a positive or negative supply voltage. Additionally, any input signal having an amplitude between supply voltages V ₊ 11502 and V_ 11508 will cause either FET 11204 and/or FET 11218 to switch to the conducting state, as will be apparent to those skilled in the art from this description.

4.3.4 Simulation results

As mentioned, an aliasing module having a complementary FET structure has an improved dynamic range compared to a single (or unipolar) FET configuration. This is further illustrated by comparing the signal waveforms of the aliasing module 11602 (Fig. 116) having a complementary FET structure with those of the aliasing module 11702 (Fig. 117) having a single (or unipolar) FET structure.

The aliasing module 11602 (Fig. 116) down-converts an EM signal 11608 using an aliasing signal 11612 to produce a down-converted signal 11610. The aliasing module 11602 has a complementary FET structure and includes an n-channel FET 11604, a p-channel FET 11606, an inverter 11614, and an aliasing signal generator 11608. The aliasing module 11602 is biased by a supply circuit 11616, as shown. An aliasing module 11702 (Fig. 117) downconverts an EM signal 11704 using an aliasing signal 11708 to produce a downconverted signal 11706. The aliasing module 11702 has a single FET structure with an n-channel FET 11712 and a

aliasing signal generator 11714 and is biased using a voltage supply circuit 11710.

Figures 118-120 show signal waveforms associated with aliasing module 11602, and Figures 121-123 show signal waveforms associated with aliasing module 11702. Figures 118, 121 show down-converted signals 11610 and 11706, respectively. Figures 119, 122 show sampled EM signal 11608 and 11704, respectively. Figures 120, 123 show aliasing signals 11612 and 11708, respectively. Aliasing signal 11612 is identical to aliasing signal 11708 to more conveniently compare modules 11602 and 11702.

The EM signals 11608, 11704 are input signals with relatively large amplitudes approaching the supply voltages of ±1.65 V, as shown in Figures 119 and 122, respectively. In Fig. 119, portions 11902 and 11904 of signal 11608 represent energy transfer from EM signal 11608 to down-converted signal 11610 through aliasing module 11602. Portion 11902 represents energy transfer near the -1.65V supply voltage, and portion 11904 represents energy transfer near the +1.65V supply voltage. The symmetry quality of energy transfer near the supply voltage rails indicates that at least one of complementary FETs 11604, 11606 is properly sampling the EM signal during each of aliasing pulses 11612. This results in a down-converted signal 11610 with minimal high frequency noise that is between

-1.0 V and +1.0 V (i.e. has a negligible DC component).

Similarly, in Fig. 122, sections 12202 and 12204 illustrate the energy transfer from the EM signal 11704 to the down-converted signal 11706 through the aliasing module 11702 (single FET configuration). Section 12202 illustrates

·

&bull; *

represents the energy transfer near the -1.65 V supply voltage, and section 12204 represents the energy transfer near the +1.65 V supply voltage. By comparing sections 12202, 12204 with sections 11902, 11904 of Fig. 119, it is clear that the energy transfer in sections 12202, 12204 near the supply voltage rails is not as asymmetric as that in sections 11902, 11904. Therefore, it is clear that EM signal 11704 partially disables single FET 11712 over a portion of the trace of signal 11704. This results in a down-converted signal 11706 that has a larger high frequency noise component and a significant negative DC component compared to the down-converted signal 11610.

The down-converted signal 11706 reflects a disturbance induced by a relatively large amplitude EM signal that disables the single FET 11712 in the aliasing module 11702. The down-converted signal 11610 generated by the aliasing module 11602 is relatively noise-free. This is because the complementary FET configuration in the aliasing module 11602 is able to handle large amplitude input signals without inducing a disturbance in the down-converted signal 11610. Therefore, the complementary FET configuration in the aliasing module 11602 has a better dynamic range compared to the single FET configuration of the aliasing module 11702.

4.4 Optimized switching structures
4.4.1 Splitters in CMOS

Fig. 124A shows an embodiment of a splitter circuit 12400 implemented in CMOS. The embodiment

22· X ¹ MM-

is for illustrative purposes only and should not be taken in a limiting sense. In one embodiment, splitter circuit 12400 is used to split a local oscillator (LO) signal into two oscillation signals that are approximately 90° out of phase. The first oscillation signal is referred to as the I-channel oscillation signal. The second oscillation signal is referred to as the Q-channel oscillation signal. The Q-channel oscillation signal is approximately 90° out of phase with the I-channel oscillation signal. The splitter circuit 12400 includes a first I-channel inverter 12402, a second I-channel inverter 12404, a third I-channel inverter 12406, a first Q-channel inverter 12408, a second Q-channel inverter 12410, an I-channel flip-flop 12412, and a Q-channel flip-flop 12414.

Figures 124F-J show exemplary waveforms used to illustrate signal relationships of splitter circuit 12400. The waveforms illustrated in Figures 124F-J reflect ideal delay times through components of splitter circuit 12400. An LO signal 12416 is illustrated in Figure 124F. The first, second, and third I-channel inverters 12402, 12404, and 12406 invert the LO signal 12416 three times, outputting an inverted LO signal 12418, as illustrated in Figure 124G. The first and second Q-channel inverters 12408 and 12410 invert the LO signal 12416 twice, outputting a non-inverted LO signal 12420, as illustrated in Figure 124H. The delay through the first, second and third I-channel inverters 12402, 12404 and 12406 is substantially equal to that through the first and second Q-channel inverters 12408 and 12410, so that the inverted LO signal 12418 and the non-inverted LO signal 12420 are approximately 180°

out of phase. The operating characteristics of the inverters can be adjusted to obtain the appropriate delay values, as will be apparent to those skilled in the art.

The I-channel flip-flop 12412 is supplied with the inverted LO signal 12418. The Q-channel flip-flop 12414 is supplied with the non-inverted LO signal 12420. In the present embodiment, the I-channel flip-flop 12412 and the Q-channel flip-flop 12414 are edge-triggered flip-flops. When either flip-flop receives a rising edge at its input, the flip-flop output changes state. Therefore, the I-channel flip-flop 12412 and the Q-channel flip-flop 12414 each output signals having approximately half the input signal frequency. In addition, as will be appreciated by those skilled in the art, because the input signals of I-channel flip-flop 12412 and Q-channel flip-flop 12414 are approximately 180° out of phase, their output signals are approximately 90° out of phase. I-channel flip-flop 12412 outputs I-channel oscillation signal 12422 as shown in Figure 124I. Q-channel flip-flop 12414 outputs Q-channel oscillation signal 12424 as shown in Figure 124J. Q-channel oscillation signal 12424 is 90° out of phase with I-channel oscillation signal 12422 as can be appreciated by comparing Figures 124I and 124J.

Fig. 124B shows a more detailed circuit structure of the splitter circuit 12400 of Fig. 124. The circuit blocks of Fig. 124B that are similar to those of Fig. 124A are designated by corresponding reference numerals. Figs. 124C-D show exemplary output waveforms related to the splitter circuit 12400 of Fig. 124B. Fig. 124C shows an I-channel oscillation signal 12422. Fig. 124D shows a Q-channel oscillation signal 12424. As can be seen by comparing Figs. 124C and 124D, the waveform of the Q-channel oscillation signal is

&bull; · ,&igr;. *&diams;

12424 of Fig. 124D is approximately 90° out of phase with respect to the waveform of the Q-channel oscillation signal 12422 of Fig. 124C.

The splitter circuit 12400 in Figures 124A and 124B is shown for illustration purposes only. The splitter circuit 12400 may be comprised of a combination of various logic and semiconductor devices, as will be apparent to those skilled in the art based on the present description.

4.4.2 I/Q circuit

Figure 124E shows an exemplary embodiment of a complete I/Q circuit 12426 in CMOS. The I/Q circuit 12426 includes a splitter circuit 12400 described in detail above. I/Q implementations are described in more detail below, including the applications referenced above.

4.5 Exemplary I and Q implementations
4.5.1 Switches with different sizes

In one embodiment, the switching modules discussed herein may be implemented as a series of switches that function in parallel as a single switch. The switch series may consist of transistors, e.g., field effect transistors (FETs), bipolar transistors, or any other suitable circuit switching elements. The switch series may consist of one type of switching element or a combination of different switching elements.

For example, Fig. 125 shows a switching module 12500. In Fig. 125, the switching module is shown as a series of FETs 12502a-n. The FETs 12502a-n can be any type of FET, e.g., a MOSFET, a JFET, a GaAsFET, etc. Each of the FETs 12502a-n has a gate electrode 12504a-n, a source electrode 12506a-n, and a drain electrode 12508a-n, similar to the FET 2802 of Fig. 28A. The series of FETs 12502-n operate in parallel. The gate electrodes 12504a-n are connected together, the source electrodes 12506a-n are connected together, and the drain electrodes 12508a-n are connected together. Each of the gate electrodes 12504a-n receives the control signal 1604, 8210 to control the switching function between corresponding source electrodes 12506a-n and drain electrodes 12508a-n. In general, the corresponding source electrodes 12506a-n and drain electrodes 12508a-n of the FETs 12502a-n are interchangeable. The number of FETs is not limited. Any limitation would depend on the specific application and the designation "a-n" is not intended to be limiting in any way.

In one embodiment, the FETs 12502a-n have similar characteristics. In another embodiment, one or more of the FETs 12502a-n have different characteristics than the other FETs. For example, the FETs 12502a-n may have different sizes. In CMOS, in general, the larger a switch is (i.e., the larger the area under the gate between the source and drain regions), the longer it takes to turn on. The longer turn on time is due in part to a higher gate to channel capacitance present in larger switches. Smaller CMOS switches turn on in a shorter time, but they have a higher channel resistance. Larger CMOS switches have a

lower channel resistance. By providing different turn-on characteristics or lines for different switch sizes, flexibility is provided in the design of the entire switch module structure. By combining smaller switches with larger switches, the channel conductance of the entire switch structure can be adjusted to meet specified requirements.

In one embodiment, FETs 12502a-n are CMOS switches of relatively different sizes. For example, FET 12502 may be a switch that is smaller than FETs 12502a-n. FET 12502b may be a switch that is larger than FET 12502a, but smaller than FETs 12502c-n. The sizes of FETs 12502c-n may also be different relative to one another. For example, increasingly larger switch sizes may be used. By changing the sizes of FETs 12502a-n relative to one another, the turn-on characteristic of the switching module may be changed accordingly. For example, the turn-on characteristic of the switching module may be adjusted to more closely approximate that of an ideal switch. Alternatively, the switching module may be adjusted to produce a shaped conductance curve.

By configuring the 12502a-n FETs so that one or more of the FETs are relatively smaller in size, their faster turn-on characteristics can improve the turn-on characteristics of the entire switch module. Because smaller switches have lower gate/channel capacitance, they can turn on faster than larger switches.

By configuring the 12502a-n FETs so that one or more of the FETs have a relatively larger size, the turn-on characteristic of the entire switching module can also be improved due to their lower channel resistance.

·

·

Because larger switches have lower channel resistance, they can provide lower channel resistance to the overall switching structure even when combined with smaller switches. This improves the ability of the overall switching structure to drive a wider range of loads. The ability to adjust switch sizes relative to one another throughout the overall switching structure therefore allows the operation of the overall switching structure to more closely approximate an ideal switch, or to meet application-specific requirements, or to balance trade-offs to enable specific tasks, as will be apparent to those skilled in the art from this description.

The representation of the switching module as a series of FETs 12502a-n in Figure 125 is for illustrative purposes only. Any element with switching functions could be used to implement the switching module (e.g., switching modules 2802, 2702, 2404, and 2406), as will be apparent to those skilled in the art from this description.

4.5.2 Reduction of the total switch area

Circuit performance can also be improved by reducing the overall switch area. As described above, smaller switches (i.e., switches with a smaller area under the gate region between the source and drain regions) have a smaller gate/channel capacitance than larger switches. The smaller gate/channel capacitance enables a lower circuit sensitivity to noise signal disturbances. Figure 126A shows an embodiment of a switch module with a large overall switch area. The switch module of Figure 126A includes 20 FETs 12602-12640.

As shown, the FETs 12602-12640 are the same size (the "Wd" and "Ing" parameters are the same). An input signal source 12646 generates the EM input signal. A pulse generator 12648 generates the energy transfer signal for the FETs 12602-12640. A capacitor Cl is the storage element for the input signal sampled by the FETs 12602-12640. Figures 126B-126Q show exemplary waveforms associated with the switching module of Figure 126A. Figure 126B shows a received EM signal having a frequency of 1.01 GHz to be sampled and down-converted to a 10 MHz intermediate frequency signal. Fig. 126C shows a power transfer signal having an aliasing rate of 200 MHz applied to the gate of each of the 20 FETs 12602-12640. The power transfer signal comprises a train of power transfer pulses with non-minor pulse widths tending away from zero in duration. The power transfer pulses repeat at the aliasing rate. Fig. 126D shows the affected received EM signal for illustrating effects of power transfer at the aliasing rate at a point 12642 of Fig. 126A. Fig. 126E shows a down-converted signal produced by the down-conversion process at a point 12644 of Fig. 126A.

Fig. 126F shows the frequency spectrum of the received 1.01 GHz signal. Fig. 126G shows the frequency spectrum of the received power transfer signal. Fig. 126H shows the frequency spectrum of the affected received EM signal at a point 12642 of Fig. 126A. Fig. 126I shows the frequency spectrum of the down-converted signal at a point 12644 of Fig. 126A.

Figures 126J-126M also show the frequency spectra of the received 1.01 GHz EM signal, the received energy transfer signal, the influenced received EM

signal at point 12642 of Fig. 126A and the down-converted signal at point 12644 of Fig. 126A in a narrower frequency range centered at 1.00 GHz. As shown in Fig. 126L, noise signal interference exists at about 1.0 GHz on the affected received EM signal at point 12642 of Fig. 126A. This noise signal interference may be radiated through the circuit, causing interference at 1.0 GHz to nearby receivers.

Figures 126N-126Q show the frequency spectra of the received 1.01 GHz EM signal, the received power transfer signal, the affected received EM signal at point 12642 of Figure 126A, and the downconverted signal at point 12644 of Figure 126A over a narrow frequency range centered near 10.0 MHz. Figure 126Q shows that at 10 MHz a signal with an amplitude of about 5 mV was downconverted.

Fig. 127A shows an alternative embodiment of the switching module, which has 14 FETs 12702-12728 instead of 20 FETs 12602-12640 as shown in Fig. 126A. In addition, the FETs have different sizes (some of the parameters "Wd" and "Ing" differ between the FETs).

Figures 127B-127Q, which illustrate exemplary waveforms associated with the switching module of Figure 127A, correspond to the similarly labeled Figures 126B-126Q. As shown in Figure 127L, a lower level noise signal interference exists at about 1.0 GHz than at the same frequency in Figure 126L. This is correlated with a lower level of circuit radiation. In addition, as shown in Figure 127Q, the lower level noise signal interference was obtained without reducing the conversion efficiency. This is illustrated in Figure 127Q by the down-converted 5 mV signal at about 10 MHz.

This voltage is substantially equal to the level down-converted by the circuit of Figure 126A. By reducing the number of switches, thereby decreasing the total switch area, and by reducing the switch area on a switch-by-switch basis, parasitic circuit capacitances can be reduced, as will be apparent to those skilled in the art from this description. In particular, this can reduce the total gate/channel capacitance, thereby obtaining lower amplitude noise signals and reducing circuit radiation.

The above depiction of the switches as FETs in Figures 126A-126Q and 127A-127Q is for illustrative purposes only. Those skilled in the art will appreciate, based on the present description, that any element having switching functions can be used to implement the switching module.

4.5.3 Charge carrier injection compensation

In embodiments where the switching modules discussed herein consist of a series of parallel switches, it may be desirable in some cases to minimize the effects of carrier injection. Minimizing carrier injection is generally desirable to reduce the unwanted circuit radiation resulting therefrom. In one embodiment, unwanted carrier injection effects can be reduced by using complementary n-channel MOSFETs and p-channel MOSFETs. Carrier injection adversely occurs in both n-channel MOSFETs and p-channel MOSFETs. However, because signals of opposite polarity are applied to their respective gate electrodes to turn the switches on and off, the resulting carrier injection is of opposite polarity. Therefore, the n-channel MOSFETs and the

p-channel MOSFETs are arranged in pairs to compensate for their respective charge carrier injection. Therefore, in one embodiment, the switching module may consist of n-channel MOSFETs and p-channel MOSFETs, the size of the respective elements being such that the undesirable effects of charge carrier injection are minimized.

Fig. 129A shows an alternative embodiment of the switching module which has 14 η-channel FETs 12902-12928 and 12 p-channel FETs 12930-12952 instead of 20 FETs 12602-12640 as shown in Fig. 126A. The η-channel and p-channel FETs are arranged in a complementary manner. In addition, the FETs have different sizes (some of the parameters "Wd" and "Ing" are different between the FETs).

Figures 129B-129Q, which illustrate exemplary waveforms associated with the switching module of Figure 129A, correspond to similarly labeled Figures 126B-126Q. As shown in Figure 129L, a lower level noise signal interference exists at about 1.0 GHz than at the same frequency of Figure 126L. This is correlated with lower circuit radiation levels. In addition, as shown in Figure 129Q, the lower level noise signal interference was obtained without reducing conversion efficiency. This is illustrated in Figure 129Q by the down-converted about 5 mV signal at about 10 MHz. This voltage is substantially equal to the level down-converted by the circuit of Figure 126A. By arranging the switches in a complementary manner, which contributes to reducing the charge carrier injection, and by adjusting the switch area switch by switch, the effects of charge carrier injection can be reduced, as will be apparent to those skilled in the art from the present description. In particular, noise signal disturbances with a lower amplitude

and obtain less unwanted circuit radiation.

The use of the FETs in Figures 129A-126Q in the foregoing description is for illustrative purposes only. Those skilled in the art will appreciate, based on the present description, how charge injection can be handled in various transistor technologies using transistor pairs.

4.5.4 Overlapped capacity

The processes used to manufacture semiconductor circuits, such as MOSFETs, have limitations. In some cases, these process limitations can result in circuits that do not function as ideally as desired. For example, a non-ideally manufactured MOSFET may have parasitic capacitances that can, in some cases, cause the surrounding circuit to radiate noise. By manufacturing circuits with structures that are as close to an ideal structure as possible, problems of non-ideal circuit function can be minimized.

Fig. 128A shows a cross-section of an exemplary n-channel boost mode MOSFET 12800 with ideally shaped (n+) regions. The MOSFET 12800 includes a gate contact 12802, a channel region 12804, a source contact 12806, a source region 12808, a drain contact 12810, a drain region 12812, and an insulator 12814. The source region 12808 and the drain region 12812 are separated by p-doped material of the channel region 12804. In the illustration, the source region 12808 and the drain region 12812 are made of an (n+) material. The (n+) material is typically formed by an ion implantation/diffusion process.

1259*

fusion process into the p-doped material of channel region 12804. Ion implantation/diffusion processes are known to those skilled in the art. Insulator 12814 insulates gate contact 12802, which bridges the p-doped material. Insulator 12814 generally comprises a metal oxide insulator. The channel current between source region 12808 and drain region 12812 for MOSFET 12800 is controlled by a voltage at gate contact 12802.

The operation of the MOSFET 12800 is described below. When a positive voltage is applied to the gate contact or gate electrode 12802, electrons in the p-doped material of the channel region 12804 are drawn to the surface beneath the insulator 12814, forming a connecting near-surface region of n-doped material between the source and drain regions, called a channel. The greater the positive voltage between the gate contact 12806 and the source region 12808, the lower the resistance across the region therebetween.

In Figure 128A, the source region 12808 and the drain region 12812 are shown to have (n+) regions formed as idealized rectangular regions by the ion implantation process. Figure 128B shows a cross-section of an exemplary n-channel boost mode MOSFET 12816 with non-ideally shaped (n+) regions. The source region 12820 and the drain region 12822 are shown as irregularly shaped regions by an ion implantation process. Due to uncertainties in the ion implantation/diffusion process, the source region 12820 and the drain region 12822 do not form rectangular regions in practical applications as shown in Figure 128A. Fig. 128B shows a source region 12820 and a drain region 12822, the irregular

■/·'/· : :2eo-:J j.

regions. Additionally, due to these process uncertainties, the (n+) regions of source region 12820 and drain region 12822 may diffuse further than desired into the p-doped region of channel region 12818 and extend under gate region 12802. The extension of source region 12820 and drain region 12822 under gate region 12802 is shown as source overlap 12824 and drain overlap 12826. Source overlap 12824 and drain overlap 12826 are shown in more detail in Figure 128C. Figure 128C shows a top view of an exemplary configuration of the MOSFET 12816. The source overlap 12824 and the drain overlap 12826 may result in unwanted parasitic capacitances between the source region 12820 and the gate contact 12802 and between the drain region 12822 and the gate contact 12802. These unwanted parasitic capacitances may interfere with circuit functions. For example, the resulting parasitic capacitances may generate noise signal interference that radiates through the circuit, causing unwanted electromagnetic interference.

As shown in Figure 128C, an exemplary MOSFET 12816 may include a gate pad 12828. The gate contact 12802 may include a gate extension 12830 and a gate pad extension. The gate extension 12830 is an unused portion of the gate contact 12802 required due to metal implantation process tolerance limits. The gate pad extension 12832 is a portion of the gate contact 12802 required to connect the gate contact 12802 to the gate pad 12828. The contact required for the gate pad 12828 requires that the gate pad extension 12832 have a non-zero length to separate the resulting contact from the portion between the source region 12820 and the drain region.

12822. This prevents the gate contact 12802 from creating a short to the channel between the source region 12820 and the drain region 12822 (the insulator 12814 of Fig. 128B is very thin in this region). Undesirable parasitic capacitances may form between the gate extension 12830 and the substrate (the FET 12816 is fabricated on a substrate) and between the gate pad extension 12832 and the substrate. By reducing the corresponding areas of the gate extension 12830 and the gate pad extension 12832, the parasitic capacitances caused thereby may be reduced. Therefore, the embodiments take into account the uncertainties of the ion implantation/diffusion process. Those skilled in the art will appreciate how the areas of the gate extension 12830 and the gate pad extension 12832 are reduced to reduce the parasitic capacitances they introduce.

The illustration of the n-channel boost mode MOSFET is for illustrative purposes only. Those skilled in the art will appreciate, based on the present description, that the present invention is also applicable to junction MOSFETs and other types of transistors.

4.6 Other implementations

The implementations described above are illustrative. These implementations are not intended to limit the invention. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention that differ slightly or substantially from the implementations described herein.

5. Optional optimizations of energy transfer with an aliasing rate

The methods and systems described in the preceding sections may be optimized by one or more of the optimization methods or systems described below.

5.1 Doubling the aliasing rate (Far) of the energy transfer signal

In one embodiment, the optional energy transfer signal module 6902 of Figure 69 includes a pulse generator module that generates aliasing pulses at twice the frequency of the oscillation source. The input signal 6828 may be any suitable oscillation source.

Figure 71 shows a circuit 7102 that generates a doubler output signal 7104 (Figure 72B) that can be used as the energy transfer signal 6306. The circuit 7102 generates pulses on both the rising and falling edges of the input oscillation signal 7106 of Figure 712A. The circuit 7102 can be implemented as a pulse generator and aliasing rate (FAR) doubler. The doubler output signal 7104 can be used as the energy transfer signal 6306.

In the example of Fig. 71, the aliasing rate corresponds to twice the frequency of the input oscillation signal F _O sc 7106, as shown by the following equation (9).

Eq. (9)

The pulse width of the aliasing pulses is determined by the delay through a first inverter 7108 of Fig. 71. As the delay increases, the pulse width

A second inverter 7112 is shown which serves to maintain polarity consistency with respect to examples described elsewhere. In an alternative embodiment, inverter 7112 is omitted. The pulses preferably have non-trivial pulse widths whose duration tends away from zero. Doubler output signal 7104 may be further suitably adapted to drive a switching module by non-trivial pulse widths. Circuit 7102 may be implemented by an integrated circuit, discretely, by an equivalent logic circuit, or by any suitable manufacturing technique.

5.2 Differential implementations

The invention is implementable by various differential configurations. Differential configurations are useful for reducing in-phase noise. This may be useful in receiver systems where in-phase interference may be caused by intentional or accidental radiators, e.g., cellular phones, CB radios, electrical equipment, etc. Differential configurations are also useful for reducing any in-phase noise caused by carrier injection of the switch in the switching module or due to the design and construction of the system in which the invention is used. Any trace signals induced in equal magnitude and phase into both input terminals of the invention are substantially reduced or eliminated. Some differential configurations, e.g., some of the configurations shown below, are also useful for increasing the voltage and/or increasing the power of the down-converted signal 1308A.

The 4402 differential system is most effective when used with a differential front end (inputs) and a differential back end (outputs). It can also be used in the following configurations, for example:

a) single-input frontend and differential backend; and

b) Differential frontend and single output backend.

Examples of these systems are described below, with a first example representing a specific method by which energy is transferred differentially from input to output.

The example of a differential energy transfer module described below is for illustrative purposes only and is not intended to limit the invention. Those skilled in the art will appreciate, based on the present description, that other embodiments are feasible (e.g., equivalent embodiments, extensions, changes, modifications, etc. of the embodiments described herein). The invention is intended and adapted to include these alternative embodiments.

5.2.1 An example to illustrate differential energy transfer

Fig. 76A shows a differential system 7602 that may be arranged in the energy transfer module 6304. The differential system 7602 has a similar inverted gated transfer construction as the construction described above with reference to Fig. 74. The differential system 7602 has inputs 7604 and 7606 and outputs 7608 and 7610. The differential system 7602 has a first inverted gated transfer module 7612 with a storage module 7614 and a switching module

7616. The differential system 7602 also includes a second inverted gated transfer module 7618 having a memory module 7620 and the switching module 7616 which it shares with the inverted gated transfer module 7612.

One or both of the inputs 7604 and 7606 are connected to an EM signal source. The inputs may be connected to an EM signal source, with the input voltages at inputs 7604 and 7606 having substantially the same amplitude but 180 degrees out of phase with each other. Alternatively, if two input signals are not available, one of the inputs 7604 and 7606 may be connected to ground.

In operation, when the switch module 7616 is closed and the memory modules 7614 and 7620 are connected in series, provided they have similar capacitance values, the memory modules are charged to equal levels but with opposite polarities. When the switch module 7616 is open, the voltage at the output 7608 is referenced to the input 7604 and the voltage at the output 7610 is referenced to the voltage at the input 7606.

Portions of the signals at outputs 7608 and 7610 comprise signals derived from energy or charge stored in storage modules 7614 and 7620, respectively, when switching module 7616 was closed. The portions of the signals at outputs 7608 and 7610 derived from the stored charge have substantially the same amplitude, but are 180 degrees out of phase with each other.

Portions of the signals at outputs 7608 and 7610 also contain voltage ripples or noise generated by the switching function of the switch module 7616. However, because the switch module is connected between the two outputs,

7608 and 7610, the noise generated by the switching module appears at the outputs to be substantially equal in magnitude and in phase relative to each other. This allows the voltage ripples to be substantially cancelled by inverting the voltage at one of the outputs 7608 or 7610 and adding the signal to the signal from the other output. In addition, any noise induced into the input terminals 7604 and 7606 by any other noise sources of substantially equal amplitude and phase will tend to be cancelled in the same manner.

5.2.1.1

Differential input to differential output

Figure 76B shows the differential system 7602 with the inputs 7604 and 7606 connected to equal and opposite EM signal sources, respectively, shown herein as dipole antennas 7624 and 7626. In this embodiment, when one of the signals from the outputs 7608 or 7610 is inverted and added to the signal from the other output, the in-phase noise generated by the switching module 7616 and other in-phase noise present at the input terminals 7604 and 7606 tend to be substantially cancelled.

5.2.1.2

Single input to differential output

Fig. 76C shows the differential system 7602 with the input 7604 connected to an EM signal source, e.g. a monopole antenna 7628, and the input 7606 connected to ground. In this configuration, the voltages at the outputs 7608 and 7610 are approximately half

the voltage value of the implementation shown in Fig. 76B, assuming that all other parameters are equal.

Figure 76E shows an exemplary single input to differential output receiver/downconverter system 7636. System 7636 includes differential system 7602 with input 7606 connected to ground as in Figure 76C. Input 7604 is connected to an EM signal source 7638 through an optional input impedance matching system 7642. The EM signal source impedance can be matched by impedance matching system 7642 as described below in Section 5.

The signals at outputs 7608 and 7610 are fed to a differential circuit 7644, e.g., a filter, which preferably inverts one of the signals at outputs 7608 or 7610 and adds it to the other signal at outputs 7608 or 7610. This substantially cancels the in-phase noise generated by switching module 7616. Differential circuit 7644 preferably filters higher frequency components of EM signal 1304 passing through memory modules 7614 and 7620. The resulting filtered signal is output as down-converted signal 1308B.

5.2.1.3 Differential Input to Single Output

Fig. 76D shows the differential input to single output system 7629, with the inputs 7604 and 7606 of the differential system 7602 connected to equal and opposite EM signal dipole antennas 7630 and 7632, respectively. In system 7629, the in-phase noise signal voltages are not canceled in the same manner as in the systems shown above. The output signal is coupled from terminal 7608 to a load 7648.

5.2.2 Specific alternative embodiments

In specific alternative embodiments, the present invention is implemented using a plurality of gated transfer modules controlled by a common power transfer signal, with a memory module connected between the outputs of the plurality of gated transfer modules. For example, Figure 99 shows a differential system 9902 having first and second gated transfer modules 9904 and 9906 and a memory module 9908 connected therebetween. The operation of the differential system 9902 will be apparent to those skilled in the art based on the present description.

As with the first implementation described in Section 5.5.1 and its subsections, the gated differential transfer system 9902 may be implemented with a single input, differential inputs, a single output, differential outputs, and combinations thereof. For example, Figure 100 shows an exemplary single input to differential output system 10002.

If in-phase rejection is desired to protect the input signal from various in-phase effects, and if in-phase rejection is not required to protect the output signal, a differential input-to-single-output implementation may be used. Fig. 102 shows an exemplary differential-to-single-ended system 10202 in which a balun 10204 is used to generate the differential input. Other input configurations are also conceivable. A first output 10206 is connected to a load

10208. A second output 10210 is connected to a ground point 10212.

In a balun system where a single output signal is taken from a differential system without using a balun (i.e., where one of the output signals is grounded), an attenuation of about 6 dB is obtained. However, in the configuration of Fig. 102, the ground point 10212 simply serves as a DC reference for the circuit. The system 10202 transfers charge from the input in the same way as if it were a fully differential system, with its conversion efficiency generally only affected by the parasitic capacitances of the circuit components used, e.g., the Rds(on) of FET switches if used in the switching module. That is, charge transfer continues in the same manner as in a single ended implementation, providing the required single ended ground for the input circuit when the time window defined by the pulse width is active, but the configuration is still such that the input is differential to enable certain in-phase rejection functions and/or an interface between a differential input and a single ended output system.

5.2.3 Specific examples of optimizations and configurations for inverted and non-inverted differential designs

Gated transmission systems and inverted gated transmission systems can be implemented with any of the various optimizations and configurations shown, e.g. with impedance matching, open circuits and resonance structures, bypass

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transmission networks, etc. For example, the differential system 10002 in Fig. 100 uses gated transmission modules with an input impedance matching system 10004 and an open circuit 10006 that use a common capacitance. Similarly, the differential system 10102 in Fig. 101 uses an inverted gated transmission module with an input impedance matching system 10104 and an open circuit 10106 that use a common capacitance.

5.3 Smoothing the down-converted signal

The down-converted signal 1308B may be smoothed by filtering if necessary. The differential circuit 7644 implemented as a filter in Figure 76E is only an example. This may be implemented by any of the described embodiments through hardware, firmware and software implementations, as will be apparent to those skilled in the art.

5.4 Impedance matching

The energy transfer module has input and output impedances that are generally defined by (1) the duty cycle of the switching module and (2) the impedance of the storage module at the frequencies of interest (e.g., at the frequencies of the EM input signal and the intermediate frequency/baseband signal).

Starting with a pulse width of approximately half the period of the EM signal, which is down-converted as a preferred embodiment, this pulse width (e.g. the "closing time") can be reduced. As the pulse width is reduced, the characteristic impedance or the

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Characteristic impedance at the input and output of the energy transfer module increases. Alternatively, as the pulse width increases from half the period of the down-converted EM signal, the impedance of the energy transfer module decreases.

One of the steps for determining the characteristic impedance at the input of the energy transfer module could be to measure its value. In one embodiment, the characteristic impedance at the input of the energy transfer module is 300 Ω. An impedance matching circuit can be used to couple an EM input signal having a source impedance of, for example, 50 Ω to the energy transfer module impedance of, for example, 300 Ω. This impedance matching can be realized in various ways, e.g. by directly providing the required impedance or by using an impedance matching circuit as described below.

Fig. 70 shows a specific embodiment in which an RF signal is used as the input signal, assuming that the impedance 7012 is a relatively low impedance, e.g., about 50 Ω, and the input impedance 7016 is about 300 Ω, an initial configuration for the input impedance matching module 7006 may include an inductor 7306 and a capacitor 7308 configured as shown in Fig. 73. The inductor 7306 and the capacitor 7308 form an L-matching system, and the calculation of the corresponding values is known to those skilled in the art.

The characteristic impedance or wave impedance at the output can be adjusted to suit the desired output frequencies. One of the

Steps to determine the characteristic impedance at the output of the energy transfer module could be to measure its value. To compensate for the very low impedance of the storage module at the frequency of the EM input signal, the storage module should have an impedance at the desired output frequencies that is preferably greater than or equal to the load to be driven (in one embodiment, the storage module impedance at a desired output signal frequency of 1 MHz is 2 kΩ and that of the load to be driven is 50 Ω). Another advantage of impedance matching is that unwanted signals can also be filtered out by the same components.

In one embodiment, the characteristic impedance at the output of the energy transfer module is 2 kΩ. An impedance matching circuit may be used to couple the down-converted signal, which has, for example, an output impedance of 2 kΩ, to a load of, for example, 50 Ω. The matching of these impedances may be done in various ways, e.g. by directly providing the required load impedance or by using an impedance matching circuit as described below.

When matching from high to low impedance, a capacitance 7314 and an inductive element 7316 may be configured as shown in Figure 73. The capacitance 7314 and the inductive element 7316 form an L-match, and the calculation of the corresponding values for the components is known to those skilled in the art.

It is assumed that the configuration of the input impedance matching module 7006 and the output impedance matching module 7008 according to the invention are initial configurations

for impedance matching. In some cases, the initial configurations may be suitable without further optimization. In other cases, the initial configurations may be optimized according to various design criteria and considerations.

If other optional optimization structures and/or components are used, their effect on the characteristic impedance or wave impedance of the power transfer module should be considered in the adaptation together with their own original criteria.

5.5 Open resonant circuits and resonance structures

Resonant open circuits and other resonant structures can be used to further optimize the energy transfer properties or characteristics of the invention. For example, resonant structures that are resonant at about the input signal frequency can be used to store energy from the input signal when the switch is open, i.e., during a period of time during which one might expect the architecture to otherwise be limited in its maximum possible efficiency. Resonant open circuits and other resonant structures include, for example, surface acoustic wave (SAW) filters, dielectric resonators, diplexers, capacitors, inductive elements, etc.

An exemplary embodiment is shown in Fig. 94A. Two additional embodiments are shown in Fig. 88 and Fig. 97. Alternative implementations will be apparent to those skilled in the art based on the present description. These implementations utilize the properties of series and parallel resonant circuits (open tank circuits).

Fig. 94A shows parallel open circuits in a differential implementation. A first parallel resonant circuit or a first parallel open circuit consists of a capacitor 9438 and an inductive element 9420 (open circuit 1). A second open circuit consists of a capacitor 9434 and an inductive element 9436 (open circuit 2).

As is obvious to experts, parallel open circuits provide:

a low impedance for frequencies below the resonance frequency;

a low impedance for frequencies above the resonance frequency; and

a high impedance for frequencies at or near the resonance frequency.

In the example shown in Fig. 94A, the first and second open tank circuits are resonant at about 920 MHz. At or near resonance, the impedance of these circuits is relatively high. Therefore, both open tank circuits in the circuit configuration shown in Fig. 94A appear as relatively high impedance circuits for the input signal frequency of 950 MHz, while simultaneously appearing as relatively low impedance circuits for frequencies in the desired output signal frequency range of 50 MHz.

A power transfer signal 9442 controls a switch 9414. When the power transfer signal 9442 controls the switch 9414 to open and close it, high frequency signal components are not allowed to pass through the open circuit 1 or the open circuit 2. However, the low frequency signal components (50 MHz in this embodiment) generated by the system are allowed to pass through the open circuit 1 or the open circuit 2.

Tank 2, where they are slightly attenuated. The effect of Open Tank 1 and Open Tank 2 is to further separate the input and output signals from the same node to create a more stable input and output impedance. Capacitors 9418 and 9440 serve to store the energy of the 50 MHz output signal between the energy transfer pulses.

Further energy transfer optimization is provided by connecting an inductive element 9410 in series with a capacitor 9412, as shown. In the example shown, the series resonant frequency of the circuit arrangement is approximately 1 GHz. The circuit improves the energy transfer characteristics of the system. The ratio of the impedance of the inductive element 9410 to the impedance of the storage capacitor 9412 is preferably kept relatively small so that the majority of the available energy is transferred to the storage capacitor 9412 during operation. Exemplary output signals A and B are shown in Figures 94B and 94C, respectively.

In Fig. 94A, circuit components 9404 and 9406 form an input impedance matching system. Circuit components 9432 and 9430 form an output impedance matching system for a 50 Ω resistor 9428. Circuit components 9422 and 9424 form a second output impedance matching system for a 50 Ω resistor 9426. Capacitors 9408 and 9412 serve as storage capacitances for the embodiment. A voltage source 9446 and a resistor 9402 generate a 950 MHz signal with a 50 Ω output impedance that is used as an input to the circuit. A circuit element 9416 includes a 150 MHz oscillator and a pulse

generator used to generate the energy transfer signal 9442.

Fig. 88 shows an open shunt tank circuit 8810 in a single-ended-to-single-ended system 8812. Fig. 97 shows a similar open shunt tank circuit 9710 in a system 9712. The open tank circuits 8810 and 9710 reduce the driver source impedance, thereby improving the transient response. The open tank circuits 8810 and 9710 are able to store energy from the input signal and provide a low driver source impedance to transfer that energy during the entire time window that the switch is closed. The switching transition can be viewed as having its behavior influenced not only by the input signal frequency, but also by higher frequency components above the input signal frequency (i.e., frequencies higher than the input signal frequency are also able to pass through the switch during the time window defined by the pulse width). Resonant circuits or structures, e.g. the 8810 or 9710 resonant open tank circuits, can exploit this by being able to transfer energy throughout the switching transition (i.e., the capacitance in the resonant open tank circuit appears as a low driver source impedance during the transition period of the time window defined by the pulse width).

The exemplary open circuits and resonant structures described above are for illustrative purposes only and are not intended to limit the invention. Alternative configurations may be used. The various resonant open circuits and resonant structures discussed may be used in combination or independently.

5.6 Concepts for charge and energy transfer

Charge transfer concepts are described below with reference to Figures 109A-F. Figure 109A shows a circuit 10902 with a switch S and a capacitor 10906 with a capacitance value C. The switch S is controlled by a control signal 10908 comprising pulses 10910 with pulse widths T.

In Fig. 109B, GL(IO) represents that the charge q of the capacitance having a capacitance value C, e.g., the capacitance 10906, is proportional to the voltage V across the capacitance, where:

q = charge in coulombs

C = capacity in Farad

V = voltage in V

A = input signal amplitude

If the voltage V is represented by Eq. (11), Eq. (10) can be rewritten as Eq. (12). The charge change Aq over time t is represented in Eq. (13) by Aq(t), which can be rewritten as Eq. (14). Using the trigonometric sum-to-product identity of Eq. (15), Eq. (14) can be rewritten as Eq. (16), which can be rewritten as Eq. (17).

The sine term in Eq. (11) is a function of only the pulse width T. Therefore, Aq(t) is maximum when T is equal to an odd multiple of π (i.e. π, 3π, 5π, ...). Therefore, the capacitor 10906 experiences the largest change in charge when the pulse width T corresponds to a value of π or a time interval corresponding to a change of 180° of the sinusoidal

gen input signal. On the other hand, if T is equal to 2π, 4π, 6π, ..., minimal charge will be transferred.

Equations (18), (19) and (20) are solved for q(t) by integrating Equation (10) so that the charge on capacitor 10906 can be plotted with respect to time on the same axis as the sinusoidal input signal sin(t) as shown in the graph of Fig. 109C. As the pulse width T decreases or tends toward zero pulse, the phase between the charge on capacitor C or q(t) and sin(t) tends toward zero. This is shown in the graph of Fig. 109D which shows that the maximum pulse charge transfer occurs near the input voltage maxima. As shown in this graph, significantly less charge is transferred as T decreases.

Energy/charge relationships are shown in equations 21-26 of Figure 109E, where it is shown that the power or energy is proportional to the charge and the charge transferred is inversely proportional to the insertion loss.

Concepts of insertion loss are illustrated in Fig. 109F. In general, the noise figure of a lossy or attenuated passive element is numerically equal to the insertion loss of the element. Alternatively, the noise figure for any element cannot be less than its insertion loss. The insertion loss can be represented by Eq. (27) or Eq. (28).

From the above discussion, it is clear that as the pulse width T increases, more charge is transferred from the input signal to the capacitor 10906, thereby increasing the energy transfer from the input to the output. It has been shown that it is not necessary to exactly reproduce the input voltage at the output because relative modulation

lated amplitude and phase information is maintained in the transmitted energy.

5.7 Optimizing and adjusting the non-minor pulse width/duration (time window)

5.7.1 Changing the input and output impedances

In one embodiment of the invention, the energy transfer signal 6306 of Figure 63 is used to change the input impedance seen by the EM signal 1304 and change the output impedance through which a load is driven. An example of this embodiment will be described below using the gated transfer module 6404 shown in Figures 68G and 82A. The method described below is not limited to the gated transfer module 6404, but can be applied to all embodiments of the energy transfer module 6304.

In Fig. 82A, when switch 8206 is closed, the impedance of circuit 8202 is substantially equal to the impedance of a parallel circuit of the storage module represented by storage capacitance 8208 and the load 8212. When switch 8206 is open, the impedance at point 8214 approaches infinity. It follows that the average impedance at point 8214 can be varied from the impedance of the parallel circuit of the storage module represented by storage capacitance 8208 and the load 8212 to the highest impedance obtainable when switch 8206 is open by varying the ratio of the time period that switch 8206 is open to the time period that switch 8206 is closed. Because switch 8206 is controlled by energy transfer signal 8210

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controlled, the impedance at point 8214 can be varied by controlling the pulse width of the power transfer signal in conjunction with the aliasing rate.

An exemplary method for altering the energy transfer signal 6306 of FIG. 63 is described below with reference to FIG. 71, in which circuit 7102 receives the input oscillation signal 7106 and outputs a pulse train shown as doubler output signal 7104. Circuit 7102 may be used to generate the energy transfer signal 6306. Exemplary waveforms of signal 7104 are shown in FIG. 72B.

It can be shown that by varying the delay of the signal passing through inverter 7108, the width of the pulses in the doubler output signal 7104 can be varied. By increasing the delay of the signal passing through inverter 7108, the width of the pulses increases. The signal passing through inverter 7108 can be delayed by inserting an R/C low-pass network in the output of inverter 7108. Other means and methods for varying the delay of the signal passing through inverter 7108 are known to those skilled in the art.

5.7.2 Real-time pulse width control

In one embodiment, the pulse width/duration or time window is adjusted in real time. For example, as shown in the timing diagrams of Figures 98B-F, a clock signal 9814 (Figure 98B) is used to generate an energy transfer signal 9816 (Figure 98F) that includes energy transfer pulses 9818 with varying pulse widths 9820. In one embodiment, the clock signal 9814 is inverted, as shown by the inverted clock signal 9822 (Figure 98D). The clock signal 9814 is also

which is delayed as represented by delayed clock signal 9824 (Fig. 98E). Inverted clock signal 9814 and delayed clock signal 9824 are then ANDed together to produce an energy transfer signal 9816 that is active - energy transfer pulses 9818 - when delayed clock signal 9824 and inverted clock signal 9822 are both active. The delay imparted to delayed clock signal 9824 essentially determines the pulse width or duration or time window 9820. By changing the delay in real time, the pulse widths are adjusted in real time.

In an alternate implementation, the inverted clock signal 9822 is delayed with respect to the original clock signal 9814 and then ANDed with the original clock signal 9814. Alternatively, the original clock signal 9814 is delayed and then inverted, and the resulting signal is then ANDed with the original clock signal 9814.

98A shows an exemplary real-time pulse width control system 9802 suitable for adjusting pulse widths in real time. The exemplary real-time pulse width control system 9802 includes an RC circuit 9804 with a voltage-variable capacitance 9812 and a resistor 9826. The real-time pulse width control system 9802 also includes an inverter 9806 and an AND gate 9808. The AND gate 9808 optionally includes an enable input 9810 for enabling/disabling the AND gate 9808. The RC circuit 9804. The real-time pulse width control system 9802 optionally includes an amplifier 9828.

The operation of the real-time pulse width control circuit is described below with reference to the timing diagrams of Figures 98B-F. The real-time

Pulse width control system 9802 receives input clock signal 9814 which is applied to inverter 9806 and RC circuit 9804. Inverter 9806 outputs inverted clock signal 9822 and applies it to AND gate 9808. RC circuit 9804 delays clock signal 9814 and outputs delayed clock signal 9824. The delay is determined primarily by the capacitance value of voltage variable capacitance 9812. In general, the delay decreases as the capacitance decreases.

The delayed clock signal 9824 is optionally amplified by the optional amplifier 9828 before being fed to the AND gate 9808. Amplification is desired, for example, if the RC constant of the RC circuit 9804 attenuates the signal below the threshold of the AND gate 9808.

The AND gate 9808 ANDs the delayed clock signal 9824, the inverted clock signal 9822, and the optional enable signal 9810 to generate the energy transfer signal 9816. The pulse widths 9820 are adjusted by changing the voltage of the voltage-variable capacitor 9812.

In one embodiment, the pulse widths 9820 are controlled to optimize energy transfer. For example, in one embodiment, the pulse widths 9820 are controlled to optimize energy transfer. Alternatively, the pulse widths 9820 are controlled for variable gain control (e.g., for automatic gain control - AGC). In this embodiment, the energy transferred is reduced by reducing the pulse widths 9820.

As will be apparent from the present description, many of the illustrated time window or pulse width circuits and others may be modified in the manner described above (e.g., the circuits in the figures

guren 68H-K). The modification or selection of the time window or pulse width can be done at the design level so that it has a fixed value in the circuit, or in an alternative embodiment it can be dynamically adjusted to compensate for or take into account different design goals, e.g. the reception of RF signals with a higher efficiency that occur in different operating regions, e.g. RF signals at 900 MHz and 1.8 GHz.

5.8 Adding a redirection network

In one embodiment of the invention, a bypass network is added to improve the efficiency of the energy transfer module. Such a bypass network can be considered as a means for artificially broadening the pulse width. The components for a bypass network are selected such that the bypass network appears to have a substantially lower impedance for transitions of the switching module (i.e., at frequencies greater than that of the received EM signal) and a medium to high impedance for the EM input signal (e.g., greater than 100 Ω at the RF frequency).

The amount of time the input signal is now connected to the opposite side of the switching module is longer due to the shaping caused by this network, which in simple implementations can be a capacitance or a series resonant circuit of an inductive element and a capacitance. A network that is series resonant above the input frequency would be a typical implementation. This shaping improves the conversion efficiency of an input signal that would otherwise be poor if only the pulse width of the power transfer signal were considered.

would have a relatively low frequency and would therefore not be optimal.

95 shows an exemplary bypass network 9502 (shown in this case as capacitor 9512) that bypasses a switching module 9504. In this embodiment, the bypass network increases the efficiency of the energy transfer module if, for example, smaller than optimal pulse widths were chosen for a given input frequency for the energy transfer signal 9506. The bypass network 9502 could have a configuration other than that shown in FIG.

95. Such an alternative construction is shown in Fig. 90. Similarly, Fig.

96 another exemplary redirection network 9602 with a capacity 9604.

The effects of a minimized pulse width and the benefit obtained by the bypass network are discussed below. Figure 103 shows an initial circuit with a 550 ps time window, its output signal has an amplitude of 2.8 mVpp as shown in Figure 107A and is applied to a 50 Ω load. By changing the pulse width or time window to 270 ps as shown in Figure 104, a reduced output signal with an amplitude of 2.5 Vpp is obtained as shown in Figure 107B and is applied to a 50 Ω load. To compensate for this attenuation, a bypass network can be added; a specific embodiment of a bypass network is shown in Figure 105. With this added network, the 50 Ω load can now be supplied with a signal with an amplitude of 3.2 Vpp, as shown in Fig. 108A. The circuit with the bypass network in Fig. 105 also had three settings in the surrounding circuit to compensate for the voltage drop generated by the bypass network and the smaller

time window induced impedance changes. Figure 106 shows that these changes introduced into the circuit but without the bypass network did not result in the increased efficiency shown by the embodiment of Figure 105 with the bypass network. Figure 108B shows the result of using the circuit of Figure 106 in which only a signal with an amplitude of 1.88 Vpp could be applied to a 50 Ω load.

5.9 Modifying the energy transfer signal using feedback

69 shows an embodiment of a system 6901 having a down-converted signal 1308B as a feedback signal 6906 for controlling various characteristics of the energy transfer module 6304 to modify the down-converted signal 1308B.

In general, the amplitude of the down-converted signal 1308B changes as a function of the frequency and phase differences between the EM signal 1304 and the energy transfer signal 6306. In one embodiment, the down-converted signal 1308B is used as the feedback signal 6906 to control the frequency and phase relationship between the EM signal 1304 and the energy transfer signal 6306. This may be accomplished using the exemplary logic in Figure 85A. The exemplary circuit in Figure 85A may be located in the energy transfer signal module 6902. Those skilled in the art will appreciate, based on the present description, that alternative implementations are possible within the scope of the present invention. In this embodiment, a state machine is used as an example.

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In the example of Figure 85A, a state machine 8504 reads an A/D converter 8502 and controls a DAC 8506. In one embodiment, the state machine 8504 has two memory locations, Previous and Current, to store and retrieve the reading results of the A/D converter 8502. In one embodiment, the state machine 8504 uses at least one memory location.

The DAC 8506 controls an input of a voltage controlled oscillator 8508. The voltage controlled oscillator 8508 controls a frequency input of a pulse generator 8510, which in one embodiment is substantially similar to the pulse generator shown in Figure 68J. The pulse generator 8510 generates an energy transfer signal 6306.

In one embodiment, the state machine 8504 operates according to a state machine flow diagram 8519 shown in Figure 85B. The goal of this processing is to modify the frequency and phase relationship between the energy transfer signal 6306 and the EM signal 1304 to maintain the amplitude of the down-converted signal 1308B at substantially an optimal level.

The amplitude of the down-converted signal 1308B may be varied by the amplitude of the energy transfer signal 6306. In an embodiment where the switching module 6502 is a FET as shown in FIG. 66A, with the energy transfer signal 6306 provided to the gate electrode 6604, the amplitude of the energy transfer signal 6306 may determine the "on" resistance of the FET, which affects the amplitude of the down-converted signal 1308B. The energy transfer signal module 6902 shown in FIG. 85C may be an analog circuit that enables an automatic gain control function. Those skilled in the art will appreciate based on the present invention.

description that alternative implementations are possible within the scope of the present invention.

5.10 Other implementations

The embodiments described above are for illustration purposes only. These embodiments are not intended to limit the invention. It will be apparent to those skilled in the art from this description that other embodiments that differ slightly or substantially from those described herein can be implemented within the scope of the present invention.

6. Exemplary power transfer buck converters

Implementation examples are described below for illustrative purposes. The invention is not limited to these examples.

Figure 86 shows a schematic diagram of an exemplary circuit for downconverting a 915 MHz signal to a 5 MHz signal using a 101.1 MHz clock.

Figure 87 shows exemplary simulation waveforms for the circuit of Figure 86. Waveform 8602 is the input to the circuit and shows the noise caused by the switch closure. Waveform 8604 is the unfiltered output of the memory unit. Waveform 8606 is the impedance matched output of the buck converter at a different time scale.

Figure 88 shows a schematic diagram of an exemplary circuit for downconverting a 915 MHz signal to a 5 MHz signal using a 101.1 MHz clock.

The circuit also features an open circuit to improve conversion efficiency.

Figure 89 shows exemplary simulation waveforms for the circuit of Figure 88. Waveform 8802 is the input to the circuit and shows the noise caused by the switch closing operation. Waveform 8804 is the unfiltered output of the storage unit. Waveform 8806 is the output of the buck converter after the impedance matching circuit.

Figure 90 shows a schematic diagram of an exemplary circuit for down-converting a 915 MHz signal to a 5 MHz signal using a 101.1 MHz clock. The circuit includes a bypass circuit to improve conversion efficiency.

Figure 91 shows exemplary simulation waveforms for the circuit of Figure 90. Waveform 9002 is the input to the circuit and shows the noise caused by the switch closing operation. Waveform 9004 is the unfiltered output of the storage unit. Waveform 9006 is the output of the buck converter after the impedance matching circuit.

Figure 92 shows a schematic diagram of the exemplary circuit of Figure 86 connected to an FSK source that switches between 913 MHz and 917 MHz at a baud rate of 500 kBaud. Figure 93 shows the original FSK waveform 9202 and the down-converted waveform 9204 at the output of the load impedance matching circuit.

IV. Further embodiments

In this section, further aspects/embodiments of the invention are discussed.

In one embodiment of the present invention there is provided a method of transmitting information between a transmitter and a receiver, comprising the steps of: a) transmitting a first sequence of signals, each having a known period, at a first repetition rate from the transmitter; b) sampling each signal in the first sequence of signals once for a predetermined time interval by the receiver, the sampling of the first sequence of signals occurring at a second repetition rate that differs from the first repetition rate by a predetermined value; and c) generating an output signal by the receiver indicative of signal levels sampled in step b) and having a period longer than the predetermined period of a transmitted signal.

In another embodiment of the invention, a communication system is provided, comprising: a) a transmitter for transmitting a first sequence of signals, each having a known period, at a predetermined first repetition rate; b) a receiver for receiving the first sequence of signals, the receiver comprising sampling means for sampling the signal level of each signal in the first sequence of signals once for a predetermined time interval at a second repetition rate that differs from the first repetition rate by a predetermined value set by the receiver. The receiver comprises a first circuit for generating a first receiver output signal indicative of the sampled signal levels and having a period that is longer than a signal of the first sequence of signals. The transmitter comprises: an oscillator for generating an oscillator output signal at the first repetition rate, switching means for receiving the oscillator

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output signal and for selectively passing the oscillator output signal, a waveform generator for receiving the oscillator output signal to generate a waveform generator output signal having a time and a frequency range which are set by the waveform generator.

The embodiment of the invention described herein includes single or multi-user communications systems that use coherent signals to obtain better system performance than conventional radio frequency techniques while reducing cost and complexity. The design allows direct conversion of radio frequencies to baseband components for processing and enables highly efficient rejection of signals unrelated to a known or controlled slew rate between the transmitter and receiver clock oscillators. The system can be designed to exploit wideband techniques that further increase its reliability and enable higher user density within a given area. The technique used allows the system to be configured as a separate transmitter-receiver pair or as a transceiver.

It is a fundamental object of the present system to enable a new type of communication technology that is applicable to both narrowband and broadband systems. In its most general form, all the advantages of broadband communication are part of the system, so that, unlike in conventional broadband designs, no complicated and expensive circuits are required according to the invention. The communication system uses coherent signals to transmit and receive information and, in its simplest form, consists of a transmitter and a receiver.

receiver. The receiver includes circuitry that turns its radio frequency input on and off in a predetermined temporal relationship to the transmitted signal. This is made possible by allowing the transmitter's clock oscillator and the receiver's clock oscillator to operate at different but known frequencies to produce a known slew rate between the oscillators. If the slew rate is small compared to the clock oscillator frequencies, the transmitted waveform will appear temporally stable, i.e., coherent (moving at the predetermined slew rate), to the switched input of the receiver. The transmitted waveform is the only waveform that will appear temporally stable with respect to the receiver, so that the receiver's input signal can be averaged to filter out unwanted signals as desired. This methodology makes the system extremely selective without complicated filters and complex coding and decoding procedures and allows direct conversion of RF energy from an antenna or cable to baseband frequencies with a minimal number of standard components, further reducing cost and complexity. The transmitted waveform can be a constant carrier signal (narrowband), a controlled pulse signal (wideband or ultra-wideband), or a combination of both, e.g. a damped sinusoidal wave or any periodic waveform, so that the system can be designed to meet essentially all broadband requirements. Simple standard modulation and demodulation techniques, e.g. AM and pulse width modulation, can be easily used in the system.

Depending on the system requirements, e.g. the information transfer speed, the process gain, and the intended use, a

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suitable of several preferred embodiments of the invention may be selected. The embodiment discussed herein will be the amplitude and pulse width modulated system. It is one of the simplest implementations of the technology and has many components in common with subsequent systems. An amplitude modulated transmitter consists of a transmitter clock oscillator, a multiplier, a waveform generator, and an optional amplifier. The transmitter clock oscillator frequency may be determined by various resonant circuits, including, for example, an inductive element and a capacitor, a crystal resonator, a SAW resonator, or a crystal. The output waveform is sinusoidal, although identical system performance would be achieved by square wave oscillators.

The multiplier component multiplies the transmitter clock oscillator output by 0 or 1 or other constants, K1 and K2, to turn on and off the oscillator output supplied to the waveform generator. In this embodiment, the supplied information may be digital or analog data in the form of pulse width modulation. The multiplier allows the transmitter clock oscillator output to be present at the waveform generator input when the supplied information exceeds a predetermined value. In this condition, the transmitter will produce an output waveform. When the supplied information falls below a predetermined value, no input signal is supplied to the waveform generator so that no transmitter output waveform is produced. The output of the waveform generator determines the bandwidth of the system in the frequency domain and hence the number of users, the process gain immunity to interference and the overall reliability, the emission level at a

any given frequency and the antenna or cable requirements. The waveform generator in this example produces a single cycle pulse output signal that produces an ultra-wideband signal in the frequency domain. An optional amplifier stage amplifies the waveform generator output signal to a desired power level.

The amplitude and pulse width modulated transmitter of the present invention is shown in Figures 130 and 131 by reference numeral 13000. The transmitter clock oscillator 13002 is a crystal controlled oscillator operating at a frequency of 25 MHz. The multiplier 13004 includes a two-input NAND gate 13102 for gating the output of the oscillator 13002 to the waveform generator 13006. The waveform generator 13006 produces a pulse output signal shown in Figures 132D and 133 by reference numeral 13208 which produces a frequency spectrum shown in Figure 134 by reference numeral 13402. The amplifier 13008 is optional. The output signal of the transmitter 13000 is applied to an antenna or cable 13010 which, as is known in the art, may be constructed in a variety of ways to suit the circumstances.

Figures 132A-D and 133-134 show the various transmitter
13000 signals. The output signal of the transmitter 13000 in Fig. 132A can be either a sinusoidal or a
The gate 13102 may be a square wave signal 13202 which is applied to an input of the N-AND gate 13102. The gate 13102 also receives an information signal 13204 in Fig. 132B which is a digital signal in the illustrated embodiment. The output signal 13206 of the multiplier 13004 may be a sinusoidal or a square wave signal depending on the original signal 13202. The waveform generator 13006 produces a single cycle pulse signal 13208 at the output. The

Voltage of single cycle pulse 13210 varies by a static value 13212 and is generated at 40 ns intervals. In the illustrated embodiment, the frequency of transmitter 13002 is 25 MHz, so that during the entire time interval that gate 13102 is "on", single cycle pulses at a frequency of 1.0 GHz are transmitted and pass through the output of transmitter oscillator 13002 every 40 ns.

Figure 135 shows a block diagram of a preferred embodiment of a receiver for recovering the amplitude or pulse width modulated information, the receiver consisting of a receiver clock oscillator 13510, a waveform generator 13508, an RF switch/integrator 13506, a decoding circuit 13514, two optional amplifier/filter stages 13504 and 13512, an antenna or cable input 13502, and an information output 13516. The frequency of the receiver clock oscillator 13510 can be determined by various resonant circuits, including, for example, an inductive element and a capacitance, a ceramic resonator, a SAW resonator, or a quartz or crystal. As in the case of the transmitter, the oscillator 13510 shown herein is a quartz or crystal oscillator. The output waveform is a square wave, although identical system performance would be achieved by a sine wave oscillator. The square wave clock oscillator output signal 13602 is shown in Fig. 136A

The receiver clock oscillator 13510 is designed to operate within a frequency range that produces a known range of slew rates with respect to the transmitter clock oscillator 13002. In this embodiment, the frequency of the transmitter clock oscillator 13002 is 25 MHz and the receiver clock oscillator 13510 outputs a frequency between 25.0003 MHz

and 25.0012 MHz, producing a slew rate of + 300 to +1200 Hz.

The receiver clock oscillator 13510 is connected to the waveform generator 13508 which shapes the oscillator signal into an appropriate output signal to adjust the length of time the RF switch 13506 is on or off. The length of time the RF switch 13502 is on should be less than one-half cycle (1/10 cycle is preferred) or, in the case of a single pulse, no wider than the pulse width of the transmitted waveform, otherwise the signal gain of the system will decrease. Examples are shown in Table Al. Therefore, the output signal of the waveform generator 13508 is a pulse of an appropriate pulse width that occurs once per cycle of the receiver clock oscillator 13510. The output signal 13604 of the waveform generator is in
Fig. 136B.

Transmitted Waveform Gain Limit Preferred On Time

Switch-on time

Single 1 ns pulse 1ns 100 ps

1 GHz output signal with 1,2,3 ... etc. 500 ps 50 ps

Cycles 50 hp

10 GHz output signal with 1,2,3... etc. 5 ps
Cycles

Table Al

The RF switch/integrator 13506 samples the RF signal 13606 as shown in pig. 136C,
when the waveform generator output signal 13604 is less than a predetermined value. When the waveform generator output signal 13604 is greater than a predetermined

ner value, the RF switch 13506 becomes a high impedance node and allows the integrator to hold the last RF signal sample 13606 until the next cycle of the output of the waveform generator 13508. The integrator portion of the switch/integrator 13506 is designed so that the integrator is charged quickly (fast attack) and discharged at a controlled rate (slow decay). This embodiment provides rejection of unwanted signals and is a factor that determines the baseband frequency response of the system. The mode of operation of the switch control depends on the actual hardware implementation.

In one embodiment of the present invention, the gating or sampling rate of receiver 13500 is 300 Hz higher than the 25 MHz transmission rate from transmitter 13000. Alternatively, the sampling rate could be less than the transmission rate. The difference in repetition rates between transmitter 13000 and receiver 13500, i.e., the "slew rate," is 300 Hz and results in a controlled shift of the sampling pulses with respect to the transmitted pulses, which therefore appear "stable" in time to receiver 13500. Referring now to Figures 132A-D and 136A-G, an example of a simple case of an output signal 13608 (Figure 136D) is constructed from four samples of four RF input pulses 13606 for ease of explanation. As can be seen, by sampling the RF pulses 13605 that are passed when the transmitter information signal 13204 (Fig. 132B) is greater than a predetermined threshold, the signal 13608 is a copy of a signal 13606, but it is mapped to a different time base. In this example, the new time base has a period that is four times longer than the real-time signal. Using an op-

·· · It ·· W

297 _tons

tional amplifier/filter 13512 leads to a further refinement of the signal 13608, which is shown in Fig. 136E
shown as signal 13610.

A decoding circuit 13514 extracts the information contained in the transmit signal and includes a rectifier that rectifies the signal 13608 or 13610 to provide a signal 13612 shown in Fig. 136G. The variable threshold generator circuit included in circuit 13514 provides a DC threshold signal level 13614 for the signal 13610 that is used to determine a HIGH (transmitter output signal ON) or LOW (transmitter output signal OFF) state and is shown in Fig. 136G. The

The final output signal 13616 is generated by an output voltage comparator in circuit 13514 which combines signals 13612 and 13614 such that when signal 13612 has a higher voltage value than signal 13614, the information output signal switches to the HIGH state. Thus, for example, signal 13616 represents a digital "1", now with respect to a time base stretched in a ratio of 1:4 with respect to the period of an original signal 13606. Although a frequency reduction of 4:1 is obtained by this representation, it is sometimes desirable to provide a reduction of more than 50,000:1; in the preferred embodiment, a reduction of 100,000:1 or more is achieved. This provides a direct shift from the RF input frequency to a low frequency band without the need for expensive intermediate frequency circuitry that would have to be used if only a 4:1 conversion or translation were used in a first stage. Information regarding time base conversion and examples are given in Table A2.

Units:

Is = 1; 1 ps = 1-1(G ¹² ; 1 ns = 1·10~ ⁹ ; 1 με = 1·10~ ⁶ ;

1 MHz = I-&Igr;&Ogr;⁶; 1 kHz = 1 10 ³

Receiver clock oscillator frequency = 25.00003 MHz Transmitter clock oscillator frequency = 25 MHz

Period = -

Transmitter - clock oscillator frequency

Period = 40 ns

Slew rate = -

(Receiver - clock oscillator frequency) - (Transmitter - clock oscillator frequency)

Slew rate = 0.003 s

Time base multiplier = seconds per nanosecond

period

Time base multiplier = 8.333·10 ⁴ Example 1 :

1 ns is converted into 83.33 \is

Time base = (1 ns)■Time base multiplier Time base = 83.33 με

Example 2 :

2 GHz is converted into 24 kHz, 2 GHz = 500 ps period
Timebase = (500 ps)·Timebase multiplier

Frequency =

Time base

Frequency = 24 kHz

Table &Aacgr;2

In the preferred embodiment shown, the signal 13616 shown at "F" has a period of 83.33 ps, a frequency of 12 KHz, and is generated once every 3.3 ms for a slew rate of 300 Hz. That is, the system converts a 1 GHz transmit signal into an 83.33 μβ-λίδα signal.

Therefore, the train of RF pulses 13210 transmitted during the presence of an "ON" signal at the information input gating circuit 13102 is used to reproduce the information input signal 13204 by sampling the pulse train at the receiver 13500. The system is designed to generate an appropriate number of RF input signals 13606 to enable signal reconstruction.

An optional amplifier/filter stage (or stages) 13504 and 13512 may be used to provide better receiver sensitivity, bandwidth control, or signal conditioning for the decoding circuit 13514. By selecting an appropriate time base multiplier, a signal is obtained at the output of the integrator 13506 that can be amplified and filtered by an operational amplifier instead of RF amplifiers, thus simplifying the design process. The signal 13610 shown at "E" illustrates the use of the amplifier.

kers/Filters 13512 (Fig. 137). The optional RF amplifier 13504, shown as the first stage of the receiver, should be included in the design if higher sensitivity and/or additional filtering processing is required. Figures 137-139 show schematic examples of receivers.

Figures 140-143 show various pulse output signals 14002 and 14202 and their corresponding frequency range at 14102 and 14302. As can be seen in Figures 140 and 141, the half-cycle signal 14002 produces a spectrum that is less affected by interference than the one-cycle pulse of Figure 133 and the 10-cycle pulse of Figure 142. The various output signals determine the system immunity to interference, the number of users in a given area, and the cable and antenna requirements. Figures 133 and 134 show exemplary pulse output signals.

Figures 144 and 145 show exemplary differential receiver designs. The operation is theoretically similar to that of the non-differential receiver of Figure 135, except that a higher signal-to-noise ratio is obtained by the differential technique due to in-phase rejection. Any signal induced in phase at both inputs of the differential receiver is attenuated by the differential amplifier shown in Figures 144 and 145, and conversely any signal producing a phase difference between the two receiver inputs is amplified.

Figures 146 and 147 show time and frequency domains of a narrowband constant carrier signal compared to ultra-wideband signals used in the illustrated embodiment.

·

Claims (158)

1. A device for down-converting an electromagnetic (EM) signal comprising: 1. a device for receiving an EM signal; and 2. means for aliasing the EM signal according to an aliasing signal at an aliasing rate to down-convert the EM signal, comprising the step of transferring energy from the EM signal at the aliasing rate. 2. The device of claim 1, further comprising: 1. a device for generating the aliasing signal, wherein the aliasing signal has a plurality of pulses with non-minor pulse widths. 3. The apparatus of claim 1, wherein the EM signal is down-converted into a demodulated baseband information signal. 4. The apparatus of claim 1, wherein the EM signal is down-converted to an intermediate frequency (IF) signal. 5. The apparatus of claim 1, wherein the EM signal is a frequency modulated (FM) signal and the EM signal is down-converted to a non-FM signal. 6. The apparatus of claim 5, wherein the non-FM signal is a phase modulated (PM) signal. 7. The apparatus of claim 5, wherein the non-FM signal is an amplitude modulated (AM) signal. 8. Device for directly down-converting a modulated carrier signal into a demodulated baseband signal, comprising: 1. a device for receiving the modulated carrier signal; and 2. means for transmitting non-negligible amounts of energy from the modulated carrier signal at a rate substantially equal to a subharmonic of the modulated carrier signal, such that the demodulated baseband signal is formed from the transmitted non-negligible amounts of energy. 9. The apparatus of claim 8, including means for generating an energy transfer signal having a harmonic substantially equal to a frequency of the modulated carrier signal, and using the asynchronous energy transfer signal to transfer the energy from the modulated carrier signal. 10. Apparatus according to claim 9 comprising means for generating a sequence of pulses with non-negligible pulse widths whose duration tends away from zero. 11. Apparatus according to claim 9 comprising means for optimizing a phase of the energy transfer signal. 12. Apparatus according to claim 9 comprising means for optimizing the frequency of the energy transfer signal. 13. Apparatus according to claim 8 comprising means for receiving an amplitude modulated carrier signal. 14. Apparatus according to claim 8 comprising means for receiving a phase-modulated carrier signal. 15. Apparatus according to claim 8 comprising means for receiving a modulated carrier signal having a frequency of more than 1 GHz. 16. Apparatus according to claim 8 comprising means for receiving a modulated carrier signal having a frequency between 10 MHz and 10 GHz. 17. Apparatus according to claim 8, comprising means for receiving the modulated carrier signal via a path having a relatively low input impedance. 18. Apparatus according to claim 8 comprising means for receiving the modulated carrier signal over a path having a relatively efficient energy transfer. 19. Apparatus according to claim 8 comprising means for receiving the modulated carrier signal via a substantially impedance matched input path. 20. Apparatus according to claim 8, comprising means for directly supplying the demodulated baseband signal to a load having a relatively low impedance. 21. Apparatus according to claim 8, comprising means for supplying the demodulated baseband signal directly to a load via a path having a relatively efficient power transfer. 22. Apparatus according to claim 8, comprising means for supplying the demodulated baseband signal directly to a load via a substantially impedance matched path. 23. The apparatus of claim 8 including means for coupling the modulated carrier signal to a reactive storage device at a rate substantially equal to a subharmonic of the modulated carrier signal. 24. Apparatus according to claim 8, comprising means for generating the energy transfer signal without synchronizing it with the modulated carrier signal. 25. Apparatus according to claim 8, comprising means for generating the energy transfer signal without synchronizing it with a phase of the modulated carrier signal. 26. Apparatus according to claim 8 comprising means for generating the energy transfer signal independently of the modulated carrier signal. 27. Apparatus according to claim 8 comprising means for generating an asynchronous energy transfer signal. 28. Apparatus according to claim 27, comprising means for receiving the modulated carrier signal over a path having a relatively low input impedance. 29. Apparatus according to claim 27, comprising means for receiving the modulated carrier signal over a path having a relatively efficient energy transfer. 30. Apparatus for directly down-converting a modulated carrier signal into a demodulated baseband signal, comprising:
a power transfer signal generator;
a switching module controlled by the energy transfer signal generator;
a memory module connected to the switching module; and
a circuit for efficiently transferring energy from the modulated carrier signal. 31. The device of claim 30, wherein the circuit comprises an input impedance matching circuit connected to an input of the device. 32. The apparatus of claim 30, wherein the circuit comprises an output impedance matching circuit connected to an output of the device. 33. The device of claim 30, wherein the switching module is connected between an input of the device and the memory module. 34. The device of claim 30, wherein the circuit comprises an open circuit connected between an input of the device and the switching circuit. 35. The apparatus of claim 30, wherein the circuit comprises a feed-forward circuit connected in parallel with the switching module. 36. The device of claim 30, wherein the memory module is connected between an input of the device and the switching module. 37. The device of claim 30, wherein the device is tuned for at least one frequency greater than 1 GHz. 38. The device of claim 30, wherein the device is tuned for at least one frequency between 10 MHz and 10 GHz. 39. The apparatus of claim 30, wherein the energy transfer signal generator comprises an asynchronous energy transfer signal generator. 40. A system for down-converting an electromagnetic (EM) signal, comprising: means for integrating and transferring energy from an EM signal to down-convert the EM signal, the means comprising a resonant structure for enhancing the energy transfer. 41. A system for down-converting an electromagnetic (EM) signal, comprising: means for integrating and transferring energy from an EM signal to down-convert the EM signal, the means having an optimized switching structure. 42. Apparatus for down-converting and demodulating a modulated carrier signal ( 1304 ) into a baseband signal ( 1308B ) comprising: 1. means for receiving ( 4614 ) the modulated carrier signal ( 1304 ); 2. means for storing ( 4618 ) energy from the received carrier signal ( 1304 ) over pulse widths or pulse windows ( 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) at an aliasing rate determined according to a frequency of the received carrier signal ( 1304 ) divided by an integer; and 3. means for generating ( 4618 ) the baseband signal ( 1308 ) from the stored energy. 43. Apparatus for down-converting a modulated carrier signal ( 1304 ) into a lower frequency modulated carrier signal ( 1308B ) comprising: 1. means for receiving (46081) the modulated carrier signal ( 1304 ); 2. means for storing ( 4612 ) energy from the received carrier signal ( 1304 ) over pulse widths or pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ) with an aliasing rate determined according to:
(a frequency of the modulated carrier signal ( 1304 ) plus or minus a frequency of the lower frequency modulated carrier signal ( 4708 ; 5016 ; 5116 ; 5216 ; 5316 ; 5416 ; 5516 ))/n,
where n is an integer; and 3. means for generating ( 4612 ) the lower frequency modulated carrier signal from the stored energy. 44. Apparatus according to claim 42 or 43, comprising: means for generating an energy transfer signal ( 4702 ; 5006 ; 5106 ; 5206 ; 5306 ; 5406 ; 5506 ; 5602 ; 5706 ; 5806 ; 5906 ; 6006 ) having a pulse train ( 4704 ; 5007 ; 5107 ; 5207 ; 5307 ; 5407 ; 5507 ; 5604 ; 5707 ; 5807 ; 5907 ; 6007 ) having the aliasing rate with non-minor pulse widths ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) and using the energy transfer signal ( 4702 ; 5006 ; 5106 ; 5206 ; 5306 ; 5406 ; 5506 ; 5602 ; 5706 ; 5806 ; 5906 ; 6006 ) to transfer the energy from the received carrier signal ( 1304 ). 45. Apparatus according to any preceding claim, comprising: means for coupling the received carrier signal ( 1304 ) at the aliasing rate to a reactive memory device ( 6801 ). 46. Apparatus according to claim 42 or 43, comprising: means for gating the received carrier signal ( 1304 ) according to a control signal ( 6306 ) comprising a sequence of pulses ( 4704 ; 5007 ; 5107 ; 5207 ; 5307 ; 5407 ; 5507 ; 5604 ; 5707 ; 5807 ; 5907 ; 6007 ) with non-minor pulse widths or pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) extending Repeat with the aliasing rate. 47. Device according to claim 42 or 43, comprising: a) means for repeating the transmission ( 4606 ) several times at the aliasing rate of a non-insignificant energy component contained in a portion of the received carrier signal ( 1304 ). 48. Apparatus according to any preceding claim, comprising means for controlling a frequency relationship between the received carrier signal ( 1304 ) and the aliasing rate. 49. Apparatus according to any one of claims 42 to 47, comprising means for controlling a phase relationship between the received carrier signal ( 1304 ) and the aliasing rate. 50. Apparatus according to any one of claims 42 to 47, comprising means for controlling a frequency relationship and a phase relationship between the received carrier signal ( 1304 ) and the aliasing rate. 51. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are approximately equal in duration to half periods of the received carrier signal ( 1304 ). 52. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately at least one tenth of a percent of half periods of the received carrier signal (1304 ) . 53. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately one percent of half periods of the received carrier signal (1304 ) . 54. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater than approximately ten percent of half periods of the received carrier signal (1304 ) with respect to their duration. 55. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately twenty- five percent of half periods of the received carrier signal (1304 ) . 56. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater than approximately fifty percent of half periods of the received carrier signal (1304 ) with respect to their duration. 57. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately seventy -five percent of half periods of the received carrier signal (1304 ) . 58. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately ninety percent of half periods of the received carrier signal (1304 ) . 59. Apparatus according to any one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal in duration to approximately N periods of the received carrier signal ( 1304 ), where N is A plus B, where A is a positive integer and B is a fraction of an integer. 60. Apparatus according to any one of claims 42 to 59, comprising means for transmitting non-trivial amounts of energy with respect to the energy contained in half-periods of the received carrier signal ( 1304 ). 61. Device according to claim 42 or 43 with a device for generating a sequence of pulses ( 4704 ; 5007 ; 5107 ; 5207 ; 5307; 5407 ; 5507 ; 5604 ; 5707 ; 5807 ; 5907 ; 6007 ) with non-minor pulse widths with respect to a period of the received carrier signal ( 1304 ). 62. Apparatus according to claim 47, comprising means for transmitting at least one-tenth of a percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 63. Apparatus according to claim 47 comprising means for transmitting at least one percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 64. Apparatus according to claim 47, comprising means for transmitting at least ten percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 65. Apparatus according to claim 47, comprising means for transmitting at least twenty-five percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 66. Apparatus according to claim 47, comprising means for transmitting at least fifty percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 67. Apparatus according to claim 47, comprising means for transmitting at least seventy-five percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 68. Apparatus according to claim 47, comprising means for transmitting at least eighty percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 69. Apparatus according to claim 47, comprising means for transmitting at least ninety percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 70. Apparatus according to any one of claims 42 to 47, wherein the pulse windows ( 5809 ) are approximately equal in duration to half periods of the received carrier signal ( 616 ) and are substantially aligned with periods of the received carrier signal ( 616 ). 71. The apparatus of claim 70, wherein the pulse widths or pulse windows ( 5809 ) are substantially aligned with positive half periods of the received carrier signal ( 616 ). 72. The apparatus of claim 70, wherein the pulse widths or pulse windows ( 6009 ) are substantially aligned with negative half periods of the received carrier signal ( 1016 ). 73. Device according to one of claims 42 to 47, wherein the pulse windows ( 5009 ) are approximately equal in duration to half periods of the received carrier signal ( 516 ) and are not aligned with the received carrier signal ( 516 ). 74. Device according to one of claims 42 to 47, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are substantially synchronized with the received carrier signal ( 1304 ). 75. Apparatus according to any one of claims 42 to 47, wherein the pulse windows ( 6306 ) are not synchronized with the received carrier signal ( 1304 ). 76. Apparatus according to claim 44 comprising means for generating the energy transfer signal ( 6306 ) independently of the received carrier signal ( 1304 ). 77. Apparatus according to any one of claims 42 to 47, comprising means for generating an asynchronous energy transfer signal ( 6306 ) having an aliasing rate, and means for using the asynchronous energy transfer signal ( 6306 ) to store the energy from the received carrier signal ( 1304 ). 78. Apparatus according to claim 42 or any of claims 47 to 77 as dependent thereon, comprising means for saving modulation information without reproducing the received carrier signal ( 1304 ). 79. Device according to one of claims 42 to 47 with a device for differential storage of energy. 80. Apparatus according to any one of claims 42 to 47, comprising means for controlling the amount of energy received from the received carrier signal ( 1304 ). 81. Apparatus according to claim 80 comprising means for controlling a pulse window ( 9820 ). 82. Apparatus according to claim 81 comprising means for controlling a pulse width. 83. Apparatus according to claim 81 comprising means for controlling a pulse window frequency. 84. Apparatus according to claim 81 comprising means for controlling a pulse window shape. 85. Apparatus according to claim 80 comprising means for controlling an energy transfer characteristic. 86. Apparatus according to any preceding claim, comprising means for discharging the stored energy between pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) in a controlled manner. 87. Apparatus according to any preceding claim, comprising means for storing sufficient energy from the received carrier signal ( 8204 ) to drive a load ( 8212 ) without additional buffering or amplification, the load being a high impedance load or a low impedance load ( 8218 ). 88. Apparatus according to any preceding claim comprising means for storing energy from the received carrier signal ( 8204 ) without pre-amplification (LNA) of the received carrier signal ( 8204 ). 89. Apparatus according to any preceding claim, comprising means for storing energy from the received carrier signal ( 1304 ) preventing exact voltage reproduction of the received carrier signal ( 1304 ) during the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ). 90. Apparatus according to any preceding claim, comprising means for receiving the modulated carrier signal ( 1304 ) via a path ( 7006 ) having a relatively low input impedance. 91. Apparatus according to any preceding claim, comprising means for receiving the modulated carrier signal ( 1304 ) via a path ( 7006 ) having a relatively efficient energy transfer. 92. Apparatus according to any preceding claim, comprising means for gating the received carrier signal ( 1304 ) using a switch ( 6502 ). 93. The apparatus of claim 92, wherein the impedance of the switch ( 7016 ) is matched to a source impedance ( 7012 ), thereby increasing the energy transmitted by the received carrier signal ( 1304 ). 94. The apparatus of claim 92, wherein the impedance of the switch ( 7016 ) is matched to a load impedance ( 7018 ), thereby increasing the energy transmitted by the received carrier signal ( 1304 ). 95. The apparatus of claim 94, wherein the received carrier signal ( 1304 ) is coupled to the switch ( 9414 ) via a resonant circuit ( 9438 , 9420 ), the resonant circuit ( 9438 , 9420 ) storing energy from components of the received carrier signal ( 1304 ) while the switch ( 9414 ) is open, and wherein energy stored in the resonant circuit ( 9438 , 9420 ) is discharged via the switch ( 9414 ) while the switch ( 9414 ) is closed, thereby increasing the energy carried by the received carrier signal ( 1304 ). 96. Apparatus for down-converting and demodulating a modulated carrier signal ( 1304 ) into a baseband signal ( 1308B ) comprising: 1. means for receiving the modulated carrier signal ( 1304 ); 2. means ( 6304 ) for storing energy from the received carrier signal ( 1304 ) over pulse widths ( 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) at an aliasing rate determined according to a frequency of the received carrier signal ( 1304 ) divided by an integer; and 3. means ( 6304 ) for generating the baseband signal ( 1308 ) from the stored energy. 97. Apparatus for down-converting a modulated carrier signal ( 1304 ) into a lower frequency modulated carrier signal ( 1308B ) comprising: 1. a device for receiving the modulated carrier signal; 2. means ( 6304 ) for storing energy from the received carrier signal ( 1304 ) over pulse widths ( 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) with an aliasing rate determined according to:
(a frequency of the modulated carrier signal ( 1304 ) plus or minus a frequency of the lower frequency modulated carrier signal (130%))/n,
where n is an integer; and 3. means ( 6304 ) for generating the lower frequency modulated carrier signal ( 1304 ) from the stored energy. 98. Apparatus according to claim 96 or 97, wherein the means ( 6304 ) for storing energy comprises: means for generating an energy transfer signal ( 6306 ) comprising a pulse train ( 4704 ; 5007 ; 5107 ; 5207 ; 5307 ; 5407 ; 5507 ; 5604 ; 5707 ; 5807 ; 5907 ; 6007 ) at the aliasing rate with non-minor pulse widths ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ), and using the energy transfer signal to transfer the energy from the received carrier signal ( 5012 ). 99. The apparatus of claim 96 or 97, wherein the means for storing energy comprises means for coupling the received carrier signal ( 1304 ) to a reactive storage device ( 6801 ) at the aliasing rate. 100. Apparatus according to claim 96 or 97, wherein the means for storing energy comprises: means ( 6404 ) for gating the received carrier signal ( 1304 ) according to a control signal ( 6306 ) comprising a sequence of pulses ( 4704 ; 5007 ; 5107 ; 5207 ; 5307 ; 5407 ; 5507 ; 5604 ; 5707 ; 5807 ; 5907 ; 6007 ) with non-minor pulse widths ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) that repeat at the aliasing rate. 101. The apparatus of claim 96 or 97, wherein the means for storing energy comprises: means for transmitting a non-de minimis energy content contained in portions of the received carrier signal ( 1304 ) at the aliasing rate. 102. The apparatus of any of claims 96 to 101, wherein the means for storing energy comprises means ( 8504 ) for controlling a frequency relationship between the received carrier signal ( 1304 ) and the aliasing rate. 103. The apparatus of any of claims 96 to 101, wherein the means for storing energy comprises means ( 8504 ) for controlling a phase relationship between the received carrier signal ( 1304 ) and the aliasing rate. 104. The apparatus of any of claims 96 to 101, wherein the means for storing energy comprises means ( 8504 ) for controlling a frequency relationship and a phase relationship between the received carrier signal ( 1304 ) and the aliasing rate. 105. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are approximately equal in duration to half periods of the received carrier signal ( 1304 ). 106. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately one tenth of a percent of half periods of the received carrier signal ( 1304 ). 107. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately one percent of half periods of the received carrier signal (1304 ) . 108. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater than approximately ten percent of half periods of the received carrier signal (1304 ) with respect to their duration. 109. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 5009 ) are equal to or greater in duration than approximately twenty- five percent of half periods of the received carrier signal (1304 ) . 110. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately fifty percent of half periods of the received carrier signal (1304 ) . 111. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately seventy -five percent of half periods of the received carrier signal (1304 ) . 112. Device according to one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal to or greater in duration than approximately ninety percent of half periods of the received carrier signal (1304 ) . 113. Apparatus according to any one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are equal in duration to approximately N periods of the received carrier signal ( 1304 ), where N is equal to A plus B, where A is a positive integer and B is a fraction of an integer. 114. Apparatus according to any one of claims 96 to 113, wherein the means for storing energy comprises means for transmitting non-trivial amounts of energy with respect to the energy contained in half-periods of the received carrier signal ( 1304 ). 115. Apparatus according to any one of claims 96 to 101, wherein the means for storing energy comprises means for generating a sequence of pulses ( 4704 ; 5007 ; 5107 ; 5207 ; 5307 ; 5407 ; 5507 ; 5604 ; 5707 ; 5807 ; 5907 ; 6007 ) having non-insignificant pulse widths ( 4701 ; 5009 ; 5109 ; 5209; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) with respect to a period of the received carrier signal ( 1304 ). 116. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least one-tenth of a percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 117. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least one percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 118. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least ten percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 119. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least twenty-five percent of the energy contained in a half-cycle of the received carrier signal ( 1304 ). 120. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least fifty percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 121. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least seventy-five percent of the energy contained in a half-cycle of the received carrier signal ( 1304 ). 122. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least eighty percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 123. The apparatus of claim 101, wherein the means for transmitting non-de minimis amounts of energy comprises means for transmitting at least ninety percent of the energy contained in a half-period of the received carrier signal ( 1304 ). 124. The apparatus of any of claims 96 to 101, wherein the pulse windows ( 5809 ) are approximately equal in duration to half periods of the received carrier signal ( 616 ) and are substantially aligned with periods of the received carrier signal ( 616 ). 125. The apparatus of claim 124, wherein the pulse windows ( 5809 ) are substantially aligned with positive half periods of the received carrier signal ( 616 ). 126. The apparatus of claim 124, wherein the pulse windows ( 6009 ) are substantially aligned with negative half periods of the received carrier signal ( 1016 ). 127. Apparatus according to any one of claims 96 to 101, wherein the pulse windows ( 5009 ) are approximately equal in duration to half periods of the received carrier signal ( 516 ) and are not aligned with the received carrier signal ( 516 ). 128. Apparatus according to any one of claims 96 to 101, wherein the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) are substantially synchronized with the received carrier signal ( 1304 ). 129. Apparatus according to any one of claims 96 to 101, wherein the pulse windows ( 6306 ) are not synchronized with the received carrier signal ( 1304 ). 130. The apparatus of claim 98, wherein the means ( 6304 ) for storing energy further comprises means for generating the energy transfer signal ( 6306 ) independently of the received carrier signal ( 1304 ). 131. The apparatus of any of claims 96 to 101, wherein the means ( 6304 ) for storing energy comprises means for generating an asynchronous energy transfer signal ( 6306 ) having the aliasing rate and for using the asynchronous energy transfer signal ( 6306 ) to store the energy from the received carrier signal ( 1304 ). 132. Apparatus according to claim 96 or any of claims 57 to 90 as dependent thereon, wherein the generating means ( 6304 ) comprises means for saving modulation information without reproducing the received carrier signal ( 1304 ). 133. The apparatus of any of claims 96 to 101, wherein the means for storing energy comprises a first means ( 7612 ) for storing energy and a second means ( 7618 ) for storing energy that are differentially configured. 134. Apparatus according to any one of claims 96 to 101, wherein the means for storing energy comprises means ( 9802 ) for controlling the amount of energy transmitted by the received carrier signal ( 1304 ). 135. Apparatus according to claim 134, wherein the means ( 9802 ) for controlling the energy transmitted by the received carrier signal comprises means for controlling the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ). 136. The apparatus of claim 135, wherein the means ( 9802 ) for controlling the pulse windows comprises means for controlling a pulse width. 137. The apparatus of claim 135, wherein the means ( 9802 ) for controlling the pulse windows comprises means ( 6902 ) for controlling a pulse window frequency. 138. The apparatus of claim 135, wherein the means ( 9802 ) for controlling the pulse windows comprises means (12502) for controlling a pulse window frequency. 139. The apparatus of claim 134, wherein the means ( 9802 ) for controlling the amount of energy transferred by the received carrier signal ( 1302 ) comprises means for controlling an energy transfer characteristic. 140. Apparatus according to any one of claims 96 to 139, wherein the generating means ( 6304 ) comprises means for discharging the stored energy between pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ) in a controlled manner. 141. The apparatus of any of claims 96 to 140, wherein the means ( 8208 ) for storing energy is capable of storing sufficient energy from the received carrier signal ( 8204 ) to drive a load ( 8212 ) without additional buffering or amplification, the load being a high impedance load or a low impedance load ( 8218 ). 142. Apparatus according to any one of claims 96 to 141, wherein the means ( 8208 ) for storing energy is capable of storing energy from the received carrier signal ( 8204 ) without pre-amplification (LNA) of the received carrier signal ( 8204 ). 143. Apparatus according to any one of claims 96 to 142, wherein the means ( 8310 ) for storing energy from the received carrier signal ( 8308 ) substantially prevents exact voltage reproduction of the received carrier signal ( 8308 ) during the pulse windows ( 4701 ; 5009 ; 5109 ; 5209 ; 5309 ; 5409 ; 5509 ; 5606 ; 5709 ; 5809 ; 5909 ; 6009 ). 144. Apparatus according to any one of claims 96 to 143, wherein the means for receiving the modulated carrier signal is arranged to receive the modulated carrier signal ( 1304 ) via a path ( 7006 ) having a relatively low input impedance. 145. Apparatus according to any one of claims 96 to 144, wherein the means for receiving the modulated carrier signal is arranged to receive the modulated carrier signal ( 1304 ) via a path ( 7006 ) having a relatively efficient energy transfer. 146. The apparatus of any of claims 96 to 145, wherein the means for storing energy comprises means for gating the received carrier signal ( 1304 ) using a switch ( 6502 ). 147. The apparatus of claim 146, wherein the means for storing energy further comprises means ( 7006 ) for matching an impedance of the switch ( 7016 ) to a source impedance ( 7012 ), thereby increasing the energy transferred by the received carrier signal ( 1304 ). 148. The apparatus of claim 146, wherein the means for storing energy further comprises means ( 7008 ) for matching an impedance of the switch ( 7016 ) to a load impedance ( 7018 ), thereby increasing the energy carried by the received carrier signal ( 1304 ). 149. The apparatus of claim 146, further comprising: means for coupling the received carrier signal ( 1304 ) to the switch ( 9414 ) via a resonant circuit ( 9438 , 9420 ), wherein the resonant circuit ( 9438 , 9420 ) stores energy from components of the received carrier signal ( 1304 ) while the switch ( 9414 ) is open, and wherein energy stored in the resonant circuit ( 9438 , 9420 ) is discharged via the switch ( 9414 ) while the switch ( 9414 ) is closed, thereby increasing the energy carried by the received carrier signal ( 1304 ). 150. The apparatus of any of claims 96 to 101, further comprising: an input impedance matching circuit ( 7006 ). 151. The apparatus of any of claims 96 to 101, further comprising: an output impedance matching circuit ( 7008 ). 152. The apparatus of any of claims 96 to 101, further comprising: an open circuit ( 7716 ). 153. The apparatus of claim 146, further comprising: a feed-forward circuit ( 9502 ) connected in parallel with the switch ( 9504 ). 154. The apparatus of any of claims 96 to 101, further comprising: a feedback loop ( 6906 ). 155. The apparatus of any of claims 96 to 101, wherein the means for storing energy comprises: a capacitive storage device ( 8208 ) sized to store substantial amounts of energy relative to the energy approximately contained in one half period of the received carrier signal ( 8204 ). 156. The device of any of claims 96 to 101, wherein the means for storing energy is implemented from a complementary metal oxide semiconductor (CMOS) material (12426). 157. The apparatus of any of claims 96 to 101, wherein the means for storing energy is implemented as a single integrated circuit ( 12426 ). 158. The apparatus of any of claims 96 to 101, further comprising a controllable output impedance ( 9802 ).