DE102007030172A1 - Electricity meter with high dynamics and small time constant - Google Patents

Abstract

The invention provides two different ammeters are available, which are based in the core each on a wired as a current-voltage converter operational amplifier. Their particular advantages over generic devices receive devices according to the invention in each case by a targeted modification of the circuit resistance in the feedback path. In the device according to the main claim ("device 1"), the Beschaltungswiderstand changes depending on the current to be measured. Device 1 is thus a linear amplifier with automatic, quasi-instantaneous measuring range switching, optionally also with multiple output channels of different sensitivity, which can be recorded simultaneously. In the device according to the independent claim ("device 2"), the circuit resistance comprises at least two biased diodes. Device 2 is thus a logarithmic amplifier with adjustable by the bias sensitivity. The advantages of the devices 1 and 2 can be intercombined arbitrarily due to their technical relationship in a device.

Description

The The invention relates to a current measuring device with high dynamics and small time constant.

State of the art

experiments charge transport in molecular metal | molecule | metal Bridges are characterized by current changes several orders of magnitude in a short time. measurements the corresponding currents (or the molecular conductivity) require high dynamics and fast response times. For measurements small currents (<1 Picoampere) have so far been linear current-voltage amplifiers used, which has a low dynamics (5 to 6 orders of magnitude) exhibit.

A large dynamic range for current-voltage measurements could be realized with a logarithmic preamplifier ( YB Acharya, SG Tikekar., Review of Scientific Instruments 64, p. 1652, 1993 ), which was later also used for conductivity measurements in scanning tunneling microscope configurations ( U. Dürig, L. Novotny, B. Michel, A. Stadler, Review of Scientific Instruments 68, page 3814, 1997; J. He, O. Sankey, M. Lee, NJ Tao, XL Li, S. Lindsay, Faraday Discussions 131, page 145, 2006 ). The logarithmic gain is realized by two antiparallel diodes as elements in the feedback of a current-voltage amplifier ( YB Acharya, SG Tikekar., Review of Scientific Instruments 64, p. 1652, 1993; U. Dürig, L. Novotny, B. Michel, A. Stadler, Review of Scientific Instruments 68, page 3814, 1997 ). The disadvantage of the high dynamics in this solution is the disadvantage compared to the low accuracy of the measurement for errors of up to 20%. To reduce the remaining error, exact temperature compensation or temperature stabilization is necessary, the practical implementation of which is not trivial. In addition, low-capacitance, low-leakage diodes do not exhibit an exact exponential current-voltage characteristic, which further complicates the calibration process.

In addition, the diodes show a high dynamic resistance in the range of small currents, which in combination with junction capacitances leads to high time constants. By installing a resistor connected in parallel with the diodes, the time constant of the circuit can be minimized ( U. Dürig, L. Novotny, B. Michel, A. Stadler, Review of Scientific Instruments 68, page 3814, 1997 ), which also reduces the current resolution and sensitivity.

It are also known measuring devices that automatically between switch over different measuring ranges (Autorange switchover) and so measurements with a great momentum of 7 to 10 Allow orders of magnitude. Indeed Each switch takes up to 15 ms. Because during this Switching time can not be measured, thereby going large Lost amounts of information.

Task and solution

It is therefore the object of the invention, an electricity meter to provide the streams in one large dynamic range with low time constant too Measure while switching times in which it is not ready to measure is minimized over the prior art.

These The object is achieved by Ammeters according to main and secondary claim. Further advantageous embodiments will be apparent from the respective referenced subclaims.

Subject of the invention

In the context of the invention, two ammeters have been developed which are based in each case on an operational amplifier connected as a current-voltage converter. Their particular advantages over generic devices receive the devices according to the invention in each case by a targeted modification of the snubber resistor in the feedback path, which converts the current flow into a voltage and makes the operational amplifier so a current-voltage converter. In the device according to the main claim ("device 1"), the circuit resistance changes depending on the current to be measured In the device according to the independent claim ("device 2"), the circuit resistance comprises two diodes. These changes in the circuit resistance each represent the particular technical feature that characterizes the contribution of the device to the prior art. In each case, they solve the problem set out above of enabling current measurements in a wide dynamic range with a simultaneously low time constant and low non-measurable dead times. In particular part of description is an embodiment of A device is specified that combines the measures used in Device 1 and Device 2 to combine the specific advantages of both devices in one device.

Device 1

device 1 is a power meter, which at least one with connected a circuit resistor as a current-voltage converter Operational amplifier includes. According to the invention the resistance of the circuit resistance at least one first flowing (to be measured) current value smaller as at at least a second flowing current value, wherein the second current value is smaller than the first one.

This measure has the effect that the measuring sensitivity changes in a defined manner depending on the current value to be measured. The output voltage U _out of an operational amplifier connected as an (inverting) current-voltage converter is given by U _out = -I _in · R, where I _in the current to be measured and R are the circuit resistance. If the circuit resistance is reduced at a certain current value, the output voltage of the operational amplifier increases in magnitude as the current continues to increase less steeply.

As a result, the measuring range of the operational amplifier and thus of the current measuring device is extended. The output voltage U _out of each operational amplifier is limited by its saturation voltage U _sat . A reduction in the circuit resistance as a function of I _in causes the saturation voltage U _{sat is reached} only at a higher value of I _in . Thus, the meter can detect a higher maximum current without suffering from a lower resolution at lower currents to be measured.

The concrete dependence of the circuit resistance on the current flowing through as well as its absolute value must be dimensioned by the person skilled in the art with regard to the dynamic range of currents to be measured specifically. Preferably, however, a factor of between 10 ² and 10 ⁹ lies between the values of the circuit resistance for the two current values flowing through. Preferably, the circuit resistance varies in a range between 10 ² and 10 ¹¹ Ω.

The circuit resistance can decrease continuously with the current I _in to be measured. In a particularly advantageous embodiment of the invention, however, it is designed such that it decreases and, in particular discontinuously decreases here, when the current to be measured I _in exceeds a threshold value. This offers the advantage that above and below the threshold value, the output voltage U _{out of} the operational amplifier is still linear in I _in , which greatly simplifies the interpretation of the measurement result. Preferably, the crossing of threshold currents in the range between 10 ^-10 A and 10 ^-4 A leads to discontinuous changes in the circuit resistance by factors in the range between 10 ² and 10 ⁶ .

A discontinuous change in the snubber resistor has been realized with a particularly advantageous embodiment of the invention, in which the snubber resistor at least two series-connected resistors R and nR and at least one resistance nR connected in parallel limiter (current limiter), which releases the current flow when the he applied voltage exceeds a breakthrough value in terms of amount. The nomenclature nR is intended to express that nR is usually much larger than R, preferably by a factor between 10 ² and 10 ⁹ . The breakdown value is preferably between 0 and 12 V, more preferably between 0 and 5 V. Ideally, the breakdown value is less than or equal to the maximum voltage detectable by the analog-to-digital converter used for data acquisition at maximum sensitivity and thus resolution of the transducer.

If the current I _in through the resistors R and nR exceeds the threshold value, the voltage drop across the resistor nR caused by it reaches the breakthrough value of the limiter. The limiter releases the current flow and thus bridges the resistance nR, so that only the resistance R is effective as a circuit resistance. NR is now much greater than R (preferably by a factor between 10 ² and 10 ^9), the sensitivity of the operational amplifier decreases abruptly from several orders of magnitude when I exceeds _the threshold value. The current meter performs an automatic measuring range changeover at this threshold value. In contrast to conventional measuring devices with automatic measuring range switching, which require up to 15 ms for this switching, this switching happens according to the invention within significantly shorter times, eg. <0.1 ms for nR = 100 MΩ. The switching time is dependent on the resistance, since it forms with unwanted capacitances in the circuit an RC circuit having a time constant.

Of the Limiter may include, for example, at least one zener diode. A single Zener diode is sufficient for a monopolar current measurement out. For a bipolar current measurement, the limiter can be two comprise oppositely poled Zener diodes connected in series.

Zener diodes have high and non-linear leakage currents. About that In addition, they have large capacities. In addition, collect itself - as with all semiconductor diodes - during the transition in the conducting state on both sides of the semiconductor junction each minority carrier in the diode. When the voltage applied to the zener diode is below the breakdown voltage is attributed to these minority carriers an unwanted electrical conduction and delay so the transition to the blocking state. As the accumulation of minority carriers by diffusion only can be degraded again and diffusion a comparatively slower Process, this diffusion capacity limits the maximum Speed at which a Zener diode from conducting to blocking Condition can be transferred. Zener diodes are, compared to small signal switching diodes, not for fast Switching processes optimized.

In a particularly advantageous embodiment of the invention the limiter therefore at least one diode and one to the diode series-connected and reverse-biased voltage source. For bipolar measurements, the limiter includes beneficial at least two antiparallel connected arrangements of a diode and one connected to the diode in series and in the reverse direction polarized voltage source. The voltage sources provide preferred Voltages up to 12 V, especially preferred voltages up to 5 V.

From the point of view of a voltage dropping across the resistor nR, one of the two diodes then blocks, while the other one is permeable. However, in the branch with the transmissive diode, the voltage opposite it from the voltage source adds to the falling voltage. If the falling voltage is smaller in magnitude than the voltage from the voltage source, the sign changes with this addition, so that the result has the wrong polarity in order to pass the voltage dropping above nR to a permeable diode. Only when the voltage drop by the same amount as that of the counter voltage from the voltage source, that is achieved _in a threshold value I, which leads to such a falling voltage, it can cause a current to flow through the diode.

Of the Term of the diode includes all semiconductor devices that have the function a diode exercise, including LEDs, bipolar transistors or field effect transistors incorporated as diodes.

Across from a limiter with Zener diodes, this arrangement has the advantage that for breakdown voltages in the range up to ± 3.5 V leakage currents below 0.1 pA and effective capacities below 0.1 pF are feasible. The leakage current is the limiting Factor for the smallest, with the ammeter still detectable electricity. The capacity is the limiting Factor for the time constant and thus for the highest frequency one with the ammeter still detectable stream.

In a particularly advantageous embodiment of the invention, the current measuring device comprises a further operational amplifier, the input of which lies on a dependent of the current to be measured I _in potential. This operational amplifier can be connected as a simple voltage follower with gain factor 1, but also as a voltage booster with a different gain factor, which is tuned to the next level of signal processing, such as the measuring range of an analog-to-digital converter. Because of the additional operational amplifier, the current I can be measured simultaneously _in a second time, wherein due to the very high input resistance of the operational amplifier an effect on the measured potential and thus on the first measurement is negligible. Advantageously, the input of the further operational amplifier is at the potential which exists between the resistors R and nR. In this case, the resistor R has a double function: it converts the current I _in for the two measurements carried out at the same time to a voltage.

The output voltage of the further operational amplifier depends much less steep, as a rule by the current to be measured I _in from than the output voltage of the first, connected as a current-voltage amplifier operational amplifier. This applies in particular if the input of the further operational amplifier is at the potential between the resistors R and nR. Thus, the further operational amplifier performs independent measurement of I _in with significantly reduced sensitivity.

This is particularly advantageous when the output voltages of the operational amplifier are detected by means of analog / digital converters. Depending on the selected measuring resolution, these transducers only detect voltages within a certain measuring range. Tensions outside of this Measuring range can no longer be distinguished. If the output voltage U _{out1 of} the first operational amplifier reaches the limit of the measuring range, according to the prior art, the measuring range would have to be switched to the next higher level, which would inevitably be associated with a lower measuring resolution.

_{By contrast, according to} the invention, the output voltage U _{out2 of} the second operational amplifier can be measured on a second channel; as it is much less sensitive to the current I _in , it is still within the measuring range. By combining the measured values for U _out1 and U _out2 in a two-channel _measurement , a much larger dynamic range of currents at the highest possible resolution of the analog-to-digital converter can be measured than was possible in the prior art. In this case, the _{ranges that can be} meaningfully detected by consideration of U _out1 and U _out2 can certainly overlap for I _in , which in turn is of great advantage for calibration _measurements .

In a further advantageous embodiment of the invention is the Limiter via at least one bleeder with the Output of the other operational amplifier and over at least two parallel connected diodes with the input of the connected to another operational amplifier. The value of this Resistance should be chosen so that the voltage drop across negligible due to the leakage currents of the limiter is. Also, the time constant of the RC circuit should be out of the resistor and the parasitic capacities of the limiter a lot be smaller than the total time constant of the invention Ammeter, preferably by a factor of 5 and especially preferably smaller by a factor of 10. Meaningful values for the Ableitwiderstand are, for example, between 3 kΩ and 100 kΩ. He must not be too low, because if the limiter releases the current flow is a certain voltage drop across the bleeder resistor required so that at least one of the parallel switched diodes between the limiter and the input of the further operational amplifier opens and conduct electricity through the limiter. If the value of the Bleeder resistance would be the available, current flowing through the limiter is insufficient Generate voltage drop across the bleeder resistor. The Diodes between the limiter and the input of the further operational amplifier remained closed, and the limiter would be disabled.

By the leakage resistance and the diodes become the input and the output the further operational amplifier and a pole of the limiter held at almost the same potential. The parallel connected diodes Lock at very small potential differences between the pole of Limiter and the input of the further operational amplifier; If too much potential difference builds up, opens one of the diodes. Leakage currents of the limiter flow over the bleeder to the output of the further operational amplifier from. As a result, the bleeder resistor suppresses feedback of leakage currents of the limiter by the resistance nR, and the diodes reduce the repercussion of parasitic Capacities of the limiter on the potential between the Resistors R and nR or neutralize this reaction even completely. Diodes and bleeder resistor must be combined can be used: If diodes are used without bleeder resistor, Constant leakage currents flow through the limiter about that. If a bleeder resistor is used, but open the parallel connected diodes between the limiter and the input waived further operational amplifier, is the further operational amplifier positively fed back and is not stable anymore.

There the device according to the invention only a few Has components and therefore requires little space, it can For example, in a piezo-positioning device for a scanning tunneling microscope can be integrated. Because the tunnel current exponential including the distance between the tip and the sample depends, it changes within the shortest possible time Time by several orders of magnitude, so that the inventive Ammeter especially for this application suitable.

Device 2

The comprises ammeters according to the invention like device 1, at least one with a circuit resistance as a current-voltage converter wired operational amplifier. According to the invention, the circuit resistance comprises this op amp, a diode array of at least two rectified series-connected diodes and one over both diodes applied voltage source, each one of the diodes between the feed point for the current to be measured connected in the diode arrangement and a pole of the voltage source is.

Of the Term of the diode includes all semiconductor devices that have the function a diode exercise, including LEDs, bipolar transistors or field effect transistors incorporated as diodes.

Since the snubber resistor comprises diodes, the operational amplifier becomes logarithmic current-voltage transformer. By biasing the diodes with the inventively provided, applied via both diodes voltage source, the sensitivity of this converter can be adjusted advantageously. It has been advantageously recognized that this creates a current measuring device whose sensitivity can typically be changed in a time interval <0.1 ms. As a result, no information is lost in fast measurements with simultaneously high dynamics of the current Im to be measured.

There According to the invention, one of the diodes between each Feed point for the current to be measured in the diode array and one pole of the voltage source are connected, both poles the voltage source from the input of the operational amplifier isolated. At the same time it was realized that with only one voltage source Both diodes are biased and thus influenced in their resistance can be. In the invention Arrangement is a pole of the voltage source with the anode of the one Diode and the other pole of the voltage source with the cathode of connected to another diode.

The Voltage source may consist of a series connection of two or more Single voltage sources exist. Then the output signal of the Ammeter tapped between the individual voltage sources become. The output signal can then be distributed the total bias on the single voltage sources with a constant Offset be provided. In order to simplify the spelling the term "voltage source" is used hereinafter in singular; it detects at several existing single voltage sources respectively one or more of these voltage sources.

The voltage source can deliver a fixed or variable voltage. For example, a digital / analog converter can be used as the voltage source, so that the user can select the optimal compromise between current resolution and time constant, for example, in the range of very small currents at any time. Although the junction capacitance C of the diodes is constant, the differential resistance R _d = dU / dI depends on the bias, and the time constant is given by C ·R _d = C ·dU / dI.

The Voltage source can also be an AC voltage or a DC voltage deliver with an AC voltage modulated thereon. So, for example in the observation of a periodic phenomenon the sensitivity of the ammeter to the time program to follow the phenomenon.

Advantageously, at least two resistors are connected in series parallel to the voltage source and to the diodes. If the output of the ammeter between the resistors is tapped, these resistors can be used to shift the signal by an offset. So that over these resistors not too high voltage drops, they should be lower impedance than the input impedance of the next signal processing stage (such as analog / digital converter), in which the output signal of the ammeter is coupled. 3-100 kΩ per resistor are examples of meaningful values. The offset does not depend on I _in because the current to be measured does not flow across these resistors. The closed circuit of the voltage source and the resistors has no influence on the measurement signal, since the voltage source is actively regulated to a constant voltage between its poles.

Out theoretically there is the possibility to choose the value of these resistors so that an optimal compromise solution between sensitivity and time resolution for a specific experimentation task results. This solution could be through the use be realized by mechanical or solid state relays. However, these components lead to big ones Leakage currents and stray capacitances, resulting in the performance of the circuit significantly worsened.

In a particularly advantageous embodiment of the invention is a further feedback path from the first feedback path, which includes the snubber resistor with the diode array, for Input of the operational amplifier provided, this Feedback path in particular an inverting Amplifier and a capacitor can contain. Thereby can the time constant of the ammeter in the range smaller Currents are reduced. The capacitor compensates for the Junction capacitance of the diodes and the other parasitic Capacities of the circuit. The inverting amplifier is necessary, because the feedback otherwise in the wrong direction would work.

In a particularly advantageous embodiment of the invention, at least one resistor is provided between the input of the operational amplifier and the feed point for the current to be measured in the diode array. The current to be measured I _in effect in this resistor a voltage drop across the current a second time may be measured _in independent of the diode array I. This is particularly important for the calibration of the logarithmic current-voltage converter according to the invention, since just for this purpose particularly suitable fast diodes with low capacitance and small leakage currents have no exactly exponential current-voltage characteristic. In addition, the characteristics of diodes are highly temperature-dependent.

The output voltage U _{out of} the operational amplifier, with which the current is measured logarithmically, can be used for an independent second measurement of the current via the additional resistor. The current I _in leads to a voltage drop both in the diode arrangement and in the resistor, and both voltage drops add up to U _out . The diode array provides an I _In logarithmic contribution and the resistance in a linear contribution.

An operational amplifier connected as a voltage follower or voltage amplifier, whose input is at the potential of the feed-in point for the current to be measured in the diode arrangement, is advantageously provided for the independent second measurement of the current. Since it is a linear component, its characteristic is subject to a much weaker temperature influence than that of a diode. In addition, it may be generally advantageous to measure the dynamics of the current I to be measured _in both a linear and a logarithmic scale simultaneously.

Out the signal of the first (with the diode array connected) operational amplifier can be the linear signal component caused by the upstream resistor is subsequently eliminated. alternative the known value of this resistance can be directly into the mathematical Model for the diode array to be included.

The sensitivity of the second measurement can be adjusted via the resistance value. It can be very different from the sensitivity of the diode array. Analogous to device 1, therefore, two-channel measurements with different sensitivities can be carried out. By the simultaneous detection of the linear and the logarithmic current measurement, overall, with the highest possible resolution of the analog / digital converter used for the data acquisition, a much larger dynamic range of currents I _{in can be} covered than was possible according to the prior art.

In a particularly advantageous embodiment of the invention is the Total resistance of the series circuit of the resistor and the Diode arrangement at least a first flowing through (to be measured) current value smaller than at least a second flowing current value, wherein the second current value is smaller is the first Analog to device 1 changes then advantageously the sensitivity of the ammeter depending on the current flowing through, which Measurements with higher dynamics possible.

For example, for this purpose between the resistor and the feed point for the current to be measured in the diode arrangement, a further resistor and parallel to this further resistor and the diode arrangement, a limiter, which releases the current flow when the voltage applied to it exceeds a breakdown value, be connected. In this case, this further cons was generally much larger than the first, as in device 1. As soon as the current I _in a threshold value and thus the voltage across the limiter exceeds the breakdown value, the further resistor and the diode array are bridged by the limiter.

Upon further increase of the current I _in , the operational amplifier then responds as a linear current to voltage converter with a sensitivity determined by the first resistor. The ammeter is thus also a device 1 for linear current measurements. As described above, the linear measurement of the current I can be carried out _in then with particularly low time constant, and with particularly high resolution over a very wide dynamic range. If the current _I is below the threshold, the power meter measures it automatically logarithmic. If it is above the threshold, it measures it linearly.

That the diode array is bridged has the additional Effect that the current through the diodes and, consequently, the warming the diodes is limited. One too small heating (> 1 K) of the diodes endangers their calibration, since the characteristics the diodes are highly temperature dependent.

If the further resistance either absent or not parallel connected with the limiter alone causes the parallel connection the limiter to the diode array that the ammeter below of the threshold current both logarithmic and linear and above of the threshold current only reacts linearly. At the same time, then too limits the heating of the diodes.

For Switching the sensitivity alone would however, analogous to device 1 suffice, only the further resistance to bridge with the limiter.

Advantageous are means for applying resistance to a calibration current intended. Then, before or even during the actual Measuring the diode array to be calibrated. The calibration current can, for example, periodically and thus to the actual measurement signal be modulated. By demodulation of the measurement results for the linear and the logarithmic measurement can then be the actual measurement be calibrated for all measuring times. This is special advantageous if the measurement takes longer and the ammeter during which temperature fluctuations are exposed.

There Device 2 just flowing in scanning tunneling microscopes very small tunnel currents between tip and sample with high dynamic range, it is for use in such a Microscope particularly suitable.

Special description part

following the subject matter of the invention will become more apparent with reference to figures explained without the subject of the invention thereby is limited. It is shown:

1 : Example for device 1 for two-channel measurement.

2 : Characteristics of the two output channels of the device 1

3 : Specification of the embodiment 1

4 : Extension of the embodiment 3 to compensate for leakage currents and parasitic capacitances of the limiter.

5 : Exemplary embodiment of a device 1 for three-channel measurement.

6 : Exemplary embodiment 5 with alternative arrangement of the voltage sources.

7 Embodiments for device 2 with diodes connected in series (partial image a) and with parallel-connected diodes (partial image b).

8th : Device 2 with additional feedback.

9 : Simplified scheme of a bipotentiostat.

10 : Demonstration of the current resolution of device 2 in the lower fA range.

11 : Combination of device 2 and device 1.

12 : Combination of device 2 and device 1 with two linear and one logarithmic amplifier stage.

1 1 shows an exemplary embodiment of device 1. This current measuring device comprises an operational amplifier OA ₁ connected to the circuit resistor R + nR as a current-voltage converter. Parallel to the resistor nR, a limiter is connected, which bridges the resistor nR at a certain voltage drop across nR and thus at a certain current flow through nR. With this current value flowing through, the resistance value of the circuit resistance R + nR is effectively smaller than with a smaller current value at which the limiter is impermeable.

In addition, another, connected as a voltage follower operational amplifier OA _{2 is} provided, the (positive) input is on the prevailing between the resistors R and nR potential. Since R is much smaller than nR, the output voltage U substantially less steep reacts _out2 of OA ₂ to changes in the current to be measured I _in as the output voltage U _out1 of OA. ₁

This connection is in 2 explained in more detail. There, the output voltages U _out1 and U _{out2 are plotted} against the current I _in to be measured. U _out2 is basically given by U _out2 = -R · I _{in .} If over the resistor nR falls less than the breakdown voltage U _{thres of} the limiter (which corresponds to the threshold value for the current I _in ), U _{out1 is} given by U _out1 = -nR * I _in (nR >> R). If a greater voltage than U _{thres is lost across nR} , U _out1 ≈ ± U _thres - R · I _in . If the values for U _out1 and U _{out2 are detected} with an analog-to-digital converter, which is only able to distinguish voltage values within the hatched measuring range, the combination of U _out1 and U _{out2 makes} the entire measuring range in the highest resolution of the analogue / digital converter. Digital converter detectable. For I _in below the threshold value U _out1 is decisive, and _in I above the threshold value is U _out2 significantly.

und

für U_out1 und U_out2. U_sat repräsentiert die Sättigungs-Ausgangsspannung für den Operationsverstärker OA₁. Typischerweise wird U₁ = U₂ gewählt. Bei Wahl einer anderen Bestückung können asymmetrische Strombereiche implementiert werden. Die Stromauflösung und die Zeitkonstante für sehr kleine Ströme werden bestimmt durch D₁, D₂, die eingangsseitigen Leckströme für OA₁ und OA₂ sowie die Größe von R₂. Die Verwendung von LEDs führt zu sehr niedrigen Leckströmen, aber großen Sperrschichtkapazitäten (4–50 pF). Die sorgfältige Auswahl von Halbleiterdioden ermöglicht den Einsatz von Komponenten mit Streukapazitäten ~1 pF und Leckströmen kleiner als 0,1 pA bei Raumtemperatur. Durch Verwenden geeigneter Operationsverstärker für OA₁ und OA₂ (z. B. AD822 von Analog Devices) können mit R₁ = 10 kΩ und R₂ = 100 MΩ im Bereich kleiner Ströme Zeitkonstanten von ~0,2 ms und ein Rauschen, welches einem Stromrauschen von ~0,5 pA rms (1 kHz Bandbreite) am Eingang entspricht, erreicht werden. Durch die Wahl von U₁ = U₂ = 3,3 V und mit der Annahme, dass U_sat = 12 V, erzielt man z. B. ±0,8 mA als maximal messbaren Strom. Das heißt, der dynamische Meßbereich erstreckt sich im vorliegenden Beispiel über 9 Größenordnungen. Die weitere Erhöhung der Werte für R₂ führt zu einer deutlichen Erhöhung der Zeitkonstante, während die Stromempfindlichkeit nach wir vor durch die Leckströme der Dioden D₁ und D₂ bestimmt wird.In 3 is the embodiment of 1 further concretised. The resistors R ₁ = 10 kΩ and R ₂ = 100 MΩ are selected such that R ₁ << R ₂ . The limiter consists of two voltage sources (U ₁ = U ₂ = 3.3 V) and two diodes (D ₁ and D ₂ ). Both diodes are inactive as long as U ₂ > U _out1 > _{-U 1} . Outside this range, one of the diodes is activated so that a current can flow. The voltage drop across the resistor R ₂ is thus limited. The voltage source can be realized by diodes, zener diodes, LEDs, integrated circuits as voltage references or other components with analog functionality. The values of U ₁ and U ₂ determine the output voltage range of U _out1 and U _out2 . The former is given by (U ₂ , -U ₁ ), and the latter by (U _sat - U ₂ - 0.7 V; - U _sat + U ₁ + 0.7 V). This results in the maximum measurable current ranges

and

for U _out1 and U _out2 . U _sat represents the saturation output voltage for the operational amplifier OA ₁ . Typically, U ₁ = U _{2 is} selected. If you choose a different component, asymmetrical current ranges can be implemented. The current resolution and the time constant for very small currents are determined by D ₁ , D ₂ , the input-side leakage currents for OA ₁ and OA ₂ and the size of R ₂ . The use of LEDs results in very low leakage currents but large junction capacitances (4-50 pF). The careful selection of semiconductor diodes enables the use of components with stray capacitances ~ 1 pF and leakage currents of less than 0.1 pA at room temperature. By using suitable operational amplifiers for OA ₁ and OA ₂ (eg AD822 from Analog Devices), with R ₁ = 10 kΩ and R ₂ = 100 MΩ in the range of small currents, time constants of ~ 0.2 ms and a noise which is one Current noise of ~ 0.5 pA rms (1 kHz bandwidth) at the input corresponds to be achieved. By choosing U ₁ = U ₂ = 3.3 V and assuming that U _sat = 12 V, z. B. ± 0.8 mA as the maximum measurable current. That is, the dynamic range extends in the present example over 9 orders of magnitude. The further increase in the values for R ₂ leads to a significant increase in the time constant, while the current sensitivity is determined by the leakage currents of the diodes D ₁ and D ₂ .

4 shows a solution for the further reduction of stray capacitance and leakage currents of the limiter. Unlike the circuit in 3 an additional leakage resistance R ₃ and two diodes D ₃ and D _{4 are} introduced. As long as D ₁ and D _{2 are} blocked due to the very small potential difference applied across them, OA ₂ and R ₃ cause the potential at point A to be identical to the potential at point B and point A to be approximately at the potential of the point C is located at the exit of OA ₂ . There is practically no voltage drop between D ₃ and D ₄ . Thereby, the feedback of leakage currents of the limiter by the resistor R ₂ and the effects of parasitic capacitances of the limiter to the potential between R ₁ and R _{2 are} drastically reduced.

The size of the bleeder resistor R ₃ is critical for the time constant of the limiter circuit. The value of this resistor should be chosen so that the voltage drop across it is negligible due to the leakage currents of diodes D ₁ and D _{2 of} the limiter. Also, the time constant of the RC circuit of the resistor R ₃ and the junction capacitances of the diodes D ₁ and D _{2 of} the limiter should be much smaller than the total time constant of the current measuring device according to the invention, preferably by a factor of 5 and more preferably by a factor of 10 smaller. Examples of meaningful values of the leakage resistance R ₃ are between 3 kΩ and 100 kΩ. It must not be too low, because if the limiter releases the current flow, a certain voltage drop across the bleeder resistor is required so that at least one of the parallel connected diodes D ₃ and / or D ₄ opens between the limiter and the input of the further operational amplifier OA ₂ and conduct electricity through the limiter. If the value of the bleeder resistor is too low, it would become available standing, flowing through the diodes D ₁ and D ₂ current does not generate a sufficient voltage drop across the bleeder. The diodes D ₃ and D ₄ remained closed and the limiter would be inoperable.

Table 1 represents a compilation of typical parameters for the in 3 shown embodiment. Example 1.1 Example 1.2 Example 1.3 R ₁ 10 kΩ 10 kΩ 1 MΩ R ₂ 100 MΩ 1 GΩ 10 GΩ _Time constant for U _out1 ~ 0.015 ms 0.15 ms 1.5 ms _Current noise for a bandwidth corresponding to the time constant of U _out1 1.3 pA 0.13 pA 13 f maximum measurable current for U _out1 ± 33 nA ± 3.3 nA ± 0.33 nA Current noise (rms) for U _out2 at 10 kHz bandwidth ~ 150 pA ~ 150 pA ~ 15 pA maximum measurable current for U _out2 ± 0.6 mA ± 0.6 mA ± 6 μA dynamic range ~ 9 decades ~ 10 decades ~ 9 decades

Example 1.3 shows that in the case of R ₂ = 10 GΩ, the value of the resistance of R ₁ must be increased (in the example to 1 MΩ) to a sufficiently large overlap of U _out1 and U _out2 with respect to the required measurement accuracy to be able to reach the associated current ranges. As a consequence, both the current range for U _out2 and the dynamic range of the overall _{measurement are} reduced.

To solve this problem, additional gain stages are introduced. 5 shows, as an example, a circuit with three outputs. The voltage generators, which _realize the displacement of the output voltage U _out1 (analogous to U ₁ and U ₂ ), consist of the zener diodes Z ₁ , Z ₂ , Z ₃ and Z ₄ and the two resistors R ₆ and R ₇ . All other components have the same functionality as in 4 , In this embodiment, OA _{1 is} connected via the consisting of R ₁ , R ₂ and R ₃ circuit resistance as a current-voltage amplifier, wherein R ₁ << R ₂ << R ₃ . The voltages of the voltage sources U ₁ , U ₂ , U ₃ and U ₄ should be selected such that U ₁ > U ₂ and U ₄ > U ₃ . OA OA ₂ and ₃ are additional wired as a voltage follower operational amplifier, the (positive) input is each on a the current to be measured I _in dependent potential (on both sides of the resistor R _2).

Because of U ₄ > U ₃ , U ₁ > U ₂ and R ₂ <R ₃ , the breakdown voltage of the limiter is first reached with increasing current I _in which R ₃ is connected in parallel. With further increasing current I _in the breakdown voltage of the limiter is later reached, which is connected in parallel to the entire series circuit of R ₂ and R ₃ . The resistor nR here corresponds to the series connection of R ₂ and R ₃ , and in addition to the nR limiter connected in parallel, only a limiter connected in parallel with R _{3 is} provided. Would the output voltages of all operational amplifiers be as in 2 plotted, the curve for U _out1 would have two kinks at two different thresholds for I _in , each decreasing its slope. Because R ₁ << R ₂ I _in behind R ₂ results in a substantially higher potential than between R ₁ and R ₂ so that the curve for U _out2 much steeper with I _in increases as the curve for U _out3. Through the synopsis of all three channels, different measuring ranges of the current I _in can be measured simultaneously by several orders of magnitude with the same high-sensitivity measuring range of the analog / digital converter.

An alternative arrangement of the voltage sources is in 6 shown. The circuit is dimensioned as follows: U ₁ * = U ₁ - U ₂ and U ₄ * = U ₄ - U ₃ .

Table 2 summarizes characteristic data of the circuit for U ₁ = U ₂ = 7V, U ₃ = U ₄ = 3.5V and U _sat = 12V together. For Example 2.3, operational amplifiers with extremely low input offset voltages must be selected. Examples are AD 549 or AD 795 from Analog Devices, or OPA 129 from Texas Instruments. Example 2.1 Example 2.2 Example 2.3 R ₁ 1 kΩ 1 kΩ 390 Ω ^R 2 330 kΩ 1 MΩ 1 MΩ R ₃ 100 MΩ 1 GΩ 10 GΩ _Time constant for U _out1 ~ 0.03 ms ~ 0.3 ms ~ 3 ms Current Noise (rms) for a bandwidth corresponding to the time constant of U _out1 1 pA 0.1 pA 10 fA maximum measurable current for U _out1 ± 33 nA ± 3.5 nA ± 0.35 nA Current noise (rms) for U _out2 at 10 kHz bandwidth ~ 10 pA ~ 3.3 pA ~ 3.3 pA maximum measurable current for U _out2 ± 10 μA ± 3.5 μA 13.5 μA Current noise (rms) for U _out3 at 10 kHz bandwidth ~ 500 pA ~ 500 pA ~ 1 nA maximum measurable current for U _out3 ± 4mA ± 4mA ± 10mA dynamic range ~ 9.5 decades ~ 10.5 decades ~ 12 decades

The Time constants for the applications in the embodiment are higher than in the case of the dual preamplifier. This results from the stray capacitances of the two limiters. Nevertheless, a high dynamic range with a large overlap achieved adjacent power ranges.

7a shows a first embodiment of the device 2. The operational amplifier is connected to a circuit resistor as a current-voltage converter, which serves as schaltungswiderstand a diode array of two rectified series connected diodes and a voltage applied via both diodes voltage source. In this embodiment, two resistors are connected in series parallel to the diodes and the ammeter. The current I _in to be measured is fed between the cathode of one diode and the anode of the other diode in the diode array. In this case, one of the diodes is in each case connected between the feed-in point for the current to be measured in the diode arrangement and a pole of the voltage source. As a result, the voltage U _out between the two resistors, on the side facing away from the diode array of the resistors, tapped.

7b shows a further embodiment of device 2. In this embodiment, the voltage source is a series circuit of two single voltage sources. The voltage U _out is tapped between the two individual voltage sources. 7b At first glance, it may seem like an alternate embodiment that works differently than the one in 7a shown embodiment. That in essence it is the same circuit, becomes apparent, if one mentally in 7b omits the voltage source U _bias2 and then _rotates D1 by 90 ° counterclockwise and U _bias1 and D ₂ by 90 ° clockwise.

8th shows a further embodiment of device 2, in which leads to reduce the time constant in the range of small currents additional feedback from the first, the diode array containing feedback path via the amplifier AMP and the capacitor C to the input of the operational amplifier. The gain A (A <0) and the capacitance of the capacitor should be chosen such that | AC | ≈ C _D1 + C _D2 , where C _D1 and C _{D2 represent} the junction capacitances of the diodes D ₁ and D ₂ . In the case of C≈C _D1 + C _D2 , z. B. A ≈ -1 as a suitable solution. The bias voltage U _bias can be realized in various ways, for. Example by Zener diodes, with a voltage reference, with digital-to-analog (D / A) converters, operational amplifiers, resistive networks, potentiometers, etc. The temperature stabilization of the critical components of the voltage source and the diodes D ₁ and D ₂ is required. R ₁ and R ₂ allow the displacement of the measurement signal by a constant offset. In the case of R ₁ = R ₂ and U _out = 0, I _in = 0 results. The connection of the two poles of the voltage source via the resistors can also be omitted; In this case both point "A" and point "B" can be used as taps of the output signal. For the selection of the diodes D ₁ and D ₂ results in a great variety. Examples are diodes with low leakage currents and capacitances, LEDs, bipolar transistors or field effect transistors built in as diodes. LEDs show extremely low leakage currents, but also high junction capacitances (4 to 50 pF). The LM394 bipolar transistor also exhibits low leakage currents and an exactly exponential current-voltage characteristic, but also high capacitances. An optimal combination of low leakage currents and stray capacitances could be achieved by using JFETs that were integrated into the circuit as diodes.

Table 3 shows typical operation parameters for different bias currents I _bias , which flow due to the bias voltage U _bias , and the optimization of the positive feedback to compensate for stray capacitance. diode JFET with low leakage currents TO-72 3mm Amber LED LM394 TO-99 Bias current (I _bias ) 1 pA 10 pA 100 pA 1 pA 100 pA 10 pA 100 pA Time constant for low currents ~ 2.5 ms ~ 0.25 ms ~ 25 μs ~ 40 ms ~ 0.4 ms ~ 10 ms ~ 1 ms Noise characteristic of the input current, rms with reference to 1 Hz bandwidth ~ 1 fA ~ 2.5 fA ~ 8 fA ~ 1.2 fA 11 f ~ 2.5 fA ~ 8 fA maximum current ± 2mA ± 2mA ± 2mA ± 0.5 mA ± 0.5 mA ± 2mA ± 2mA

Of the maximum current in this circuit is determined by the heat dissipation of the diodes due to current flow. In the presented solution the current can be limited so that the temperature of the diodes does not change by more than ± 0.5K. The maximum Electricity was taking into account the thermal coefficients of the components.

Device 2 was incorporated as a basic component in a four-electrode potentiostat. 9 shows a simplified scheme.

The bipotentiostat controls the potential of two electrodes with respect to a reference electrode RE using a counter electrode CE, the current being recorded separately by both working electrodes WE ₁ and WE ₂ . For this purpose, the current flowing through each of the working electrodes is transformed by a respective device 2 into a voltage signal. In addition, the potential of each of the two working electrodes is measured via another operational amplifier connected as a voltage follower. Also, the potential of the reference electrode RE is measured by a voltage follower. The measurement is controlled by means of a 16-bit digital-to-analog (D / A) converter and an 8-channel analog-to-digital (A / D) converter via a microcontroller MC which has a buffer BUF. The user interface and the presentation of the data were realized via a self-developed Windows program. The microcontroller MC communicates with a PC via a galvanically isolated USB interface.

The control unit supplies two voltage signals. A signal determines the potential of the reference electrode RE and therefore represents the potential of the working electrode WE ₁ , since this is basically measured relative to the reference electrode RE and WE _{1 is kept} at ground potential. The second signal determines the potential of the working electrode WE ₂ and thus represents the potential difference between the working electrodes WE ₁ and WE ₂ . Both signals are generated according to a user-specified, discrete time program. The signals are then sectionally constant and can have jumps between these sections. Alternatively, an analog ramp generator SG which can be connected via the switch is implemented. The use of a ramped, steady voltage waveform has advantages for cyclic voltammetric experiments with high capacitive systems.

From the voltage signal that each generates a device 2 from the current flowing through a working electrode current, the potential of this working electrode is removed via a differential amplifier before the signal is passed to the A / D converter. Otherwise, at the working electrode WE _{2, in} addition to the voltage drop caused by the current to be measured by WE ₂ , the potential applied to this electrode via the D / A converter would also be measured. At the working electrode WE ₁ would be measured without the differential amplifier in addition to the caused by the current to be measured by WE ₁ voltage drop also a parasitic potential of this nominally at ground potential working electrode.

The 16 bit A / D converter operates at a sampling rate of 80 kHz. 5 internal signals are stored successively. In the data logging, both the time interval between two data points (acquisition time) and the integration time within which data of the same internal source can be averaged be freely set by the user. Thus, the user can make an optimal selection of the recording parameters, adapted to the relevant time scale of the current experiment.

The current resolution of the instrument according to 9 in the lower fA range is in 10 using the example of a cyclic voltammogram for an isolated to the outermost peak gold nanoelectrode in 0.1 MH ₂ SO ₄ , demonstrated. In this illustration, the current I through the working electrode WE _{1 is} plotted against the voltage U between the working electrode WE ₁ and the potential RHE of the reference electrode. The voltage was changed at a sweep rate of 50 mV / s.

In the voltage range between 0.2 and 1.1 V, no electrochemical processes take place. The constant current in this voltage range, apart from the noise, results from an undesired capacitance of approximately 2 pF. At point P ₁ , oxides form on the surface of the electrode. After reversing the voltage, these oxides are electrochemically reduced again at point P ₂ .

From the current peak caused by the reduction at the point P ₂ , the charge converted in this reduction and, in turn, the effective uninsulated area at the foremost tip of the nanoelectrode can be determined to be approximately 0.7 μm ² .

The quiescent current of the sensor diode was set to ~ 3 pA. This results in a time constant of ~ 2 ms and a noise of 5 fA RMS for a bandwidth of 1 Hz. The integration time is 0.2 s. Alone 10 plotted data are unfiltered. All currents are less than ± 1 pA.

The one for the measurement of 10 used construction according to 9 was used to study the electrochemical properties of nanoelectrodes, molecular bridges metal | molecule | metal bridges as well as nanowires on templates prepared by lithographic methods.

11 shows an embodiment of a combination of device 2 with device 1. Between the input of the operational amplifier OA ₁ and the feed point for the current to be measured in the diode array two resistors R and nR are connected. A limiter is connected in parallel to the resistor nR and to the diode arrangement. The operational amplifiers OA ₂ and OA ₃ are voltage followers which pick up the potential behind the resistors R + nR and R, respectively. For I to be measured flows _in below the threshold current for which the limiter is impermeable, I is measured _in so simultaneously to U _out1 logarithmically as well as U and U _{out2 out3} with two different sensitivities linear. Above the threshold, further increases of I _in only via the signal U _{out 3} can be detected.

This circuit combines the advantages of a multi-channel current-voltage converter (device 1) with a tunable logarithmic current-voltage converter (device 2). This solution reduces the complex calibration routine of the logarithmic unit. A routine for self-calibration of the diodes may be implemented, which may be retrieved simultaneously with a current measurement. Due to the overlap of the current _ranges (analogous to device 1), which correspond to the output voltages U _out2 and U _out1 , the output signal at U _out1 can be calibrated with U _out2 , since U _out2 does not require calibration (linear components). In addition, the logarithmic output stage is used only in a limited dynamic current range in which the diodes used exhibit an exactly exponential current-voltage characteristic. By this combination, the thermal control of the current-sensing diodes can be saved. In addition to simplifying the components, this also results in considerable space savings.

In 12 Two linear amplifier stages (consisting of the components OA ₁ , OA ₂ , OA ₃ , R ₁ and R ₂ ) are used. The limiting circuit (limiter) consists of U ₁ , U ₂ , D ₁ , D ₂ , D ₅ , D ₆ and R ₃ , and limits the voltage drop across R ₂ . The voltage source U _bias , D ₃ , D ₄ , R ₄ , R ₅ , AMP and C represent the capacitance-compensated logarithmic current-voltage converter (see description to 8th ). This circuit has two advantages: (1) Very small currents flow through the diodes D ₃ and D ₄ . As a consequence, the heat dissipation at these diodes is negligible. The choice of suitable values for R ₁ allows the extension of the measurable current range to significantly higher values. (2) Careful selection of R ₂ allows for a sufficiently large overlap of the current _ranges corresponding to U _out1 and U _out2 . This results in easy access to autocalibration of the logarithmic component simultaneously to a current, ongoing measurement. As a consequence, no temperature control of D ₃ , D ₄ and the voltage source U _{bias is} necessary, whereby the circuit requires significantly less space.

The circuit is equipped with a calibration resistor R ₆ and a switch SW. This allows the generation of a stream for performing the autocalibration routine. The switch can be a me be a chanical or a "solid state" relay. By actuating the switch SW and applying a corresponding calibration voltage, a current flow through D ₁ , D ₂ , R ₂ , D ₃ and D ₄ can be generated. By careful selection of the values for R ₃ , R ₆ and the voltage U _cal presented before the switch SW, such a current can be generated so that a sufficiently large overlap of the measurable currents for U _out1 and U _out2 can be achieved, which in turn the optimal auto-calibration is possible.

Table 4 shows performance data for typical constructions of the embodiment 12 compiled. diode type Low leakage JFET R ₁ 390 Ω 390 Ω 390 Ω R ₂ 1 MΩ 1 MΩ 1 MΩ Bias current (I _bias ) 1 pA 10 pA 100 pA Time constant for small currents ~ 4 ms ~ 0.4 ms ~ 40 μs Noise level of the input current, rms, with reference to a bandwidth of 1 Hz, U _out1 ~ 1 fA ~ 2.5 fA ~ 8 fA Maximum measurable current at U _out1 ± 3.5 μA ± 3.5 μA ± 3.5 μA Noise Level, rms, for U _out2 with reference to a bandwidth of 10 Hz ~ 3.3 pA ~ 3.3 pA ~ 3.3 pA Maximum measurable current at U _out2 ± 3.5 μA ± 3.5 μA ± 3.5 μA Noise Level, rms, for U _out3 with reference to a bandwidth of 10 Hz ~ 1 nA ~ 1 nA ~ 1 nA Maximum measurable current at U _out3 ± 10mA ± 10mA ± 10mA

QUOTES INCLUDE IN THE DESCRIPTION

This list The documents listed by the applicant have been automated generated and is solely for better information recorded by the reader. The list is not part of the German Patent or utility model application. The DPMA takes over no liability for any errors or omissions.

Cited non-patent literature

• - YB Acharya, SG Tikekar., Review of Scientific Instruments 64, p. 1652, 1993 [0003]
• U. Dürig, L. Novotny, B. Michel, A. Stadler, Review of Scientific Instruments 68, page 3814, 1997; J. He, O. Sankey, M. Lee, NJ Tao, XL Li, S. Lindsay, Faraday Discussions 131, page 145, 2006 [0003]
• YB Acharya, SG Tikekar., Review of Scientific Instruments 64, p. 1652, 1993; U. Dürig, L. Novotny, B. Michel, A. Stadler, Review of Scientific Instruments 68, page 3814, 1997 [0003]
• U. Dürig, L. Novotny, B. Michel, A. Stadler, Review of Scientific Instruments 68, page 3814, 1997 [0004]

Claims (20)

Ammeter, comprising at least one with a circuit resistor as a current-voltage converter wired operational amplifier, characterized by a schaltungswiderstand whose resistance value at least a first flowing current value is less than at least a second flowing current value, wherein the second current value is smaller than the first one. Electricity meter according to claim 1, characterized by a snubber resistor whose resistance value decreases, when the current to be measured exceeds a threshold. Electricity meter according to one of the claims 1 to 2, characterized by a circuit resistance comprising at least two resistors R and R connected in series nR and at least one parallel to the resistor nR Limiter, which releases the flow of current when the applied to him Voltage exceeds a breakthrough value. Electricity meter according to claim 3, characterized by a limiter comprising at least one zener diode. Electricity meter according to claim 4, characterized by a limiter comprising two oppositely poled, in series switched zener diodes. Electricity meter according to one of the claims 3 to 5, characterized by a limiter comprising at least a diode and a diode connected in series and in the reverse direction polarized voltage source. Electricity meter according to claim 6, characterized by a limiter comprising at least two anti-parallel connected Arrangements of a diode and a series connected to the diode and in the reverse direction ge polten voltage source. Electricity meter according to one of the claims 1 to 7, characterized by another, as a voltage follower or voltage amplifier wired operational amplifier, whose input depends on a current to be measured Potential lies. Electricity meter according to claim 8, characterized characterized in that the input of the further operational amplifier lying on the potential, that between the resistances R and nR prevail. Electricity meter according to claim 9, characterized characterized in that the limiter has at least one bleeder resistor with the output of the further operational amplifier and over at least two parallel connected diodes with the input of the connected to another operational amplifier. Ammeter, comprising at least one wired with a circuit resistor as a current-voltage converter Operational amplifier, characterized by a circuit resistance comprising a diode array of at least two equally addressed series connected diodes and one across both diodes applied voltage source, wherein each one of the diodes between the Feed point for the current to be measured in the diode array and a pole of the voltage source is connected. Electricity meter according to claim 11, characterized by a voltage source comprising a series circuit of at least two single voltage sources. Electricity meter according to one of the claims 11 to 12, characterized by at least two parallel to the voltage source and resistors connected in series with the diodes. Electricity meter according to one of the claims 11 to 13, characterized by a further feedback path from the first feedback path to the input of the operational amplifier. Electricity meter according to claim 14, characterized by a further feedback path containing a Capacitor and an inverting amplifier. Ammeter according to one of claims 11 to 15, characterized by at least one resistor between the input of the operational amplifier and the feed point for the measured Current in the diode array. Electricity meter according to claim 16, characterized by at least one as a voltage follower or voltage amplifier wired operational amplifier, whose input on the Potential of the entry point for the current to be measured lies in the diode array. Electricity meter according to one of the claims 16 to 17, characterized in that the total resistance of the Series connection of the resistor and the diode array at a first flowing current value is smaller than at least a second current flowing value, wherein the second current value is smaller than the first. Electricity meter according to claim 18, characterized characterized in that between the resistor and the feed point for the current to be measured in the diode array another Resistance as well as parallel to this further resistance and to Diode arrangement a limiter, which releases the flow of current when the voltage applied to it exceeds a breakthrough value, are switched. Electricity meter according to one of the claims 16 to 19, characterized by means for applying the resistance with a calibration current.