CN117792087A - A low-voltage non-isolated DC converter and its control method - Google Patents

Abstract

The present invention discloses a low-voltage non-isolated DC converter and a control method thereof . The low-voltage non-isolated DC converter comprises an input power supply DC1 , an input filter capacitor _C1 , a switch tube Q1 , a resonant capacitor Cr _{, a} first rectifier diode D1 _{, a} second rectifier diode D2 _, a coupling inductor Tr _, a resonant inductor Lr and an output filter capacitor _C2 ; the input filter capacitor C1 is connected _in parallel with the input power supply DC1 , the output filter capacitor C2 is connected in parallel with the output end, the coupling inductor _Tr , the resonant inductor Lr , the resonant capacitor Cr and the second rectifier diode _D2 are connected in series in sequence , wherein the coupling inductor Tr comprises a first inductor L1 and _a second inductor L2 _, the converter realizes bidirectional excitation of the coupling inductor Tr _, so that the coupling inductor has a transformer function, the utilization rate of the magnetic device is improved, and the voltage step-up ratio of _the converter can be improved.

Description

A low voltage non-isolated DC converter and control method thereof

Technical field

The invention relates to a low-voltage non-isolated DC converter and a control method thereof, and belongs to the technical fields of power electronics technology and battery equipment.

Background technique

At present, primary clean energy such as wind power generation is widely used. Since primary clean energy such as wind power itself is unstable and has a strong time-varying property that is difficult to predict, the electric energy generated is unstable and the voltage variation range is wide, which brings great challenges to the grid connection of new energy. In order to achieve the grid connection of new energy, the power electronic converter in the new energy product needs to convert the input voltage with a wide range of variations into a stable high voltage, and the boost ratio has a high requirement. As a classic single-tube boost circuit, the non-isolated Boost boost circuit is a commonly used circuit for power electronic converters in new energy products. For example, to achieve a 20-fold boost ratio, the duty cycle of the switch in the non-isolated Boost boost circuit needs to reach 0.95. It is difficult to achieve this extreme duty cycle in engineering, and a high duty cycle can easily cause the system to lose control, increase converter losses, and reduce converter load capacity.

In order to avoid the occurrence of extreme duty cycles, coupled inductors are currently used to improve the boosting capability of non-isolated Boost circuits. For example, the Chinese patent with patent number ZL201511000295.3 and publication date of 2018-06-01 has a solution. As shown in Figure 1, it uses a coupled inductor to replace the input filter inductor in the non-isolated Boost boost circuit. The current directions of the two inductors of the coupled inductor in the patent change in one direction, the core utilization rate is low, and due to insufficient coupling , the leakage inductance of the coupling inductor causes the switching tube to generate voltage spikes at the switching moment, so an additional auxiliary absorption circuit is provided.

Alternatively, multiple filter inductors can be added to different locations of the non-isolated Boost circuit to couple them, and the circuit's operating mode can be changed by adding additional switch tubes. By superimposing the energy storage of different filter inductors, the purpose of increasing the boost capability can be achieved; however, the current direction of the coupled inductor changes in one direction, and it cannot be bidirectionally excited like a transformer, and more switch tubes are used. Switching changes the connection mode of the energy storage inductor, causing a sudden change in the voltage across the energy storage inductor, introducing new high-frequency components and exacerbating the EMI problem.

Summary of the invention

In order to solve the problems existing in the prior art, the present invention provides a converter in which a coupled inductor has a transformer function, thereby improving the voltage step-up ratio of the converter.

In order to achieve the above objects, the technical solutions proposed by the present invention are:

A low-voltage non-isolated DC converter, including input power supply DC 1, input filter capacitor C ₁ , output filter capacitor C ₂ , coupling inductor Tr , switching tube Q ₁ and second rectifier diode D ₂ ;

The input filter capacitor C ₁ is connected in parallel with the input power supply DC 1, and the output filter capacitor C ₂ is connected in parallel with the output terminal;

The coupling inductor Tr includes a first inductor L ₁ and a second inductor L ₂ . The two ends of the first inductor L ₁ are endpoint 1 and endpoint 2 respectively. The endpoint 1 of the first inductor L ₁ is connected to the positive pole of the input power supply DC 1 . The cathode of the second rectifier diode D ₂ is connected to the positive terminal of the output;

The drain of the switch tube Q1 is connected to the terminal 2 of the first inductor L1 , and the source is connected to the negative electrode of _the input power supply DC1 and the negative terminal of _the output;

It also includes a resonant capacitor C _r , a resonant inductor L _r and a first rectifier diode D ₁ ; the inductance value of the coupling inductor Tr after two inductors L ₁ and L ₂ are connected in series is several times the inductance value of the resonant inductor L _r ;

The second inductor L ₂ , the resonant capacitor C _r and the resonant inductor L _r are connected in series with each other to form a series branch. One end of the series branch is connected to the end point 2 of the first inductor L ₁ and the other end is connected to the anode of the second rectifier diode D ₂ , the anode of _the first rectifier diode D1 is connected to the terminal 1 of L1 , and the cathode _is connected to _the anode of D2 ;

The switch tube Q1 is turned off or on, so that _the currents on _the first inductor L1 and _the second inductor L2 flow in the same direction or in the opposite direction, thereby realizing bidirectional excitation of the coupled inductor Tr .

A further design of the above technical solution is as follows: the inductance value of the two inductors L ₁ and L ₂ connected in series of the coupling inductor Tr is greater than or equal to 5 times of the resonant inductor L _r .

The converter also includes a third rectifier diode D ₃ and a voltage dividing branch connected in parallel to the output end and formed by at least two voltage dividing capacitors in series. The anode of the third rectifying diode D ₃ is connected to the drain and cathode of the switching tube Q ₁ Connect to one voltage dividing end of the voltage dividing branch.

The two ends of the second inductor L ₂ are terminal 3 and terminal 4 respectively; the terminal 1 of the first inductor L ₁ and the terminal 3 of the second inductor L 2 are terminals of the same name, or the terminal 2 of the first inductor L ₁ and the terminal 3 of the second inductor L ₂ are terminals of the same name. The terminal 4 of the second inductor L ₂ is the terminal of the same name.

The second inductor L ₂ , the resonant inductor L _r and the resonant capacitor C _r are connected in series in sequence. The end point 3 of the second inductor L ₂ is connected to the end point 2 of the first inductor L ₁ , and the end point 4 is connected to one end of the resonant inductor L _r .

The resonant capacitor C _r , the second inductor L ₂ and the resonant inductor L _r are connected in series in sequence. One end of the resonant capacitor C _r is connected to the end point 2 of the first inductor L ₁ , and the other end is connected to the end point 3 of the second inductor L ₂ .

The resonant capacitor Cr , _the resonant inductor Lr and the second inductor L2 are connected in series in sequence. One end of _the resonant capacitor Cr is connected to the terminal ₂ of _the first inductor L1 _, and the other end is connected to one end of the resonant inductor Lr . The other end _of the resonant _inductor Lr is connected to the terminal ₃ of the second inductor L2 .

The second inductor L ₂ , the resonant capacitor C _r and the resonant inductor L _r are connected in series in sequence. The end point 3 of the second inductor L ₂ is connected to the end point 2 of the first inductor L ₁ , and the end point 4 is connected to one end of the resonant capacitor C _r .

The resonant inductor L _r , the second inductor L ₂ and the resonant capacitor C _r are connected in series in sequence. One end of the resonant inductor L _r is connected to the end point 2 of the first inductor L ₁ , and the other end is connected to the end point 3 of the second inductor L ₂ . .

A control method for a low-voltage non-isolated DC converter, used to control the above-mentioned low-voltage non-isolated DC converter. The control method specifically includes:

Apply a fixed conduction time to the switching tube Q ₁ so that the resonant inductor L _r and the resonant capacitor C _r complete half-cycle resonance during the conduction period of Q ₁ ;

By adjusting the switching frequency f _s of Q ₁ to change the conduction ratio of Q ₁ , the change in the boost ratio of the converter output voltage compared to the input voltage is achieved.

Compared with the existing technology, the present invention has the following advantages:

In the present invention, a resonant inductor L _r and a resonant capacitor C _r are connected in series to the second inductor L ₂ of the coupled inductor of the existing boost converter to form a series branch. The series branch is connected to the first inductor L ₁ and passes through the third inductor L 2 . The anode of the rectifier diode D1 is connected to _the first inductor L1 , and the cathode is connected to the series branch. Therefore, when the switch tube Q1 is turned _on , _a current loop of _the second inductor L2 in the coupling inductor Tr is constructed, realizing coupling. The bidirectional excitation of the inductor Tr enables the coupled inductor to function as a transformer, and the utilization rate of the magnetic device is improved, which in turn can increase the boost ratio of the converter and achieve high-gain boost.

When the converter of the present invention is used, the first inductor L ₁ and the second inductor L ₂ of the coupled inductor and the resonant capacitor C _r are used to jointly store energy, thereby reducing the voltage stress of the diode and switching tube in the circuit; at the same time, the resonant inductor L is used _r resonates with the resonant capacitor C _r , allowing the leakage inductance energy of the coupling inductor to be absorbed, solving the voltage spike problem caused by the leakage inductance energy without the need for additional peripheral absorption circuits.

The converter of the present invention uses only one switch tube, does not require multiple switch tubes to switch and change the connection mode of the energy storage inductor, does not introduce new high-frequency components, and has little EMI problem.

Description of the drawings

Figure 1 is an existing DC boost conversion circuit diagram;

Figure 2 is a schematic diagram of a low-voltage non-isolated DC converter according to an embodiment of the present invention;

Figure 3 is a schematic diagram of the simulation verification results of Embodiment 1 of the present invention;

4 is a schematic diagram of a low voltage non-isolated DC converter according to a second embodiment of the present invention;

5 is a schematic diagram of a low voltage non-isolated DC converter according to a third embodiment of the present invention;

FIG6 is a schematic diagram of simulation verification results of Embodiment 3 of the present invention;

FIG. 7 is a schematic diagram of a low voltage non-isolated DC converter according to a fourth embodiment of the present invention.

Detailed ways

The present invention will be described in detail below with reference to the accompanying drawings and specific embodiments.

Example 1:

As shown in Figure 2, it is a schematic diagram of the low-voltage non-isolated DC converter of this embodiment and a specific implementation diagram. The DC converter includes an input power supply DC 1, an input filter capacitor C ₁ , a switching tube Q ₁ , a resonant capacitor C _r , and a Rectifier diode D ₁ , second rectifier diode D ₂ , coupling inductor Tr , resonant inductor L _r and output filter capacitor C ₂ ; the input filter capacitor C 1 is connected in parallel with the input power supply DC ₁ , and the output filter capacitor C ₂ is connected in parallel with the input power supply DC 1 The output end is connected in parallel, and the coupling inductor Tr , the resonant inductor L _r , the resonant capacitor C _r and the second rectifier diode D ₂ are connected in series in sequence. The coupling inductor Tr includes the first inductor L ₁ and the second inductor L ₂ . The two ends of L ₁ are respectively Endpoint 1 and endpoint 2, the two ends of L ₂ are endpoint 3 and endpoint 4 respectively. In this embodiment, endpoint 1 and endpoint 3 are ends with the same name. Alternatively, endpoint 2 and endpoint 4 can be ends with the same name. Endpoint 1 of L ₁ is connected to DC. The positive electrode _of 1, endpoint 2 is connected to the drain of switch Q1 and endpoint ₃ of L2 , endpoint ₄ of L2 is connected to one _end of resonant _inductor Lr , the other end of resonant inductor Lr is connected to one _end of Cr , the other end of Cr One end is connected to the anode of the second rectifier diode D ₂ , the cathode of D ₂ is connected to the positive terminal BUS+ of the output, the anode of the first rectifier diode D ₁ is connected to the anode of DC 1, and the cathode of D ₁ is connected to the anode of C _r and the rectifier diode D ₂ the anode; the negative terminal of DC 1, the negative terminal BUS- of the output and the source of the switch tube Q ₁ are connected and grounded.

In this embodiment, the switch tube Q1 can be _a high-frequency switch tube with an anti-parallel diode, or a high-frequency switch tube connected in parallel with an integrated diode, a parasitic diode or an external diode; the resonant capacitor Cr can be a high-frequency non-polar capacitor or a high-frequency polar capacitor. The resonant inductor _Lr can be the leakage inductance of the coupled inductor Tr , or an external inductor, or a combination of the leakage inductance and the external inductor . The coupled inductor Tr in the converter of this embodiment can realize bidirectional excitation, and the specific principle is:

Assume that the turns ratio of the first inductor L ₁ and the second inductor L ₂ is 1: n , n ≥ 1, and the inductance of L ₁ is set to L ₀ , then the inductance of L ₂ is n² * L ₀ ; the resonant inductor L _r It is the coupling leakage inductance of the two inductors or the sum of the external inductance and the leakage inductance. Let the voltage of the input power supply DC 1 be V ₁ and the output voltage be V _o .

When the switch Q ₁ is turned on and the second rectifier diode D ₂ is turned off, the voltage on the first inductor L ₁ is V _{L 1_on} = V ₁ , terminal 1 is positive and terminal 2 is negative; the second inductor passes through the Coupling of an inductor, the current flows from terminal 4 to terminal 3, the voltage on it is V _{L 2_on} = nV ₁ voltage, the voltage direction is positive at terminal 3, and negative at terminal 4; pull down the cathode of the first rectifier diode D ₁ voltage, causing the first rectifier diode D ₁ to conduct, the input power supply DC 1 charges the resonant capacitor C _r through the first rectifier diode D ₁ , and at this time, the resonant inductor L _r resonates with the resonant capacitor C _r , and the resonant frequency . When the resonant inductor L _r and the resonant capacitor C _r resonate for half the resonant period 0.5/ f ₀ , the current on the resonant inductor L _r naturally starts from 0 and ends at 0, because the current flowing through the first rectifier diode D ₁ at this time is in line with the resonance. The current on the inductor L _r is the same current, and the current of the first rectifier diode D ₁ naturally starts from 0 and ends at 0, and the first rectifier diode D ₁ realizes soft switching. Therefore, in order to achieve soft switching of the first rectifier diode D ₁ , the conduction time of the switching tube Q ₁ cannot be less than half the resonance period 0.5/ f ₀ of the inductor L _r and the resonant capacitor C _r . In addition, when the resonant inductor L _r resonates with the resonant capacitor C _r for half the resonant period 0.5/ f ₀ , the average voltage on the resonant inductor L _r is 0, and the voltage on the resonant capacitor C _r V _Cr = V _{L 1_on} + V _{L 2_on} =( n +1) V ₁ .

When the switch tube Q1 is turned off _and the second rectifier diode D2 is turned on, since the current direction of the first inductor L1 cannot be changed, _the current of the second inductor L2 flows in the opposite direction from terminal ₃ to terminal 4 through coupling, thereby realizing bidirectional excitation of _the coupled inductor and making the coupled inductor have transformer function. Due to the reverse excitation of the coupled inductor, at this time, the voltage at terminal 1 _of the first inductor L1 is negative, and terminal 2 is positive, the voltage at terminal ₃ of the second inductor L2 is negative, and terminal 4 is positive, the first inductor L1 and _the second inductor L2 are in a series relationship, and the first _inductor L1 _{and the} second inductor L2 are jointly divided into V _o - V ₁ - _V Cr , where V _Cr is the voltage on the resonant capacitor C _r , and the voltage stress of the switch tube Q1 at this time is the sum of the voltage V _{L 1_off} on the first inductor and _the voltage V ₁ of the input power supply DC 1; since the resonant capacitor C _r plays an energy storage role in this mode, the voltage fluctuation on it is very small, and it can be approximately a voltage source characteristic, and the voltage on it remains unchanged as V _Cr =( n +1) V _1. In order to achieve this purpose, it is required that L _{r is} small and C _r is large to reduce the fluctuation range of V _Cr caused by _resonance when the switch tube Q1 is turned on. Therefore, in this embodiment, the inductance value of the coupled inductor Tr after the two inductors are connected in series is greater than or equal to 5 times the inductance L _r ; thus, the voltage V _L on the first inductor _{1_off} =( V _o - V _Cr - V ₁ )/(1+ n )=( V _o - V ₁ )/( n +1)- V ₁ , the voltage stress of the switch tube Q ₁ is ( V _o - V ₁ )/( n +1), which is reduced by ( n +1) times compared with the voltage stress V _o - V ₁ of the traditional solution.

Assuming that the inductor current is continuous, the turn-on duty cycle of the switch Q ₁ is D , and the turn-off duty cycle is 1- D. From the above analysis, it can be seen that when Q ₁ is turned on, the voltage of the first inductor of the coupled inductor V _{L 1_on} = V ₁ , when Q ₁ is turned off, the voltage of the first inductor of the coupling inductor V _{L 1_off} =( V _o - V _Cr - V ₁ )/(1+ n )=( V _o - V ₁ )/( n +1 )- V ₁ , according to the volt-second balance theorem, V _{L 1_on} D = V _{L 1_off} (1- D ), the boost gain can be obtained as:

;

The boost gain of the traditional non-isolated Boost boost circuit is G 1=1/(1- D ). The boost gain in the technical solution of the Chinese patent with patent number ZL201511000295.3 and publication date of 2018-06-01 ;

Because 0< D <1, and n≥1, the boost gain G of this embodiment is much larger than G 1 and G2 of the existing technical solution, that is, G > G ₂ > G ₁ ; therefore, the solution of this embodiment can boost the conversion The boost ratio of the converter achieves high-gain boost.

Simulation example:

Simulate the above technical solution of this embodiment, set the parameters n =4, V ₁ =25V, D =0.63, switching frequency fs =100K, C _r =2μF, L _r is 2μH, the first inductor L ₁ is 5μH, and the The second inductor L ₂ is 80 μH, and the load is 100 ohms. The output voltage calculated according to the above gain formula G is 362V, the output voltage calculated according to G ₂ is 237V, the output voltage calculated according to G ₁ is 67V, and the stable simulation voltage obtained according to the simulation is The average value Vo _is as high as 354V, as shown in Figure 3 (the abscissa in the figure is time in ms , the ordinate is voltage in V ), which is far more than the theoretical calculation value of the existing case, and is consistent with the theory mentioned above in this embodiment. The reason for the difference in calculated gain is the diode voltage drop, switch conduction resistance and drive duty cycle (rise rate, fall rate) loss, so in actual use, closed-loop control is required to adjust the duty cycle correction error to reach the required voltage.

In summary, the converter circuit of this implementation example has a higher boost gain than the prior art, avoiding the low efficiency of the traditional ultra-large duty cycle boost.

In this embodiment, by applying a fixed conduction time to the switch Q ₁ , the resonant inductor L _r and the resonant capacitor C _r can complete half-cycle resonance during the conduction period of the switch Q ₁ , so that the first rectifier diode D ₁ and switch transistor Q ₁ to achieve low loss as much as possible. By adjusting the driving off time of switch transistor Q ₁ , the switching frequency fs is changed, thereby changing the conduction ratio D of Q ₁ in the boost gain G , and achieving an increase in the output voltage. Different boost ratio requirements.

Example 2:

The converter structure of this embodiment is basically the same as that of Embodiment 1. The resonant inductor L _r , the coupling inductor Tr, the second inductor and the resonant capacitor Cr are connected in series. In this embodiment, the connection position of the resonant capacitor Cr in the series circuit can be The inductors L _r and L ₂ are exchanged, and their exchanged positions are shown in Figure 4. The negative terminal of the resonant capacitor Cr is connected to the terminal 2 of the inductor L ₁ and the drain of Q ₁ , and the positive terminal of Cr is connected to the terminal 3 of L ₂ . , terminal 3 of L2 is connected to one end of _the inductor Lr , and the other _{end of} the inductor Lr is connected to _the cathode of D1 and _the anode of D2 .

The above series branch can also be set as a resonant capacitor Cr _, a resonant inductor Lr and a second inductor L2 connected in _series in sequence, one end of the resonant capacitor Cr is connected to terminal 2 of the first inductor L1 , _the other _end is connected to one end of _the inductor Lr , and the other _end of the resonant inductor Lr is connected to terminal ₃ of the second inductor L2 ; or _the second inductor L2 , the resonant capacitor Cr and the resonant _inductor Lr are connected in series _in sequence, terminal 3 of _the second inductor _L2 is connected to terminal 2 of the first _inductor L1 , and terminal ₄ is connected to one end of the resonant capacitor Cr ; or the resonant _inductor Lr , the inductor L2 and the resonant capacitor Cr are connected in series in sequence, one end of the resonant _{inductor Lr} is connected to terminal 2 of the first inductor L1 , and the _other end is connected to terminal ₃ of the second loop L2 of the wire; the circuit function and principle of the converter in this embodiment are the same as those in the _first embodiment, so they are not repeated here.

Embodiment three:

As shown in Figure 5, this embodiment is based on Embodiment 1. Two voltage dividing capacitors C ₃ and C ₄ are connected between the positive terminal BUS+ and the negative terminal BUS- of the output terminal. The positive terminal of C ₃ is connected to the positive terminal of the output terminal. terminals are connected, the negative terminal of C3 is connected to the positive terminal of C4 , the negative terminal of C4 is connected to the negative terminal of the output; the drain of _the switch tube _Q1 is connected _to the anode of _the third rectifier diode D3 , and the _third rectifier The cathode of diode D ₃ is connected to the negative terminal of C _3. This point can be used as the intermediate output voltage terminal Mid+ , and is jointly connected with the output negative terminal BUS- to provide external power supply to the load. At the same time, the C ₄ capacitor can also absorb the peak voltage of the switching tube Q ₁ and play a voltage clamping role.

Simulation example:

Simulate the above technical solution of this embodiment, set the parameters n =4, V ₁ =25V, D =0.63, switching frequency f _s =100kHz, C _r =2μF, L _r is 2μH, the first inductor L ₁ is 5μH, The second inductor L ₂ is 80 μH, the voltage dividing capacitors C ₃ and C ₄ are both 110 μF, the load connected between the positive terminal BUS+ and the negative terminal BUS- at the output terminal is 100 ohms, and the load connected between the terminal Mid+ and the negative terminal BUS- is 100 ohms. As shown in Figure 6 (the abscissa in the figure is time _in ms , the ordinate is voltage in V ), the simulated output voltage Vo between the positive terminal BUS+ and the negative terminal BUS- is 354V, and the endpoint The output voltage V _mid between Mid+ and the negative terminal BUS- is 240V. It can be seen that the two voltages are output together through the voltage dividing capacitor.

Embodiment 4:

As shown in Figure 7, this embodiment is based on the second embodiment and connects two voltage dividing capacitors C3 and C4 between the positive terminal BUS+ and the negative terminal BUS- of _the output terminal. Similarly, the drain of the switch tube Q1 is connected to C3 The negative terminal and the positive terminal of C4 , this point can be used as the intermediate output voltage endpoint Mid+ , and is connected to the load together with the output negative terminal BUS- for external power supply.

The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. For those skilled in the art, the present invention may have various modifications and variations. Any modification, equivalent replacement, improvement, etc. made within the spirit and principle of the present invention shall be included in the protection scope of the present invention.

Claims (10)

1. A low-voltage non-isolated DC converter, including input power supply DC 1, input filter capacitor C ₁ , output filter capacitor C ₂ , coupling inductor Tr , switching tube Q ₁ and second rectifier diode D ₂ ; The input filter capacitor C ₁ is connected in parallel with the input power supply DC 1, and the output filter capacitor C ₂ is connected in parallel with the output terminal; The coupling inductor Tr includes a first inductor L ₁ and a second inductor L ₂ . The two ends of the first inductor L ₁ are endpoint 1 and endpoint 2 respectively. The endpoint 1 of the first inductor L ₁ is connected to the positive pole of the input power supply DC 1 . The cathode of the second rectifier diode D ₂ is connected to the positive terminal of the output; The drain of the switch tube Q1 is connected to the terminal 2 of the first inductor L1 , and the source is connected to the negative electrode of _the input power supply DC1 and the negative terminal of _the output; _It is characterized by: it also includes a resonant capacitor Cr , a resonant inductor Lr and a first rectifier diode D1 ; the inductance value of the two inductors L1 _{and L2 of} the _coupling inductor Tr connected in series is several times _the inductance value of the resonant inductor Lr ; The second inductor L ₂ , the resonant capacitor C _r and the resonant inductor L _r are connected in series with each other to form a series branch. One end of the series branch is connected to the end point 2 of the first inductor L ₁ and the other end is connected to the anode of the second rectifier diode D ₂ , the anode of _the first rectifier diode D1 is connected to the terminal 1 of L1 , and the cathode _is connected to _the anode of D2 ; The switch Q1 is turned off or turned on, causing _the currents in _the first inductor L1 and _the second inductor L2 to flow in the same direction or in the opposite direction, thereby realizing bidirectional excitation of the coupled inductor Tr . 2. The low-voltage non-isolated DC converter according to claim 1, characterized in that: the inductance value of the two inductors L 1 _and L ₂ connected in series of the coupling inductor Tr is greater than or equal to 5 times of the resonant inductor L _r . 3. The low-voltage non-isolated _DC converter according to claim 2, further comprising: a third rectifier diode D3 and a voltage dividing branch connected in parallel to the output end and formed by at least two voltage dividing capacitors in series. The anode of the three rectifier diodes D ₃ is connected to the drain of the switching tube Q ₁ , and the cathode is connected to a voltage dividing end of the voltage dividing branch. 4. The low-voltage non-isolated DC converter according to claim 3, characterized in that: the two ends of the second inductor L2 are endpoint 3 and endpoint 4 respectively; _the endpoint 1 of _the first inductor L1 and the second inductor The terminal 3 of L ₂ is the terminal of the same name, or the terminal 2 of the first inductor L ₁ and the terminal 4 of the second inductor L ₂ are terminals of the same name. 5. The low-voltage non-isolated DC converter according to claim 4, characterized in that: the second inductor L ₂ , the resonant inductor L _r and the resonant capacitor C _r are connected in series in sequence, and the end point 3 of the second inductor L ₂ is connected to the second inductor L 2 . The terminal 2 and terminal 4 of an inductor L ₁ are connected to one end of the resonant inductor L _r . 6. The low-voltage non-isolated DC converter according to claim 4, characterized in that: the resonant capacitor C _r , the second inductor L ₂ and the resonant inductor L _r are connected in series in sequence, and one end of the resonant capacitor C _r is connected to the first inductor. Terminal 2 of L ₁ , and the other end is connected to terminal 3 of the second inductor L ₂ . 7. The low-voltage non-isolated DC converter according to claim 4, characterized in that: the resonant capacitor C _r , the resonant inductor L _r and the second inductor L ₂ are connected in series in sequence, and one end of the resonant capacitor C _r is connected to the first inductor. The endpoint 2 of L1 is connected to one end of _the resonant inductor Lr , and the other _end of the resonant _inductor Lr is connected to the endpoint ₃ of the second inductor L2 . 8. The low-voltage non-isolated DC converter according to claim ₄ , characterized in that: the second inductor L2 , _the resonant capacitor Cr and the resonant inductor _Lr are connected in series in sequence, the terminal 3 of the second inductor L2 is connected to the _terminal 2 of the first _inductor L1 , and the terminal 4 is connected to one end of _{the resonant capacitor Cr} . 9. The low-voltage non-isolated DC converter according to claim 3, characterized in that: the resonant inductor L _r , the second inductor L ₂ and the resonant capacitor C _r are connected in series in sequence, and one end of the resonant inductor L _r is connected to the first The terminal 2 of an inductor L ₁ and the other end are connected to the terminal 3 of the second inductor L ₂ . 10. A control method for a low-voltage non-isolated DC converter, used to control the low-voltage non-isolated DC converter according to any one of claims 1 to 9, characterized in that: the control method specifically includes: Apply a fixed conduction time to the switching tube Q ₁ so that the resonant inductor L _r and the resonant capacitor C _r complete half-cycle resonance during the conduction period of Q ₁ ; By adjusting the switching frequency f _s of Q ₁ to change the conduction ratio of Q ₁ , the change in the boost ratio of the converter output voltage compared to the input voltage is achieved.