CN114759792B - A single-stage high-gain modular multi-level resonant DC boost converter - Google Patents

Abstract

The invention discloses a single-stage high-gain modularized multi-level resonant direct-current boost converter, and belongs to the technical field of power generation, power transformation or power distribution. The converter comprises M switch inductance units, a quasi-switch boost network, N switch capacitance units, K upper bridge units and a lower bridge unit, wherein the upper bridge unit and the lower bridge unit are sequentially connected in series, the branches formed by connecting an output inductance and an output filter capacitance in series are connected at two ends of the bridge arms in parallel, K units are input in total by controlling only one unit in the upper bridge units in a non-input mode, the K input units are in series resonance with the output inductance and the output filter capacitance, the switching stress of the device is small, the nesting number of the switch inductance units, the nesting number of the switch capacitance units and the number of the upper bridge units connected in series are changed, so that the converter can obtain flexibly variable high voltage gain and high step-up ratio. The direct current boost converter can work stably under the open loop condition, and meanwhile, self-balancing of the capacitor voltages of the upper bridge unit and the lower bridge unit is realized.

Description

A single-stage high-gain modular multi-level resonant DC boost converter

Technical Field

The present invention relates to high voltage direct current converter technology, in particular to a single-stage high gain modular multi-level resonant direct current boost converter applied to medium voltage direct current transmission technology, belonging to the technical field of power generation, power transformation or power distribution.

Background technique

In today's renewable energy power generation systems, the DC voltage provided by a single solar cell or a single fuel cell is low, which is difficult to meet the power voltage requirements of existing electrical equipment. Therefore, it is necessary to introduce a high voltage gain DC converter in the renewable energy power generation system to boost the low input voltage on the DC side. At the same time, since medium-voltage DC transmission has incomparable advantages and broad application prospects in large-capacity long-distance transmission and new energy power generation, the development and improvement of high voltage gain DC-DC converters have attracted the attention of many scholars at home and abroad. However, current research is focused on medium and low voltage DC-DC converters, and DC converters that can be used in high voltage applications are still under study. In this context, modular multi-level converters with variable high gain and self-balancing of upper and lower bridge unit capacitor voltages are increasingly attracting attention.

The prior art improves the voltage gain of the modular multi-level converter by changing the number of upper bridge units and the inductor charging ratio, but the boost capability is limited. In order to realize the self-balancing of the capacitor voltage of the upper and lower bridge units of the modular multi-level converter, the prior art needs to perform redundant operations on the lower bridge units in the lower bridge arm and relies on complex control algorithms, which increases the size and cost of the converter.

It can be seen that how to adjust the gain according to the actual application of the modular multi-level DC converter and achieve self-balancing of the upper and lower arm unit capacitor voltages through a simple control method is a technical problem that needs to be solved urgently.

Summary of the invention

The invention aims to address the deficiencies of the above-mentioned background technology and propose a single-stage high-gain modular multi-level resonant DC boost converter suitable for small and medium-power new energy access occasions and flexible DC transmission systems. The quasi-switch boost network is used as the boost unit. By adjusting the number M of switch inductor units in the power supply circuit on the low-voltage input side of the quasi-switch boost network and the number N of switch capacitor units connected in series on the output side of the quasi-switch boost network, the boost ratio of the DC booster is greatly improved, and a variable high voltage gain is achieved. Through the carrier phase-shift pulse width modulation strategy, only one upper bridge unit is not put into use in the inductor energy release mode, so that the same number of bridge arm unit capacitors are connected in series with the output inductor and the output filter capacitor in each working cycle, and the number of lower bridge arm unit configurations is reduced. At the same time, the self-balancing of the upper and lower bridge unit capacitor voltages when the DC booster is in open-loop operation can be achieved, and the volume and cost of the converter are reduced. The converter effectively solves the problems of low gain, large volume, and high output voltage harmonics of traditional DC converters.

The present invention adopts the following technical solutions to achieve the above-mentioned invention object:

A single-stage high-gain modular multi-level resonant DC boost converter comprises: a first inductor and M switch inductor units, a quasi-switch boost network, N switch capacitor units, an upper bridge arm and a lower bridge arm connected in series with the upper bridge arm, an output inductor and an output filter capacitor; the first inductor and the M switch inductor units are used as energy storage elements on the low-voltage input side of the DC boost converter, and the inductors in the M switch inductor units are connected in series or in parallel with the first inductor by controlling the switches in the switch inductor units; the input side of the quasi-switch boost network is connected in series in a DC power supply circuit on the low-voltage side; the output side of the quasi-switch boost network is connected in series in a DC power supply circuit on the low-voltage side; the output side of the quasi-switch boost network is connected in series in a DC power supply circuit on the low-voltage side by controlling the switch in the switch inductor units ... The switches in the switch capacitor unit enable the capacitors in the N switch capacitor units to be connected in series and parallel and then connected between the drain of the first IGBT in the quasi-switch boost network and the midpoint of the output bridge arm. The capacitance value between the drain of the first IGBT in the quasi-switch boost network and the midpoint of the output bridge arm is adjusted by adjusting the number N of the switch capacitor units. The upper bridge arm includes K upper bridge units connected in series in sequence, and the lower bridge arm includes 1 lower bridge unit. The upper bridge arm and the lower bridge arm are connected in series to form the output bridge arm of the DC boost converter, and the branch after the output inductor and the output filter capacitor are connected in series is connected in parallel at both ends of the output bridge arm.

The circuit structures of the M switching inductor units are the same. The Mth switching inductor unit includes the 3M-2th diode, the 3M-1th diode, the 3Mth diode and the M+1th inductor, wherein the cathode of the 3M-2th diode, the cathode of the 3Mth diode and one end of the M+1th inductor are connected to serve as port 4 of the Mth switching inductor unit, the anode of the 3M-1th diode is connected to the anode of the 3Mth diode to serve as port 3 of the Mth switching inductor unit, the cathode of the 3M-1th diode is connected to the other end of the M+1th inductor to serve as port 2 of the Mth switching inductor unit, and the 3M-2th diode is connected to the cathode of the 3Mth diode to serve as port 4 of the Mth switching inductor unit. The anode of the -2 diode serves as port 1 of the Mth switching inductor unit; one end of the first inductor is connected to port 1 of the first switching inductor unit and the positive terminal of the low-voltage side voltage source, and the other end of the first inductor is connected to port 3 of the first switching inductor unit; thereafter, port 1 and port 3 of the i+1th switching inductor unit are respectively connected to port 4 and port 2 of the i-th switching inductor unit, and the M switching inductor units are nested according to this principle. Finally, port 2 of the Mth switching inductor unit is connected to the drain of the first IGBT in the quasi-switch boost network, 1≤i≤M, and M is an integer greater than 0.

The quasi-switch boost network includes a first IGBT, a fourth diode, a fifth diode and a fifth capacitor, wherein the drain of the first IGBT and the anode of the fifth diode are connected to port 2 of the Nth switch inductor unit; the source of the first IGBT is connected to the anode of the fourth diode and the negative plate of the fifth capacitor; the cathode of the fifth diode is connected to the positive plate of the fifth capacitor; the cathode of the fourth diode is connected to the negative terminal of the low-voltage side voltage source;

The circuit structures of N switch capacitor units are the same. The Nth switch capacitor unit includes a capacitor and a diode. The positive plate of the capacitor is connected to the cathode of the diode. The negative plate of the capacitor of the first switch capacitor unit is connected to the drain of the first IGBT in the quasi-switch boost network. The anode of the diode of the first switch capacitor unit is connected to the positive plate of the fifth capacitor. The connection point between the positive plate of the capacitor and the cathode of the diode in the first switch capacitor unit is connected to the anode of the diode in the second switch capacitor unit and the negative plate of the capacitor in the third switch capacitor unit. The negative plate of the capacitor in the second switch capacitor unit is connected to the drain of the first IGBT in the quasi-switch boost network. The positive plate of the capacitor; the connection point between the positive plate of the capacitor in the second switch capacitor unit and the cathode of the diode is connected to the anode of the diode in the third switch capacitor unit and the negative plate of the capacitor in the fourth switch capacitor unit; the connection point between the positive plate of the capacitor in the third switch capacitor unit and the cathode of the diode is connected to the anode of the diode in the fourth switch capacitor unit and the negative plate of the capacitor in the fifth switch capacitor unit; the fourth switch capacitor unit to the N-1th switch capacitor unit are connected according to the above principle, and the connection point between the positive plate of the capacitor and the cathode of the diode in the Nth switch capacitor unit is connected to the connection point of the upper and lower bridge arms of the converter.

The Kth upper bridge unit includes a Kth upper bridge IGBT, a Kth upper bridge diode and a Kth upper bridge capacitor, wherein an anti-parallel diode is inside the Kth upper bridge IGBT; the source of the Kth upper bridge IGBT is connected to the cathode of the Kth upper bridge diode, the drain of the Kth upper bridge IGBT is connected to the positive plate of the Kth upper bridge capacitor; the negative plate of the Kth upper bridge capacitor is connected to the anode of the Kth upper bridge diode; the source of the Kth upper bridge IGBT serves as the input end of the Kth upper bridge unit, and the negative plate of the Kth upper bridge capacitor serves as the output end of the Kth upper bridge unit.

The lower bridge unit includes a first lower bridge IGBT, a second lower bridge IGBT and a lower bridge capacitor, wherein the first lower bridge IGBT and the second lower bridge IGBT are both anti-parallel diodes; the drain of the first lower bridge IGBT is connected to the positive plate of the lower bridge capacitor, the source of the first lower bridge IGBT is connected to the drain of the second lower bridge IGBT; the source of the second lower bridge IGBT is connected to the negative plate of the lower bridge capacitor; the source of the first lower bridge IGBT serves as the input end of the lower bridge unit, and the source of the second lower bridge IGBT serves as the output end of the lower bridge unit.

The input end of the first upper bridge unit is connected to one end of the output inductor, the output end of the pth upper bridge unit is connected to the input end of the p+1th upper bridge unit, the output end of the p+1th upper bridge unit is connected to the input end of the p+2th upper bridge unit, the pth upper bridge unit to the Kth upper bridge unit are connected in series in sequence, 1≤p≤K, K is an integer greater than 0, the output end of the Kth upper bridge unit is connected to the input end of the lower bridge unit, the other end of the output inductor is connected to the positive plate of the output filter capacitor, the output end of the lower bridge unit, the cathode of the fourth diode in the quasi-switch boost network, the negative electrode of the low-voltage side voltage source, and the negative plate of the output filter capacitor are connected; a load is connected between the two plates of the output filter capacitor, and the capacitance value of the output filter capacitor is much larger than the capacitance values of the upper bridge capacitor and the lower bridge capacitor.

The lower bridge arm can adopt multiple lower bridge units connected in series in sequence for redundancy processing. As a further optimization scheme of a single-stage high-gain modular multi-level resonant DC boost converter of the present invention, the lower bridge arm adopts one lower bridge unit. At this time, the converter can achieve self-balancing of the capacitor voltage of the upper bridge unit and the lower bridge unit under open-loop operation.

As a further optimization scheme of a single-stage high-gain modular multi-level resonant DC boost converter of the present invention, a carrier phase-shift pulse width modulation strategy is adopted to control the charging and discharging states of the first inductor, the inductance of M switching inductor units, the capacitor in the quasi-switch boost network, the capacitor of N switching capacitor units, the upper bridge capacitors in K upper bridge units, and the lower bridge capacitor in 1 lower bridge unit.

Furthermore, a driving method of a single-stage high-gain modular multi-level resonant DC boost converter is provided, wherein a switching cycle T _s is divided into K working cycles _Te , _Te =T _s /K. In each working cycle, the converter operates in two modes, the first IGBT and the second lower bridge IGBT are turned on to make the converter operate in an inductive energy storage mode, and when the inductive energy storage mode lasts for a time duration of d* _Te , the first IGBT and the second lower bridge IGBT are turned off to make the converter operate in an inductive energy release mode.

In each working cycle, the driving method of the converter includes the following steps:

Step 1), let j = 1;

Step 2), calculate P = mod (j + K, K), mod is the modulo function;

Step 3), turn off the first lower bridge IGBT of the lower bridge unit, turn on the second lower bridge IGBT of the lower bridge unit, the 3M-2 diode and the 3M-1 diode in the switch inductor unit are turned on under the forward voltage drop, and the 3M diode is turned off under the reverse voltage drop; turn on the first IGBT in the quasi-switch boost network, and the fourth diode and the fifth diode are turned off under the reverse voltage drop; the diode in the switch capacitor unit is turned on under the forward voltage drop; the external low-voltage side voltage source charges and stores energy for the first inductor, the M+1 inductor in the switch inductor unit, and the capacitor in the switch capacitor unit, respectively, and the converter is in the inductor energy storage mode; in this mode, the upper bridge IGBTs in all upper bridge units are turned on, the upper bridge diodes in all upper bridge units are turned off under the reverse voltage drop, and the upper bridge capacitors in the K upper bridge units work in series resonance with the output inductor and the output filter capacitor. Since the output filter capacitor value is much larger than the capacitance value of the upper bridge capacitor in the upper bridge unit, the resonant frequency is determined by the output inductor and the upper bridge capacitors in the K upper bridge units, and the output filter capacitor releases energy to power the load;

Step 4), turn off the first IGBT in the quasi-switch boost network, the 3Mth diode in the switch inductor unit is turned on by the forward voltage drop, the 3M-2th diode and the 3M-1th diode are turned off by the reverse voltage drop, turn off the second lower bridge IGBT of the lower bridge unit, turn on the first lower bridge IGBT of the lower bridge unit, turn off the upper bridge IGBT of the Pth upper bridge unit, turn on the upper bridge IGBT of the remaining upper bridge units, the upper bridge diode of the Pth upper bridge unit is turned on by the forward voltage drop, and the upper bridge diodes of the remaining upper bridge units are turned off by the reverse voltage, so that the upper bridge capacitors of the upper bridge units except the Pth upper bridge unit, the lower bridge capacitors of the lower bridge units, the output inductor and the output filter capacitor work in series resonance; since the output filter capacitor value is much larger than the upper bridge capacitor and the lower bridge capacitor value, the resonant frequency is determined by the upper bridge capacitor and the output inductor of the upper bridge units except the Pth upper bridge unit, and when the capacitance values of the upper bridge capacitor and the lower bridge capacitor are equal, the resonant frequency is the same as the resonant frequency of step 3);

Step 5), let j=j+1, and jump to step 2).

When the upper bridge IGBT of the first upper bridge unit is selected to be turned off, the phase shift of the IGBT of the first upper bridge unit is 0, and the phase shift of the IGBT of the remaining upper bridge units is based on the phase shift of the IGBT of the first upper bridge unit; when the upper bridge IGBT of the second upper bridge unit is selected to be turned off, the phase shift of the upper bridge IGBT of the second upper bridge unit is 2π/K; when the upper bridge IGBT of the third upper bridge unit is selected to be turned off, the phase shift of the upper bridge IGBT of the third upper bridge unit is 2*2π/K; when the upper bridge IGBT of the Pth upper bridge unit is selected to be turned off, the phase shift of the upper bridge IGBT of the Pth upper bridge unit is (P-1)*2π/K; when the upper bridge IGBT of the Kth upper bridge unit is selected to be turned off, the phase shift of the upper bridge IGBT of the Kth upper bridge unit is (K-1)*2π/K. The inductor energy release mode controls the trigger signals of the upper bridge IGBTs in all upper bridge units through a carrier phase-shift pulse width modulation strategy so that K-1 upper bridge capacitors are always put into the upper bridge arm in the inductor energy release mode of each working cycle.

Compared with the prior art, the present invention adopts the above technical solution and has the following technical effects:

(1) The present invention discloses a single-stage high-gain modular multi-level resonant DC boost converter, which combines switching inductors, switching capacitors, and quasi-switching boost network technologies. Under the same input voltage and duty cycle, variable high voltage gain can be achieved by changing the number of nested switching inductor units, switching capacitor units, and the number of half-bridge sub-modules connected in series on the upper bridge arm.

(2) The present invention sequentially switches the switch tubes of the upper bridge unit in the upper bridge arm through a carrier phase-shift pulse width modulation strategy, thereby changing the number of capacitors connected in series in the upper bridge arm. The capacitors put into the upper bridge arm match the capacitors put into the lower bridge arm, so that in each working cycle, there are always K capacitors that resonate in series with the output inductor and the output filter capacitor. In the case of non-redundant operation with only one lower bridge unit, the capacitor voltages of the upper and lower bridge units are self-balanced, the switching stress of the device is reduced, the control method is simple, and the size and cost of the converter are reduced.

(3) The converter disclosed in the present invention has continuous input current on the low-voltage side and continuous load current, and has broad application prospects in the field of small and medium-power medium-voltage DC power transmission and distribution.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG1 is a block diagram of a single-stage high-gain modular multi-level resonant DC boost converter system configured with M switch inductor units, N switch capacitor units, and K upper bridge units according to the present invention.

FIG2 is a topological structure diagram of a single-stage high-gain modular multi-level resonant DC boost converter with M=1, N=1, and K=3 in an embodiment of the present invention.

Figure 3(a) is a circuit diagram of a single-stage high-gain modular multi-level resonant DC boost converter with M=1, N=1, K=3 in working mode 1 in an embodiment of the present invention, and Figure 3(b) is a circuit diagram of a single-stage high-gain modular multi-level resonant DC boost converter with M=1, N=1, K=3 in working mode 2 in an embodiment of the present invention.

FIG4 is a simulation waveform diagram of a single-stage high-gain modular multi-level resonant DC boost converter with M=1, N=1, and K=3 in an embodiment of the present invention when Vi=25V, D=11/15, d=0.2, _L1 = _L2 =1.5mH, C=50μF, _LS =1.76μH, C _H =2200μF, R=1500Ω, and a switching frequency _fs =10kHz. FIG4(a) is a simulation waveform diagram of the output voltage V _i (i=1,2,3,4) of each unit of the upper and lower bridge arms of the converter. FIG4(b) is a simulation waveform diagram of the voltages V _C5 and V C6 of capacitors C ₅ and C ₆ in the second boost circuit of the converter. FIG4(c) is a simulation waveform diagram of the output voltage V _o of the high-voltage side of the converter. FIG4(d) is a simulation waveform diagram of the currents i _L1 and i _L6 of the inductors L ₁ and L ₂ in the converter. _FIG4 (e) is a simulation waveform diagram of the converter load current i _o .

Explanation of the symbols in the figure: Vi _is the low-voltage side input voltage source, Cell ^Mth is the Mth switching inductor unit, _L1 and _L2 are the first and second inductors, L _M+1 is the M+1th inductor, _D1 , _D2 , _D4 , _D5 , _D6 are the first, second, fourth, fifth, and sixth diodes, _D3M , _D3M-1 , _D3M-2 are the 3M, 3M-1, and 3M-2 diodes, _C1 ~ _C3 are the first to third upper bridge capacitors, _C4 is the lower bridge capacitor, _C5 is the fifth capacitor, _S0 is the first IGBT, _S1 ~ _S3 are the first to third upper bridge IGBTs, _S4 and _S5 are the first and second lower bridge IGBTs, _Ls is the output inductor, Co is the output filter capacitor, and _Vo is the high-voltage side output voltage.

Detailed ways

The technical solution of the invention is described in detail below with reference to the accompanying drawings.

The technical solution disclosed in the present invention can be implemented in many different forms, and the protection scope of the present invention should not be considered to be limited to the embodiments described herein. On the contrary, these embodiments are provided to fully disclose the present invention so that those skilled in the art can fully understand the technical solution of the present invention through the disclosed embodiments. In order to clearly show the specific circuit structure of the converter of the present invention, the specific circuit structure of the switch inductor unit and the switch capacitor unit is partially enlarged in the accompanying drawings.

It should be understood that, although the terms first, second, third, etc. may be used to describe various elements, components and/or parts, these elements, components and/or parts are not limited by these terms. These terms are only used to distinguish elements, components and/or parts from each other. Therefore, the first element, component and/or part discussed below can be expressed as a second element, component or part without departing from the purpose of the present invention.

In this embodiment, the number of switching inductor units is set to M; the number of switching capacitor units is set to N; the number of upper bridge units in the upper bridge arm is set to K; the lower bridge units in the lower bridge arm can be set to multiple to achieve redundancy, but because the number of lower bridge units in the lower bridge arm has no effect on the gain under the control strategy of the present invention, in order to reduce the volume, weight and cost of the converter, and at the same time to achieve self-balancing of the capacitor voltages of the upper and lower bridge units, the present invention uses a lower bridge arm of a lower bridge unit, and takes M=1, N=1, K=3 as an example to describe in detail the single-stage high-gain modular multi-level resonant DC boost converter of the present invention.

Referring to FIG1 , the present invention discloses a single-stage high-gain modular multi-level resonant DC boost converter circuit, which includes: a first inductor L ₁ , M switch inductor units (SL Cell) nested with the first inductor L ₁ , a quasi-switch boost network, N switch capacitor units (SC Cell), an upper bridge arm, and a lower bridge arm connected in series with the upper bridge arm, and an output circuit composed of an output inductor and an output filter capacitor. The upper bridge arm is connected in series with K upper bridge units, and the K upper bridge units are respectively submodule 1, submodule 2, ..., submodule K, and submodule 1, submodule 2, ..., submodule K are all composed of a half-bridge circuit of the same structure, in which an upper bridge IGBT of an anti-parallel diode is connected in series with a diode in reverse series, and the two ends of the reverse series branch are connected in parallel with a capacitor; the lower bridge arm adopts a lower bridge unit, and the lower bridge unit is composed of a half-bridge circuit of another structure, in which two IGBTs of anti-parallel diodes are connected in series, and the two ends of the series branch are connected in parallel with a capacitor.

In a single-stage high-gain modular multi-level resonant DC boost converter circuit shown in FIG1 , the connection relationship of the energy storage elements in the low-voltage side power supply circuit is as follows: the positive electrode of the low-voltage side voltage source V _i is connected to one end of the first inductor L ₁ , the first inductor L ₁ is nested with M switching inductor units, the Mth switching inductor unit includes the 3M-2th diode D _3M-2 , the 3M-1th diode D _3M-1 , the 3Mth diode D _3M and the M+1th inductor, wherein the cathode of the 3M-2th diode D _3M-2 , the cathode of the 3Mth diode and one end of the M+1th inductor L _M+1 are connected as port 4 of the Mth switching inductor unit, the anode of the 3M-1th diode D _3M-1 is connected to the anode of the 3Mth diode D _3M as port 3 of the Mth switching inductor unit, the cathode of the 3M-1th diode D _3M-1 is connected to the M+1th inductor L M+1 The other end of _M+1 is connected to serve as port 2 of the Mth switching inductor unit, the anode of the 3M-2th diode D _3M-2 serves as port 1 of the Mth switching inductor unit, one end of the first inductor L ₁ is connected to port 1 of the first switching inductor unit and the positive terminal of the low-voltage side voltage source _Vi , the other end of the first inductor L ₁ is connected to port 3 of the first switching inductor unit, and port 2 of the Mth switching inductor unit is connected to the drain of the first IGBT S ₀ in the quasi-switch boost network and the anode of the fifth diode D ₅ .

In a single-stage high-gain modular multi-level resonant DC boost converter circuit shown in FIG1 , the connection relationship of the components in the quasi-switch boost network is as follows: the quasi-switch boost network includes a first IGBT S ₀ , a fourth diode D ₄ , a fifth diode D ₅ and a fifth capacitor C ₅ , wherein the drain of the first IGBT S ₀ and the anode of the fifth diode D ₅ are connected to port 2 of the Nth switching inductor unit; the source of the first IGBT S ₀ is connected to the anode of the fourth diode D ₄ and the negative plate of the fifth capacitor C ₅ ; the cathode of the fifth diode D ₅ is connected to the positive plate of the fifth capacitor C ₅ ; and the cathode of the fourth diode D ₄ is connected to the negative terminal of the low-voltage side voltage source V _i .

In a single-stage high-gain modular multi-level resonant DC boost converter circuit shown in FIG1 , the connection relationship of each device in the switch capacitor unit is as follows: each switch capacitor unit includes a capacitor and a diode, and the positive plate of the capacitor is connected to the cathode of the diode, the negative plate of the capacitor of the first switch capacitor unit is connected to the drain of the first IGBT S ₀ in the quasi-switch boost network, the anode of the diode of the first switch capacitor unit is connected to the positive plate of the fifth capacitor C ₅ , the connection point between the positive plate of the capacitor and the cathode of the diode in the first switch capacitor unit is connected to the anode of the diode of the second switch capacitor unit and the negative plate of the capacitor of the third switch capacitor unit, and the negative plate of the capacitor in the second switch capacitor unit is connected to the fifth capacitor C ₅ , the connection point between the capacitor positive plate of the second switching capacitor unit and the cathode of the diode is connected to the anode of the diode of the third switching capacitor unit and the capacitor negative plate of the fourth switching capacitor unit; the connection point between the capacitor positive plate of the third switching capacitor unit and the cathode of the diode is connected to the anode of the diode of the fourth switching capacitor unit and the capacitor negative plate of the fifth switching capacitor unit, the fourth switching capacitor unit to the N-1th switching capacitor unit are connected according to the above principle, and the connection point between the capacitor positive plate of the Nth switching capacitor unit and the cathode of the diode is connected to the connection point of the upper and lower bridge arms of the converter.

In a single-stage high-gain modular multi-level resonant DC boost converter circuit shown in FIG1 , the connection relationship between the devices in the upper and lower bridge arms of the converter is as follows: the upper bridge arm of the converter is connected in series with K identical upper bridge units, and the upper bridge units are represented as submodule 1, submodule 2, ..., submodule K, respectively. Each upper bridge unit includes an upper bridge IGBT, an upper bridge diode and an upper bridge capacitor. An anti-parallel diode is connected inside the upper bridge IGBT. The source of the upper bridge IGBT is connected to the cathode of the upper bridge diode, the drain of the upper bridge IGBT is connected to the positive plate of the upper bridge capacitor, the negative plate of the upper bridge capacitor is connected to the anode of the upper bridge diode, the source of the upper bridge IGBT serves as the input end of the upper bridge unit, and the negative plate of the upper bridge capacitor serves as the upper bridge unit. The output end of the lower bridge arm adopts a lower bridge unit, the lower bridge unit includes a first lower bridge IGBT, a second lower bridge IGBT and a lower bridge capacitor, the first lower bridge IGBT and the second lower bridge IGBT are both anti-parallel diodes, the drain of the first lower bridge IGBT is connected to the positive plate of the lower bridge capacitor, the source of the first lower bridge IGBT is connected to the drain of the second lower bridge IGBT; the source of the second lower bridge IGBT is connected to the negative plate of the lower bridge capacitor, the source of the first lower bridge IGBT serves as the input end of the lower bridge unit, and the source of the second lower bridge IGBT serves as the output end of the lower bridge unit; the output port of the upper bridge unit K is connected to the input port of the lower bridge unit, and the connection point is the connection point of the upper and lower bridge arms; the output inductor L One end of _s is connected to the input port of the first upper bridge unit, and the other end of the output inductor _Ls is connected to the positive plate of the output filter capacitor Co; the negative electrode of the low-voltage side voltage source _Vi is connected to the cathode of the fourth diode _D4 in the quasi-switch boost network, the source of the second lower bridge IGBT in the lower bridge unit and the negative plate of the filter capacitor Co, and the load is connected in parallel at both ends of the filter capacitor.

To illustrate the working principle of the converter, this embodiment selects M=1, N=1, K=3. Referring to FIG2, the upper and lower bridge units are named as: submodule 1, submodule 2, submodule 3, submodule 4, and the capacitance values of the submodule capacitors _C1 , _C2 , _C3 , and _C4 are set equal. Therefore, in steady state, the average value of the DC side capacitor voltage of each submodule is equal. The output voltage value is equal to the sum of the series capacitor voltage values of the upper and lower bridge arms input circuit.

In the present invention, the switching frequency of the switch tube and the lower bridge unit switch tube in the quasi-switch boost network is _fe , the switching frequency of the upper bridge unit switch tube is _fs , the converter operating frequency is _fe , and the converter operating frequency _fe is equal to K times the upper bridge unit switching frequency _fs . Therefore, within one switching cycle _Ts , there are K working cycles _Te , and each working cycle _Te = _Ts / K. In this embodiment, K = 3, so _fe = _3fs , _Te = _Ts / 3.

In one embodiment of the circuit of the present invention, as shown in FIG2 , the working mode and specific implementation method of a working cycle are as follows:

Mode 1: Inductor Energy Storage Mode

As shown in FIG3(a), the first IGBT _S0 and the second lower bridge IGBT _S5 of the lower bridge arm submodule 4 in the quasi-switch boost network are turned on. At this time, the lower bridge capacitor _C4 is bypassed, the first diode _D1 , the second diode _D2 and the sixth diode _D6 are turned on under the forward voltage drop, the third diode _D3 , the fourth diode _D4 and the fifth diode _D5 are turned off under the reverse voltage drop, and the low-voltage side voltage source forms a loop through _D1 - _S0 - _C5 - _D6 - _S5 and _D2 - _S0 - _C5 - _D6 - _S5 to charge the first inductor _L1 and the second inductor _L2 ; the sixth capacitor _C6 forms a loop through _S0 - _C5 - _D6 to absorb energy from the fifth capacitor _C5 ; obviously, the first inductor _L1 , the second inductor _L2 and the sixth capacitor C ₆ The energy stored in the mode 1 stage is related to the charging time. Let the time proportion of mode 1 in a working cycle be d (d=mode 1 duration/working cycle _Te ). At the same time, the first upper bridge IGBT _S1 , the second upper bridge IGBTS2, and the third upper bridge IGBTS3 in the upper bridge arm submodule 1, submodule ₂ , and submodule ₃ are turned on, and the diodes reversely connected to the first upper bridge IGBT _S1 , the second upper bridge _IGBTS2 , and the third upper bridge _IGBTS3 are cut off due to reverse voltage. The three capacitors of the upper bridge arm, the first to the third upper bridge capacitors _C1 ~ _C3 , the output inductor _Ls , and the output filter capacitor _CO form a series resonant circuit. Since the capacitance value of the output filter capacitor C _O is much larger than the capacitance values of the first upper bridge capacitor C ₁ , the second upper bridge capacitor C ₂ , and the third upper bridge capacitor C ₃ , the filter capacitor C _O can be ignored when C ₁ , C ₂ , C ₃ , C _O , and _LS are in series resonance. Therefore, the resonant frequency of the resonant circuit of this embodiment is:

In this mode, the output voltage on the high-voltage side is equal to the sum of the capacitor voltages of the three sub-modules on the upper bridge arm.

Mode 2: Inductor energy release mode

As shown in FIG3(b), the first IGBT _S0 and the second lower bridge IGBT _S5 of the lower bridge arm submodule 4 in the quasi-switch network are turned off, and the first lower bridge IGBT S4 of the lower bridge arm submodule ₄ is turned on. At this time, the lower bridge capacitor _C4 is put into the circuit, the third diode _D3 , the fourth diode _D4 and the fifth diode _D5 are subjected to a forward voltage drop, the first diode _D1 , the second diode _D2 and the sixth diode _D6 are subjected to a reverse voltage drop and are turned off, the first inductor _L1 and the second inductor _L2 are connected in series to release energy, and the low-voltage side input voltage source V _i and the first inductor _L1 and the second inductor _L2 are connected in series to form a loop through _D3 - _D5 - _D4 to charge the fifth capacitor _C5 ; in addition, the low-voltage side voltage source V _i , the first inductor _L1 , the second inductor _L2 and the sixth capacitor _C6 are connected in series to form a loop through _D3 - _S4 to charge the lower bridge capacitor _C4 in the lower bridge unit; at the same time, the first upper bridge IGBT _S1 in the turned-off submodule 1 is turned off. , the diode connected in reverse series with the first upper bridge IGBT _S1 is subjected to a forward voltage drop and is turned on, and the first upper bridge capacitor _C1 in submodule 1 is bypassed; the second upper bridge capacitor _C2 , the third upper bridge capacitor C3, the fourth lower bridge capacitor _C4 in submodule 2, submodule 3, and submodule ₄ form a series resonant circuit with the output inductor _LS and the output filter capacitor _CO . Since the number of capacitors working in series resonance is the same as that in mode one, the resonant frequency in this mode is the same as that in mode one.

In mode 1, capacitors _C1 , _C2 , and _C3 are connected in series to supply power to the output terminal _Vo , and the voltage across the capacitors is _Vcj (j=1,2,3), so the high-voltage side voltage _Vo can be expressed as: _Vo =V _C1 +V _C2 +V _C3 ; in mode 2, capacitors _C2 , _C3 , and _C4 are connected in series to supply power to the output terminal _Vo , so the high-voltage side voltage _Vo can be expressed as: _Vo =V _C2 +V _C3 +V _C4 . By comparing the above two formulas, it can be seen that: V _C1 =V _C4 . Combined with the working state of the above working cycle, it can be concluded that the voltage across the capacitors connected to the upper and lower bridge arms of the loop is equal, achieving the internal balance of the capacitor voltage, that is: V _C1 =V _C2 =V _C3 =V _C4 . Therefore, in the entire working cycle, the high-voltage side voltage _Vo is constant: V _o =3V _C.

In one working cycle, the energy stored in the energy storage inductor L _N (N＝1,2) in mode 1 should be equal to the energy released in mode 2. According to the volt-second balance theorem, we get:

In formula (2), d=1-3(1-D), where d is the charging ratio of inductors _L1 and _L2 , that is, the duty cycle of the first IGBT _S0 and the second lower bridge IGBTS ₅ in one working cycle; and D is the duty cycle of the first upper bridge IGBT _S1 , the first upper bridge IGBTS ₂ , and the first upper bridge IGBT _S3 in the upper bridge arm submodule.

From formula (2), we can get:

In formula (3), 0<d<1/3, combined with d=1-3(1-D), 2/3<D<7/9, since the voltages of capacitors _C5 and _C6 are equal in the steady state, that is, _VC5 = _VC6 , and _VC4 = _VC5 + _VC6 , the gain factor expression G of the embodiment of the present invention is:

When the number of switch inductor units is M, the number of switch capacitor units is N, and the number of upper bridge arm half-bridge sub-modules is K, the gain factor expression G of the circuit of the present invention is:

This embodiment adopts a phase-shift control strategy to control each IGBT in the converter. In mode 1, the IGBTs of the K sub-modules of the upper bridge arm are all triggered and turned on. In mode 2, the IGBTs of the K-1 sub-modules of the upper bridge arm are triggered and turned on. Therefore, the upper bridge arm switch tubes are triggered and turned on in sequence by shifting the phase by 2π/K. To ensure that there are always K-1 capacitors connected to the loop in the upper bridge arm, it is usually required that the denominator of the gain G is greater than 0, and the duty cycles d and D are greater than 0 and less than 1, so as to ensure that the switching state of the lower bridge arm sub-module is complementary to the switching state of each sub-module of the upper bridge arm, so as to ensure that K capacitors are connected in series to the loop at any time.

It should be noted that the number of lower bridge arm sub-modules in the present invention can be multiple, but under the phase-shifted pulse width modulation control strategy used in the present invention, the number of lower bridge arm sub-modules has no effect on the converter boost ratio, and when the number of upper bridge arm sub-modules is K, the lower bridge arm can use one sub-module to achieve stable operation.

As shown in Figures 4(a), 4(b), 4(c), 4(d) and 4(e), Matlab/Simulink simulation results of the output voltages Vi (i=1,2,3,4) of each sub-module of the upper and lower bridge arms of the converter, the voltages V _C5 and V _{C6 of} the _fifth capacitor C 5 and the sixth capacitor C 6 in the quasi-switch boost network, the high-voltage side output voltage V _o , the currents _i L1 and i L2 of the first inductor L ₁ and the first inductor L 2, and the load current i o of the circuit of the present invention are respectively given when M=1, N=1, K= ₃ , Vi=25V, D ₌₁₁ / ₁₅ , d= _0.2 , L ₁ =L ₂ =1.5mH, C=50μF, LS =1.76μH, C _{H =2200μF} , R= _1500Ω and the switching frequency _{f s} =10kHz. Among them, Figure 4(a) is the output voltage _Vi (i=1,2,3,4) of each submodule, _V4 works complementary to _V1 , _V2 , and _V3 , Figure 4(b) is the voltage V _C5 and V _C6 of the fifth capacitor _C5 and the sixth capacitor _C6 , Figure 4(c) is the waveform of the output voltage _Vo on the high-voltage side of the converter, Figure 4(d) is the waveform of the current i _L1 and i _L2 of the first inductor _L1 and the second inductor _L2 , and Figure 4(e) is the waveform of the current i _o flowing through the load.

In summary, the circuit of the present invention has a higher voltage step-up ratio, continuous input current on the low-voltage side, continuous load current, and can operate stably in an open-loop condition.

It is understood by those skilled in the art that, unless otherwise defined, all terms (including technical terms and scientific terms) used herein have the same meaning as the general understanding of ordinary technicians in the field to which the present invention belongs. It should also be understood that terms such as those defined in general dictionaries should be understood to have the same meaning as the meaning in the context of the prior art, and the technical terms and scientific terms in the present invention should not be interpreted in an idealized or overly formal sense.

The above specific implementation methods further describe in detail the purpose, technical solutions and beneficial effects of the present invention. It should be understood that the above specific implementation methods are only a specific implementation method of the present invention rather than a limitation of the present invention. For example, those skilled in the art can also select the number of switching inductor units, the number of switching capacitor units and the number of upper bridge units connected in series with the upper bridge arms according to the high gain required in the actual application scenario. The converter topology of the present invention can be constructed with switching inductors and switching capacitors of other circuit structures. Any modifications, equivalent substitutions and improvements made within the spirit and principles of the present invention should be included in the protection scope of the present invention.

Claims (5)

1. A single-stage high-gain modular multi-level resonant DC boost converter, characterized in that it comprises: M switching inductor units, a quasi-switch boost network, N switching capacitor units, an upper bridge arm formed by connecting K upper bridge units in series, a lower bridge arm including only one lower bridge unit, an output inductor, and an output filter capacitor; wherein, The circuit structures of the M switching inductor units are the same. The Mth switching inductor unit includes: a 3M-2th diode, a 3M-1th diode, a 3Mth diode and an M+1th inductor. The cathode of the 3M-2th diode, the cathode of the 3Mth diode and one end of the M+1th inductor are connected to serve as the fourth port of the Mth switching inductor unit. The anode of the 3M-1th diode is connected to the anode of the 3Mth diode to serve as the third port of the Mth switching inductor unit. The cathode of the 3M-1th diode is connected to the other end of the M+1th inductor to serve as the second port of the Mth switching inductor unit. The 3M-2th diode The anode of the tube serves as the first port of the Mth switching inductor unit, one end of the first inductor in the low-voltage side DC power supply loop is connected to the first port of the first switching inductor unit and the positive terminal of the low-voltage side voltage source, the other end of the first inductor is connected to the third port of the first switching inductor unit, the first port of the i+1th switching inductor unit is connected to the fourth port of the i-th switching inductor unit, the third port of the i+1th switching inductor unit is connected to the second port of the i-th switching inductor unit, and the second port of the Mth switching inductor unit is connected to the drain of the first IGBT in the quasi-switch boost network, 1≤i≤M, M is an integer greater than 0; The quasi-switch boost network includes: a first IGBT, a fourth diode, a fifth diode and a fifth capacitor, wherein the drain of the first IGBT and the anode of the fifth diode are connected to the second port of the Nth switch inductor unit, the source of the first IGBT is connected to the anode of the fourth diode and the negative plate of the fifth capacitor, the cathode of the fifth diode is connected to the positive plate of the fifth capacitor, and the cathode of the fourth diode is connected to the negative terminal of the low-voltage side voltage source; The circuit structures of the N switching capacitor units are the same, each switching capacitor unit includes a capacitor and a diode, and the positive plate of the capacitor in each switching capacitor unit is connected to the cathode of the diode, the negative plate of the capacitor in the qth switching capacitor unit is connected to the drain of the first IGBT in the quasi-switch boost network, the anode of the diode in the qth switching capacitor unit is connected to the positive plate of the fifth capacitor, the connection point between the positive plate of the capacitor in the qth switching capacitor unit and the cathode of the diode is connected to the anode of the diode in the q+1th switching capacitor unit and the negative plate of the capacitor in the q+2th switching capacitor unit, and the q+ The negative plate of the capacitor in the first switching capacitor unit is connected to the positive plate of the fifth capacitor, the connection point between the positive plate of the capacitor in the q+1th switching capacitor unit and the cathode of the diode is connected to the anode of the diode in the q+2th switching capacitor unit and the negative plate of the capacitor in the q+3th switching capacitor unit, the connection point between the positive plate of the capacitor in the q+2th switching capacitor unit and the cathode of the diode is the anode of the diode in the q+3th switching capacitor unit, and the negative plate of the capacitor in the q+4th switching capacitor unit, and the connection point between the positive plate of the capacitor in the Nth switching capacitor unit and the cathode of the diode is connected to the connection point of the upper and lower bridge arms; The circuit structures of the K upper bridge units are the same, and the Kth upper bridge unit includes: a Kth upper bridge IGBT, a Kth upper bridge diode and a Kth upper bridge capacitor. A diode is connected in anti-parallel in the Kth upper bridge IGBT, the source of the Kth upper bridge IGBT is connected to the cathode of the Kth upper bridge diode, the drain of the Kth upper bridge IGBT is connected to the positive plate of the Kth upper bridge capacitor, the negative plate of the Kth upper bridge capacitor is connected to the anode of the Kth upper bridge diode, and the source of the Kth upper bridge IGBT is connected to the cathode of the Kth upper bridge diode. As the input end of the Kth upper bridge unit, the negative plate of the Kth upper bridge capacitor serves as the output end of the Kth upper bridge unit, the input end of the first upper bridge unit is connected to one end of the output inductor, the output end of the pth upper bridge unit is connected to the input end of the p+1th upper bridge unit, the output end of the p+1th upper bridge unit is connected to the input end of the p+2th upper bridge unit, the pth upper bridge unit to the Kth upper bridge unit are connected in series in sequence, the output end of the Kth upper bridge unit is connected to the input end of the lower bridge unit, 1≤p≤K, K is an integer greater than 0; The lower bridge unit includes: a first lower bridge IGBT, a second lower bridge IGBT and a lower bridge capacitor. The first lower bridge IGBT and the second lower bridge IGBT are both anti-parallel connected with a diode. The drain of the first lower bridge IGBT is connected to the positive plate of the lower bridge capacitor, the source of the first lower bridge IGBT is connected to the drain of the second lower bridge IGBT, the source of the second lower bridge IGBT is connected to the negative plate of the lower bridge capacitor, the source of the first lower bridge IGBT serves as the input end of the lower bridge unit, the source of the second lower bridge IGBT serves as the output end of the lower bridge unit, and the output filter capacitor is connected in parallel between the other end of the output inductor and the output end of the lower bridge unit. 2. According to claim 1, a single-stage high-gain modular multi-level resonant DC boost converter is characterized in that M, N, and K are expressed according to the converter output gain expression: Select, where G is the output gain of the converter, d is the on-duty cycle of the first IGBT and the second lower bridge IGBT in one working cycle, d=1-K(1-D), D is the duty cycle of the upper bridge IGBT in the upper bridge unit, and int(*) is the rounding function. 3. A driving method for a single-stage high-gain modular multi-level resonant DC boost converter according to claim 2, characterized in that a switching cycle _Ts is divided into K working cycles _Te , _Te = _Ts /K, in each working cycle, firstly, the first IGBT, the second lower bridge IGBT, and the upper bridge IGBTs in all upper bridge units are turned on to make the converter work in an inductive energy storage mode, and the duration of the inductive energy storage mode accounts for d in one working cycle, and then the first IGBT, the second lower bridge IGBT, and the upper bridge IGBT of an upper unit are turned off to make the converter work in an inductive energy release mode, and the inductive energy release mode controls the upper bridge IGBTs in all upper bridge units through a carrier phase-shift pulse width modulation strategy so that in the inductive energy release mode of each working cycle, there are always K-1 upper bridge capacitors put into the upper bridge arm. 4. According to the driving method of a single-stage high-gain modular multi-level resonant DC boost converter described in claim 3, it is characterized in that the method of controlling the upper bridge IGBTs in all upper bridge units through the carrier phase-shift pulse width modulation strategy is: the trigger turn-on signals of all upper bridge IGBTs in all upper bridge units are sequentially phase-shifted by 2π/K to turn on K-1 upper bridge IGBTs. 5. According to claim 4, a driving method for a single-stage high-gain modular multi-level resonant DC boost converter is characterized in that the specific method of sequentially performing a 2π/K phase shift on the triggering and conducting signals of all upper bridge IGBTs in all upper bridge units is as follows: determining the serial number P of the turned-off upper bridge IGBT according to the expression P=mod(j+K, K), taking the phase of the triggering and conducting signal of the upper bridge IGBT in the first upper bridge unit as a reference, and determining the phase shift amount of the triggering and conducting signal of the upper bridge IGBT in the Pth upper bridge unit according to the serial number P of the turned-off upper bridge IGBT, wherein the phase shift amount of the triggering and conducting signal of the upper bridge IGBT in the Pth upper bridge unit is (P-1)*2π/K, and mod is a modulo function.