CN108063605B

CN108063605B - Radio frequency filter and method for tuning radio frequency filter

Info

Publication number: CN108063605B
Application number: CN201710952804.5A
Authority: CN
Inventors: 格尼希·楚祖基; 巴拉姆·A·威廉森
Original assignee: Resonant Inc
Current assignee: Murata Manufacturing Co Ltd
Priority date: 2014-03-14
Filing date: 2014-12-31
Publication date: 2022-01-18
Anticipated expiration: 2034-12-31
Also published as: CN108063605A; JP6158780B2; KR101691264B1; JP2015177542A; KR20150107585A

Abstract

The invention discloses a radio frequency filter and a method for tuning the radio frequency filter. A Radio Frequency (RF) filter includes: the apparatus includes a signal transmission path having an input and an output, a plurality of resonant elements arranged along the signal transmission path between the input and the output, and a plurality of non-resonant elements coupling the resonant elements together to form a stop band having a plurality of transmission zeroes corresponding to respective frequencies of the resonant elements and at least one sub-band between the transmission zeroes. The non-resonant elements include at least one variable non-resonant element for selectively introducing at least one reflection zero within the stop band to create a pass band within a selected one of the sub-bands. Wherein the stop band is within the microwave frequency range.

Description

Radio frequency filter and method for tuning radio frequency filter

The application has application date of 2014, 12 and 31, application number of 201410854568X, and inventive name of: patent application for low loss tunable rf filters.

Cross Reference to Related Applications

This application is a partial continuation of U.S. patent application serial No. 13/282,289 filed on 26/10/2011, U.S. patent application serial No. 13/282,289 is a continuation of U.S. patent application serial No. 12/959,237 filed on 2/2010 on 12/2010 on U.S. patent number 8,063,714, U.S. patent application serial No. 12/959,237 is a continuation of U.S. patent application serial No. 12/620,455 filed on 17/11/2009 on U.S. patent number 7,863,999, U.S. patent application serial No. 12/620,455 is a continuation of U.S. patent application serial No. 12/163,814 filed on 27/2008 on 27/6/7,639,101 on 2008 on U.S. patent application serial No. 12/163,814 claims priority to U.S. provisional patent application serial No. 60/937,462 filed on 27/2007 on 6/17/2006 on 11/7,719,382 on 2006 on U.S. patent application serial No. 7,719,382 The continuation-in-part application of/561,333, which is incorporated herein by reference in its entirety.

Technical Field

The present invention relates generally to microwave circuits, and in particular to microwave band pass filters.

Background

Electrical filters have long been used for the processing of electrical signals. In particular, such electrical filters are used to select a desired electrical signal frequency from an input signal by passing the desired signal frequency while blocking or attenuating other undesired electrical signal frequencies. Filters can generally be classified into categories including low pass filters, high pass filters, band pass filters, and band stop filters, which indicate the type of frequency selectively passed by the filter. Further, the filters may be classified by type as butterworth, chebyshev, inverse chebyshev, and elliptic filters, which indicate the type of bandshaped frequency response (frequency cut-off characteristic) the filter provides with respect to an ideal frequency response.

The type of filter typically used depends on the intended use. In communication applications, band pass filters are commonly used in cellular base stations and other telecommunication equipment to filter out or block RF signals in all but one or more predefined frequency bands. Such filters are commonly used, for example, in the front end of receivers to filter out noise and other unwanted signals that could damage the receiver components in a base station or telecommunications equipment. Placing a well-defined band-pass filter directly at the receiver antenna input will typically eliminate a number of adverse effects caused by strong interfering signals at frequencies near the desired signal frequency. Because of the placement of the filter at the receiver antenna input, the insertion loss must be very low so as not to degrade the noise figure. In most filter technologies, achieving low insertion loss requires a corresponding compromise in filter steepness or selectivity.

In commercial telecommunications applications, it is often desirable to use a narrow band filter to screen out as small a passband as possible, so that a fixed spectrum can be split into as large a number of frequency bands as possible, thereby increasing the number of actual users that can fit in the fixed spectrum. With the steep rise in wireless communications, such filtering should provide a high degree of selectivity (the ability to distinguish signals separated by small frequency differences) and sensitivity (the ability to receive weak signals) in an increasingly unfavorable frequency spectrum. Of most particular importance are the 800MHz-900MHz range for modular cellular communications and the 1,800MHz-2,200MHz range for Personal Communication Services (PCS).

Of most interest to the present invention is the need for a high quality factor Q (i.e., measuring the ability to store energy, and thus inversely related to its power consumption or loss), low insertion loss, tunable filter in military (e.g., RADAR), communications, and electronic intelligence (ELINT), and commercial fields such as in communications applications, including cellular, in a wide range of microwave and RF applications. In many applications, the receiver filter must be tunable to select a desired frequency or to capture an interfering signal frequency. Thus, the introduction of a linear, tunable bandpass filter between the receiver antenna and the first non-linear element in the receiver (typically a low noise amplifier or mixer) provides the significant advantages of a wide range of RF microwave systems, provided that the insertion loss is very low.

For example, in commercial applications, the frequency range of 1,800MHz-2,200MHz used by PCS may be split into several narrower frequency bands (a-F bands), only a subset of which may be used by telecommunications operators in any given area. It would therefore be beneficial for base stations and handheld units to be able to be reconfigured to operate with any selected subset of these frequency bands. As another example, in a RADAR system, high amplitude interfering signals, whether from "friendly" nearby sources or from interferers, may desensitize the receiver or intermodulate with high amplitude clutter signal levels to give false target indications. Thus, in high-density signal environments, RADAR alarm systems often become completely unavailable, in which case frequency hopping would be useful.

Microwave filters are generally constructed using two circuit building blocks: very efficient storage of a frequency f₀A plurality of resonators of energy; and coupling the electromagnetic energy between the resonators to form a multi-step or multi-pole coupler. For example, a four-pole filter may include four resonators. The strength of a given coupler is determined by its reactance (i.e., inductance and/or capacitance). The relative strength of the coupler determines the shape of the filter, an The topology of the coupler determines whether the filter performs a band-pass function or a band-stop function. Resonant frequency f₀Is substantially determined by the inductance and capacitance of each resonator. For conventional filter designs, the frequency at which the filter is effective (active) is determined by the resonant frequencies of the resonators that make up the filter. Each resonator must have a very low internal resistance so that the response of the filter is sharp and highly selective for the reasons discussed above. The demand for low resistance has pushed more for smaller size and lower cost resonators for a given technology.

Typically, fixed frequency filters are designed to minimize the number of resonators required to achieve a certain shape, since the size and cost of a conventional filter will grow linearly with the number of resonators required to achieve it. As in the case of semiconductor devices, lithographically defined filter structures, such as those in High Temperature Superconductor (HTS), micro-electro-mechanical systems (MEMS), and thin Film Bulk Acoustic Resonator (FBAR) filters, are much less sensitive to such size and cost than conventional combination or dielectric filter scaling.

The method now used to design the tunable filter follows the same method described above with respect to the fixed frequency filter. They therefore result in very efficient, effective and simple circuits, i.e. they result in the simplest circuits necessary to achieve a given filter response. In existing tuning techniques, all resonant frequencies of the filter are adjusted to tune the frequency of the filter. For example, if it is desired to increase the operating band of the device by 50MHz, then the resonant frequency of all the narrow band filters must be increased by 50 MHz. Although this prior art has been generally successful in tuning the frequency band, it inevitably introduces resistance into the resonators, thereby disadvantageously increasing the insertion loss of the filter.

Although an HTS filter can be tuned to change its resonant frequency by mechanically moving the HTS plates on each resonator in the filter without introducing significant resistance to the resonators, this technique is inherently slow (on the order of seconds) and requires a relatively large three-dimensional tuning structure. Insertion loss can be reduced in so-called switched filter designs; however, these designs still introduce a significant amount of loss between switching times and require additional resonators. For example, the insertion loss of the filter system can be reduced by providing two filters and a pair of single pole double throw switches (SP2T) that select between the filters, effectively reducing the tuning range requirements, but increasing the number of resonators by a factor of two and introducing losses from the switches. The losses of the filter system can be further reduced by introducing more switches and filters, but each additional filter will require the same number of resonators as the original filter and will introduce more losses by the required switches.

Therefore, there is still a need to provide a bandpass filter that can be tuned quickly with reduced insertion loss.

Disclosure of Invention

According to a first aspect of the present invention there is provided a radio frequency, RF, filter comprising: a signal transmission path having an input and an output; a plurality of resonant elements arranged along a signal transmission path between the input and the output; and a plurality of non-resonant elements coupling the resonant elements together to form a stop band having a plurality of transmission zeroes corresponding to respective frequencies of the resonant elements and at least one sub-band between the transmission zeroes, wherein the non-resonant elements comprise at least one variable non-resonant element for selectively introducing at least one reflection zero within the stop band to create a pass band in one of the at least one sub-band, wherein the stop band is in a microwave frequency range.

According to a further aspect of the present invention there is provided a method of tuning an RF filter as described above, comprising removing a portion of each tuning element, thereby modifying the frequency of the respective resonant element.

According to yet another aspect of the present invention, there is provided a method of tuning the RF filter described above, comprising: measuring a response of the RF filter to generate a set of measurement data; analyzing the set of measurement data to extract one or more filter design parameters for optimization; optimizing the one or more filter design parameters to achieve a desired filter response; generating a tuning scheme based on the optimized one or more filter design parameters; and tuning the RF filter by altering one of at least one of the non-resonant elements to selectively move a corresponding reflection zero along the stop band to move the pass band within a selected one of the sub-bands.

According to yet another aspect of the present invention, a Radio Frequency (RF) filter is provided. The RF filter includes a signal transmission path having an input and an output, a plurality of resonant elements disposed along the signal transmission path between the input and the output, and a plurality of non-resonant elements coupling the resonant elements together. The resonant elements are coupled together to form a stop band having a plurality of transmission zeroes corresponding to respective frequencies of the resonant elements and at least one sub-band between the transmission zeroes. The non-resonant elements have susceptance values located at least one reflection zero within the stopband to create a passband within a subband of the at least one subband.

The non-resonant elements include at least one variable non-resonant element for selectively introducing at least one reflection zero within the stop band to create a pass band in one of the sub-bands. In one embodiment, a plurality of sub-bands are provided, in which case the variable non-resonant elements can be used to displace the reflection zeroes along the stop band to create a pass band within a selected one of the sub-bands. The pass bands may have significantly different bandwidths within the selected sub-bands. In another embodiment, the variable non-resonant elements are used to shift at least one other reflection zero within the stop band to create another pass band within another one of the sub-bands.

For example, the variable non-resonant element may have, for example, an adjustable susceptance, and may include one or more of a variable capacitor, a low-loss switch (loss-loss switch), a varactor, and a switched capacitor. In one embodiment, each resonant element comprises a thin-film lumped element structure (such as, for example, a High Temperature Superconductor (HTS)), although the resonant element may take the form of any structure that resonates at a desired frequency.

The RF filter further includes an electrical controller configured to receive an operating temperature and adjust the variable non-resonant element based on the received operating temperature to selectively move the reflection zero along the stop band to move the pass band within a selected one of the sub-bands. In one embodiment, the electrical configuration controller is configured to adjust the variable non-resonant elements to selectively introduce reflection zeroes within the stop band to create a pass band within one of the sub-bands. For example, each non-resonant element may have a plurality of capacitors coupled in parallel with each other to form a capacitive circuit and at least one switch coupled to at least one capacitor. The electrical controller can then be configured to vary the reactance of each of the non-resonant elements by varying the capacitance of the capacitive circuit by operating the switch to selectively include or exclude at least one capacitor of the capacitive circuit, thereby selectively moving the reflection zeroes within the stop band to move the pass band within the selected sub-band.

The electrical controller can be configured to adjust the variable non-resonant element to selectively move the reflection zero along the stop band to return the pass band to a nominal as-designed location within the frequency range. In this case, the electrical controller can be configured to adjust the at least one resonant element based on the received operating temperature to selectively move the transmission zeroes corresponding to each frequency of the resonant element along the stop band to return the pass band to a nominal design position within the frequency range.

In one embodiment, the RF filter further comprises a temperature sensor configured to measure an operating temperature, in which case the electrical controller is configured to receive the operating temperature measured by the temperature sensor. The RF filter may further include a memory storing a look-up table including a plurality of reference operating temperatures and a plurality of sets of adjustment settings, each corresponding to a different operating temperature. In this case, the electrical controller is configured to compare the measured operating temperature to a plurality of reference operating temperatures in a look-up table, select a set of adjustment settings corresponding to the reference operating temperature that is closest to the measured operating temperature, and adjust the variable non-resonant element according to the set of adjustment settings.

Other and further aspects and features of the present invention will become apparent from a reading of the following detailed description of the preferred embodiments, which are intended to illustrate, but not to limit the invention.

Drawings

The drawings illustrate the design and utility of preferred embodiments of the present invention, in which like reference numerals refer to like elements. In order to better appreciate how the above-recited and other advantages and objects of the present invention are obtained, a more particular description of the invention briefly described above will be rendered by reference to specific embodiments thereof which are illustrated in the appended drawings. Understanding that these drawings depict only typical embodiments of the invention and are not therefore to be considered to be limiting of its scope, the invention will be described and explained with additional specificity and detail through the use of the accompanying drawings, in which:

fig. 1 is a block diagram of a tunable Radio Frequency (RF) filter constructed in accordance with an embodiment of the invention.

Fig. 2 is a graph of modeled frequency response for an exemplary wide stop band using 8 resonant elements.

Fig. 3 is a graph of the frequency response of fig. 2, wherein the pass band has been introduced within the sub-bands of the stop band.

Fig. 4 a-4 g are graphs of the frequency response of fig. 2, in which pass bands have been introduced within selected sub-bands of the stop band.

FIGS. 5 a-5 d are graphs of the frequency response of FIG. 2 in which the stop band has been frequency shifted and the pass band has been introduced at various locations of the sub-bands of the shifted stop band

Fig. 6 is a graph illustrating the simultaneous shifting of the transmission zeroes of the frequency response of fig. 2 to extend the range of pass bands introduced within selected sub-bands of the stop bands of fig. 4 a-4 g.

Figures 7 a-7 f are graphs of modeled frequency responses of an exemplary wide stop band using 9 resonant elements, where a pass band has been introduced within selected sub-bands of the stop band to cover the frequency range of a Personal Communication Service (PCS);

FIG. 8 is a graph illustrating the independent shifting of the transmission zeroes of the frequency responses of FIGS. 7 a-7 f to accommodate the passband introduced within selected sub-bands of the stopband;

FIGS. 9 a-9 f are graphs of the modeled frequency response of FIG. 2 in which a plurality of pass bands have been introduced within selected sub-bands of the stop band;

figure 10 is a block diagram of a tunable RF filter constructed in accordance with another embodiment of the invention.

FIG. 11 is a graph of a modeled frequency response of the filter of FIG. 10, wherein pass bands have been introduced at various locations of sub-bands of the shifted stop band;

FIG. 12 is a graph showing the variation of the frequency shift of the coupling values of the non-resonant elements relative to the pass band of FIG. 11 for use in the tunable RF filter of FIG. 10;

13 a-13 d show circuit representations of the tunable RF filter of FIG. 1;

FIG. 14 is a table showing component values used to model the RF filter of FIG. 14 in three filter states;

15 a-15 c are circuit implementation diagrams of the tunable RF filter of FIG. 1, particularly illustrating various filter states and corresponding frequency responses;

16 a-16 c are graphs of the frequency response of the RF filter of FIG. 14 in three states;

fig. 17 is a graph showing tuning of the RF filter of fig. 14 versus the insertion loss of the filter;

FIG. 18 is a graph comparing the insertion loss of the RF filter of FIG. 14 when tuned to the same frequency range versus the insertion loss of a conventional filter;

FIG. 19 is a graph comparing the insertion loss of the filter of FIG. 1 versus the insertion loss of a switched mode filter when tuned to the same frequency range;

FIG. 20 is a graph comparing the frequency response of two-, four-, and six-resonator tunable filters constructed in accordance with the present invention with the frequency response of a standard bandpass filter;

FIG. 21 shows another circuit representation of the tunable RF filter of FIG. 1;

FIG. 22 shows a coupling matrix of the circuit representation of FIG. 21;

fig. 23 a-23 c are graphs of the frequency response of the RF filter of fig. 21 and corresponding coupling matrices;

fig. 24 is a graph graphically illustrating coupling values in the coupling matrices of fig. 23 a-23 c for tuning the RF filter of fig. 21;

FIG. 25 is a graph graphically illustrating another set of coupling values that may be used to tune the RF filter of FIG. 21;

FIG. 26 is a graph graphically illustrating yet another set of coupling values that may be used to tune the RF filter of FIG. 21;

FIG. 27 is a plan view layout of one resonator in the tunable RF filter of FIG. 1, particularly illustrating a tuning fork for tuning the resonator;

FIG. 28 is a plan view layout of one resonator in the tunable RF filter of FIG. 1, particularly illustrating a trim tab for tuning the resonator; and

figure 29 is a block diagram of a tunable RF filter constructed in accordance with an embodiment of the invention.

Detailed Description

Referring to fig. 1, a tunable Radio Frequency (RF) filter 10 constructed in accordance with the present invention will now be described. In the illustrated embodiment, the RF filter 10 is a bandpass filter having a passband that is tunable over a desired frequency range, such as 800MHz-900MHz or 1,800MHz-2,220 MHz. In a typical scenario, the RF filter 10 is placed in the front end of a receiver (not shown) after a wide passband filter that rejects energy outside the desired frequency range. The RF filter 10 generally includes a signal transmission path 12 having an input 14 and an output 16, a plurality of nodes 17 disposed along the signal transmission path 12, a plurality of resonant branches 19 each extending from a node 17, and a plurality of non-resonant branches 21 each extending from a node 17. The RF filter 10 further comprises a plurality of (in this case, four) resonant elements 18 coupled between the input 14 and the output 16 and in particular between the resonant branch 19 and ground, a plurality of tuning elements 20 for adjusting the frequency of the resonant elements 18, a plurality of non-resonant elements 22 coupling the resonant elements 18 together, four of the non-resonant elements 22 being coupled between the non-resonant branch 21 and ground. The RF filter 10 further includes an electrical controller 24 configured to tune the RF filter 10 to a selected narrow band within the frequency range.

The signal transmission path 12 may include physical transmission lines to which the non-resonant elements 22 are directly or indirectly coupled, but in alternative embodiments, no physical transmission lines are used. In the illustrated embodiment, the resonant elements 18 comprise lumped element electrical components, such as inductors and capacitors, and in particular, thin film lumped structures, such as planar spiral structures, saw tooth serpentine structures, single coil structures, and dual coil structures. Such structures may include thin film epitaxial High Temperature Superconductors (HTS) patterned to form capacitors and inductors on low loss substrates. U.S. patent No. 5,616,539, which is incorporated herein by reference in its entirety, sets forth further details discussing high temperature superconductor lumped element filters.

In the illustrated embodiment, the resonant element 18 is formed by a susceptance B^RShowing, and the non-resonant element 22 is formed by susceptance B^NIt is shown that the non-resonant elements 22 are coupled in parallel with the resonant elements 18 and that the admittance transformers J are coupled between the resonant elements 18. Selected ones of the non-resonant elements 22 can be changed while any remaining ones of the non-resonant elements 22 remain fixed.

As will be described in greater detail below, the non-resonant elements 22 can be varied to tune the passband substantially throughout the frequency range, with only minor adjustments, if necessary, to accommodate and/or shift the passband in opposite portions of the frequency range. In this manner, the insertion loss of the filter 10 is significantly reduced since the non-resonant elements 22 are used as the primary means of tuning the filter 10 rather than the resonant elements 18. That is, because adjusting non-resonant elements 22 contributes less to the loss of filter 10 than adjusting the significantly loss sensitive resonant elements 18, filter 10 will have less loss than prior art filters that employed resonant elements as the primary means for tuning filter 10. Furthermore, the tuning speed of the filter 10 is faster, if any, due to the very slight adjustment of the frequency of the resonator elements 18.

The RF filter 10 achieves this by introducing a narrow pass band with a selected area of a wide stop band. That is, although the RF filter 10 is employed as a bandpass filter, the resonant elements 18 are actually coupled together by the non-resonant elements 22 (rather than creating a passband), but rather creating a wide-bandstop response with transmission zeroes (in this case, four) corresponding to the respective frequencies of the resonant elements 18. The electrical controller 24 then adjusts the non-resonant elements 22 to introduce and displace the reflection zeroes along the stop band to move the narrow pass band within the desired frequency range. The electrical controller 24 may also adjust the frequency of the resonant elements 18 through the tuning elements 20 to move the transmission zeroes along the frequency range to optimize the filter response. In the illustrated embodiment, an electrical controller 24 including a memory (not shown) for storing the values of the non-resonant elements 22 is necessary to achieve the desired location of the pass band over the frequency range.

This technique will now be described with reference to various exemplary filter responses modeled according to the following equations:

wherein S is₁₁Is the input reflection coefficient, S, of the filter₂₁Is the forward transmission coefficient, s is the normalized frequency, F and P are polynomials of order N of the generalized complex frequency s (where N is the number of resonant elements), and epsilon is a constant defined to equal the return loss. Since the molecule has an order N, the coefficient S₁₁And S₂₁Can have up to N zeros. When coefficient S₁₁、S₂₁All N zeros, the filter response is considered to be a complete ellipse. Further details of discussing filter modeling are set forth in "Microtrip Filters for RF/Microwave Application," Jia-Shen G. Hong and M.J.Lancaster, Wiley-Interscience 2001. The normalized frequency s-iw can be mapped to the actual frequency according to the following equation:

where f is the actual frequency, f_cIs the center frequency and BW is the bandwidth of the filter. Further details on the discussion of normalized frequency conversion to actual frequency are set forth in "Microwave Filters, Impedance-Matching Networks, and Coupling Structures," g.matthaei, l.young and e.m.t.jones, McGraw-Hill (1964).

Fig. 2 shows an exemplary wide band-stop filter response, which is modeled using 8 resonant elements, creating 8 (only 6 shown) corresponding transmission zeroes 30 (as best shown in the right side view of fig. 2) at each resonant element frequency to form a stop band 32, and 8 (only 6 shown) reflection zeroes 34 (as best shown in the left side view of fig. 2) that fall outside of the stop band 32. In this particular example, the transmission zeroes 30 are located at-1.05, -0.75, -0.45, -0.15, 0.45, 0.75, and 1.05 in the normalized frequency range, thereby creating a stop band having a normalized frequency range between-1.05 and 1.05. As shown in the right side view of fig. 2, the filter response includes 7 "beats" at-0.90, -0.60, -0.30, 0.0, 0.30, 0.60, and 0.90, respectively, in the region 36 between the transmission zeroes 30. Thus, in general, a band-stop filter includes N transmission zeroes (corresponding to N resonant elements), up to N reflection zeroes, and N-1 bounce regions 36.

Notably, the pass band can be formed by any one of the jumps in the region 36 (hereinafter referred to as a "sub-band") shown in FIG. 2 by displacing at least one of the reflection zeroes 34 to the stop band 32 (i.e., by adjusting the values of the non-resonant elements). For example, FIG. 3 shows an exemplary filter response in which 4 reflection zeroes 34 have been introduced into the stop band of FIG. 2 to create a pass band 38 within the center sub-band 36(4) (i.e., at 0). The reflection zeroes 34 can be displaced along the stop band 32 (i.e., by adjusting the values of the non-resonant elements) to create a pass band 38 within a selected one of the sub-bands 36. That is, the reflection zeroes 34 can be displaced along the stop band 32 to "hop" the pass band 38 between the sub-bands 36.

For example, fig. 4 a-4 g illustrate an exemplary filter response in which 4 reflection zeroes 34 have been displaced within the stop band 32 to selectively create a pass band 38 in the center of all 7 sub-bands 36. That is, with reference to fig. 4 a-4 g in sequence, the pass band 38 hops from the first sub-band 36(1) (fig. 4a) to the second sub-band 36(2) (fig. 4b), to the third sub-band 36(3) (fig. 4c), to the fourth sub-band 36(4) (fig. 4d), to the fifth sub-band 36(5) (fig. 4e), to the sixth sub-band 36(6) (fig. 4f), and finally to the seventh sub-band 36(7) (fig. 4 g). Thus, in the illustrated embodiment, the center of the pass band 38 can hop between-0.90, -0.60, -0.30, 0.0, 0.30, 0.60, and 0.90. It should be noted that while the sequence of fig. 4 a-4 g means that the pass band 38 hops between adjacent sub-bands 36, the pass band 38 may hop between non-adjacent sub-bands 36, for example, from the second sub-band 36(2) to the fifth sub-band 36 (5).

While the pass band 38 can hop between the sub-bands 36 to discretely cover the desired frequency range, the transmission zeroes 30 can be simultaneously moved in unison (i.e., by adjusting the frequencies of the resonant elements) from their nominal positions to displace the entire stop band 32, and thus the pass band 38, within the normalized frequency range. Thus, the pass band 38 can be moved from the center of the sub-bands 36 (i.e., -0.90, -0.60, -0.30, 0.0, 0.30, 0.60, and 0.90) to cover a continuum of the desired frequency range. Thus, if all of the transmission zeroes can be displaced by +/-0.15 from their nominal positions (i.e., the resonant elements tune together in a frequency range of +/-0.15), each of the pass bands 38 shown in FIGS. 4 a-4 g will cover 15% of the normalized frequency range from-1.05 to 1.05.

By way of example, if the center of the desired pass band 38 is at-0.20, then the pass band 38 may be positioned in the third sub-band 36(3) (at the center-0.30 of FIG. 4 c) and the transmission zeroes 30 may be displaced 0.10 from their nominal positions to move the pass band 38 from-0.30 to-0.20. If the center of the desired pass band 38 is at 0.85, the pass band 38 can be positioned in the seventh sub-band 36(7) (at the center 0.90 of FIG. 4 g) and the transmission zeroes 30 can be shifted-0.05 from their nominal positions to move the pass band 38 from 0.90 to 0.85.

Although the pass band 38 is shown in fig. 4 a-4 g with the pass band 36 positioned centered within the sub-bands 36, the reflection zeroes 34 can be displaced within the stop band 32 (i.e., by adjusting the values of the non-resonant elements) to selectively move the pass band 38 within selected sub-bands 36. In such a case, the pass band 38 can be hopped between the sub-bands 36 and moved within the sub-bands 36, thereby reducing the amount of transmission zeroes 30 needed to adjust the continuum of pass bands 38 to cover the desired frequency range. For example, fig. 5 a-5 d illustrate exemplary filter responses for the center sub-band 36(4) in which all of the transmission zeroes 30 are displaced 0.05 from their nominal positions (i.e., by increasing the frequency of the resonant elements 18 by 0.05), and the reflection zeroes are incrementally displaced 0.05 from their nominal positions (i.e., by adjusting the non-resonant elements 22).

Specifically, with sequential reference to fig. 5 a-5 d, the transmission zeroes are shifted 0.05 from their nominal positions, thereby shifting the pass band 38 from 0 (fig. 5a) to 0.05 (fig. 5 b). Then, after fixing the transmission zeroes 30, the reflection zeroes 34 are incrementally displaced by 0.05 from their nominal positions to move the pass band 38 from the center (0.05 in fig. 5 b)) of the sub-band 36(4) to a position 0.05 (0.10 in fig. 5 c) to the right of the center of the sub-band 36(4), and then to a position 0.10 (0.15 in fig. 5 d) to the right of the center of the sub-band 36 (4).

Although this approach may break the symmetry of the attenuation slope of the bandpass filter, in this case it reduces the required displacement of the transmission zeroes 30, and therefore the tuning range of the resonant elements is reduced from 15% to 5% to obtain the same tuning range as in the case where the reflection zeroes 34 are not displaced within the sub-bands 36. As a result, the loss of the filter is further reduced.

Note that while it is theoretically possible to displace the transmission zeroes 30 within the entirety of the sub-bands 36, in which case each pass band 38 can cover approximately 15% of the entire stop band 32 without tuning the resonant elements, in practice, the loss of the filter increases significantly due to the close proximity of the reflection zeroes 34 to the transmission zeroes 30. To this end, it is preferable to shift the transmission zeroes 30 along with the reflection zeroes 34 to allow the pass band 38 to move throughout the frequency range without significant loss.

For example, referring to FIG. 6, the transmission zeroes 30 are displaced within a range of) +/-0.05 relative to their nominal positions (shown by the horizontal dashed lines) to position the pass band 38 anywhere within the nominal frequency range of-1.05 to 1.05 (as represented by the oblique dashed lines). As the frequency of the pass band 38 moves from-1.05 to 1.05, the reflection zeroes 34 hop from one sub-band 36 to the next sub-band 36, the reflection zeroes 34 are displaced within a range of +/-0.10 along the sub-band 36, and the transmission zeroes 30 are displaced within a range of +/-0.05, with the total range between hops being 0.30.

Specifically, at the beginning of the tuning range, the transmission zeroes 30 are initially positioned at-0.05 (i.e., -1.05, -0.75, -0.45, -0.15, 0.45, 0.75, 1.05) relative to their nominal positions, which centers the first sub-band 36(1) at-0.95, in which case the reflection zeroes 34 are initially positioned at-0.10 relative to their nominal positions in the first sub-band 36(1) to place the pass band 38 at-1.05. While the transmission zeroes 30 are fixed, the reflection zeroes 34 in the first sub-band 36(1) can be displaced to their nominal positions to move the pass band 38 from-1.05 to-0.95. While the reflection zeroes 34 are fixed, the transmission zeroes 30 can then be displaced 0.05 relative to their nominal positions, which moves the center of the first sub-band 36(1) to-0.85, thereby moving the pass band from-0.95 to-0.85. While the transmission zeroes are again fixed, the reflection zeroes 34 can be displaced 0.10 relative to their nominal positions to displace the pass band from-0.85 to-0.75.

Once the pass band 38 reaches-0.75, the reflection zeroes 34 will hop from the first sub-band 36(1) to the second sub-band 36(2), and then the transmission zeroes 30 will again be displaced-0.05 relative to their nominal positions, which moves the center of the second sub-band 36(2) to-0.65, in which case the reflection zeroes 34 will be initially positioned-0.10 relative to their nominal positions to maintain the pass band 38 at-0.75. The transmission zeroes 30 and reflection zeroes 34 are then moved in coordination with one another to move the pass band 38 from-0.75 to-0.45 in the same manner as described above with respect to the first sub-band 36 (1). Once the pass band 38 reaches-0.45, the reflection zeroes 34 will hop from the second sub-band 36(2) to the third sub-band 36(3), and so on, until the pass band 38 reaches 1.05.

Although the RF filter 10 is described above as being capable of tuning a narrow passband within a continuum of a desired frequency range (i.e., the RF filter may be reconfigured in a continuous manner), the RF filter 10 may be reconfigured in a discrete manner such that the passband 38 may be discretely centered within a selected band region. For example, in a PCS application, the RF filter 10 may be reconfigured to operate in any of the six a-F bands by locating a narrow pass band at a selected one of these bands.

Fig. 7a to 7f show exemplary filter responses corresponding to six different reconfiguration states of the RF filter. In this case, the modeled filter has nine transmission zeroes 30 (only seven shown) to create a stop band 32 having eight sub-bands 36 located between the respective transmission zeroes 30, and seven reflection zeroes 34 that can be displaced into the stop band 32 to create a pass band 38 within selected ones of the six sub-bands 36. Thus, the RF filter may be reconfigured to operate in the a-band (fig. 7a), D-band (fig. 7B), B-band (fig. 7C), E-band (fig. 7D), F-band (fig. 7E), or C-band (fig. 7F) of the PCS communication protocol. As shown, the width of the pass band 38 within the sub-bands 36 varies, as determined by the spacing of adjacent transmission zeroes 30. Specifically, the width of A, B, and C bands, was approximately 2.5 times the width of D, E, and F bands.

Note that because in this reconfigurable embodiment, the pass band 38 need not be moved within a continuum of the desired frequency range, but the pass band 38 is designed to be wide enough to cover the desired frequency range, the transmission zeroes 30 are not shifted to extend the range of the pass band 38. Also, as shown in fig. 8, the transmission zeroes 30 are independently displaced from their nominal positions to make room for the pass band 38 or otherwise improve attenuation performance. For example, the second and third transmission zeroes 30(2), 30(3) are moved away from each other to make room for the reflection zero 34 on the a band; moving the fourth and fifth transmission zeroes 30(4), 30(5) away from each other to make room for reflection zeroes on the B band, and moving the seventh and eighth transmission zeroes 30(7), 30(8) away from each other to make room for reflection zeroes 34 on the C band; moving the third and fourth transmission zeroes 30(3), 30(4) away from each other to make room for the reflection zeroes 34 on the D band, and moving the fifth and sixth transmission zeroes 30(5), 30(6) away from each other to make room for the reflection zeroes 34 on the E band; and the sixth and seventh transmission zeroes 30(6), 30(7) are moved away from each other to make room for the reflection zeroes 34 on the F band.

Although the above-described techniques are described as introducing a single pass band 38 within the stop band 32 (i.e., one pass band at a time), multiple pass bands may be introduced within the stop band 32. For example, fig. 9 a-9 f illustrate exemplary filter responses in which two sets of four reflection zeroes 34 have been displaced within the stop band 32 to selectively create two pass bands 38(1), 38(2) at the center of a selected pair of sub-bands 36. That is, with reference to fig. 9a to 9f in sequence, the pass bands 38(1), 38(2) are introduced into the second and third sub-bands 36(2), 36(3) (fig. 9a), into the third and fifth sub-bands 36(3), 36(5) (fig. 9b), into the third and fourth sub-bands 36(3), 36(4) (fig. 9c), into the second and fourth sub-bands 36(2), 36(4) (fig. 9d), into the second and sixth sub-bands 36(2), 36(6) (fig. 9e), and into the second and fifth sub-bands 36(2), 36(5) (fig. 9 f).

Referring now to fig. 10 and 11, a basic tunable filter 50 will be described for the purpose of explaining the correlation between the variable non-resonant elements (in terms of coupling values) and the resulting shifted values of the narrow pass band within the wide stop band. As shown in fig. 10, RF filter 50 generally includes a signal transmission path 52 having an input 54 and an output 56, a plurality (in this case, two) of resonant elements 58 between input 54 and output 56, and a plurality of non-resonant elements 62 coupling resonant elements 58 together. Tuning elements may be used to adjust the frequency of the resonant element 58 and an electrical controller (not shown) may be used to tune the RF filter 50 to a selected narrow band within the frequency range. Like filter 10 shown in fig. 1, resonant element 58 of filter 50 is comprised of susceptance B ^RShowing, and the non-resonant element 62 is formed by susceptance B^NShown, coupled in parallel with the resonant elements 58, and an admittance transformer J coupled in parallel between the resonant elements 58. Selected non-resonances in the non-resonant elements 22Component (in this case, susceptance B)^N) Can be varied while the remaining non-resonant elements of the non-resonant elements 22 (in this case, the admittance transformer J) remain fixed.

The filter 50 is modeled to create an exemplary filter response as shown in fig. 11. The frequencies of the two resonant elements 58, and thus the two transmission zeroes 70, are set at 0.95GHz and 1.05GHz, creating a stop band (not shown) having a normalized frequency range between 0.95GHz and 1.05 GHz. In this case, because there are only two resonant elements 58, a single sub-band 76 is positioned at the center 1.00GHz between the transmission zeroes 70. Accordingly, reflection zeroes (not shown) are introduced and displaced along the stop band to move the pass band 78 (five positions of the pass band 78 are shown) within a single sub-band 76.

As further shown in FIGS. 11 and 12, a variable non-resonant element 66 (designated B in FIG. 12) may be adjusted^N(L) and B^N(S)) to move the passband 78 by changing their coupling value by about 1.00GHz of nominal frequency. Specifically, the frequency of the pass band 78 will follow the non-resonant element B on the load side ^N(L) increase in percentage coupling value and B of non-resonant element on power supply side^N(S) decreases with decreasing percentage coupling value (left shift), and with load-side non-resonant element B^N(L) reduction in percentage coupling value and non-resonant element B on power supply side^NThe percentage coupling value of (S) increases.

Referring to fig. 13 a-13 c, the non-resonant elements 22 of the filter 10 of fig. 1 may be replaced with actual components such that the filter 10 may be modeled and implemented. As shown in fig. 13a, the circuit is first simplified to only use the non-resonant elements 22 to reconfigure the constituent components necessary for the filter 10. In this case, the tuning element 20 is not necessary for the reconfiguration of the analog (modelling) filter 10, and is therefore removed from the circuit representation of fig. 13 a. As shown in fig. 13b, the block components represented by the circuit of fig. 13a have been replaced with actual circuit components. By replacing B with a capacitor^N Non-resonant elements 22 are shown, the non-resonant elements 22 shown by J being replaced by a capacitive pi-network and by a parallel capacitor-inductor combinationChange from B^R A resonator element 20 is shown. The circuit representation of fig. 13b is further simplified to that of fig. 13c, the non-resonant elements 22 of which may be altered to effect reconfiguration of the filter 10.

The filter 10 of fig. 13c was simulated using actual circuit component values. The circuit of fig. 13c is modeled in accordance with the polynomial equation discussed above, except that the component values relate to the coefficients of the polynomial. As discussed above, the filter 10 has four resonant elements 18 and, therefore, four transmission zeroes and three sub-bands formed therebetween in its frequency response. The values of the capacitor non-resonant elements 22 in the circuit representation of figure 13c can therefore be adjusted according to one of the three sets of values shown in figure 14 to hop the passband between the three sub-bands to place the filter 10 in a selected one of the three states. The circuit representation according to fig. 13d models each capacitor in the circuit representation of fig. 13 c. Specifically, each capacitor C is represented as having a variable capacitor C_dParallel fixed capacitor C₀And a variable capacitor C_dA series circuit of resistors R (representing switches).

Referring now to fig. 15a to 15c, a filter 10 using the basic architecture shown in fig. 13c can be reconfigured between one of three states by adjusting selected ones of the non-resonant elements 22. As shown, all of the frequency responses of filter 10 have four transmission zeroes 30 corresponding to the frequencies of the four resonant elements 18 and three subbands 36 formed between transmission zeroes 30. Thus, the pass band 38 can be created in each of the three sub-bands 36 to support a total of three different states: create a left state for the pass band 38 in the first sub-band 36 (1); creating an intermediate state for the pass band 38 in the second sub-band 36 (2); and a right state that creates the pass band 38 in the third sub-band 36 (3).

As shown, each non-resonant element 22 has three capacitors C connected in parallel₁-C₃With two capacitors C outside each having a switched capacitance₁And C₂And boost switch S₁And S₂Resistance loss resistor R of₁And R₂Are connected in series.Thus, by closing switch S₂And S₃Can be combined with a capacitor C₁And C₂Included in the circuit, and by independently opening the switch S₁And S₂Removing capacitor C from the circuit₁And C₂. Thus, assume capacitor C₁-C₃Having equal values, each non-resonant element 22 may have a selected one of three values: c₁(S₁、S₂None closed), C₂+C₃(switch S)₁、S₂One of them is closed), or C₁+C₂+C₃(switch S)₁、S₂Both closed). Switch S₁And S₂May be any suitable low loss switch, for example, a low loss GaAs switch. Alternatively, other variable elements capable of adjusting the capacitance value may be used, such as a variable capacitor, GaAs varactor diode, or switched capacitor.

It has been determined that when the non-resonant elements 22 have values determined by the switching states shown in fig. 15a, the pass band 38 can be placed in the first sub-band 36(1) (left state); when the non-resonant elements 22 have values determined by the switching states shown in fig. 15b, are placed in the second sub-band 36 (2) (intermediate state); and placed in the third sub-band 36(3) (intermediate state) when the non-resonant elements 22 have values determined by the switch states shown in fig. 15 c. The filter 10 may be tuned using the parameter extraction and analysis techniques disclosed in U.S. patent application serial No. 11/289,463, entitled Systems and Methods for Tuning Filters, which is expressly incorporated herein by reference. For purposes of illustration, the light bulb adjacent to the switch in the closed state has been shown lit (colored), and the light bulb adjacent to the switch in the open state has been shown extinguished (not colored). Although the filter 10 has been described with respect to fig. 15 a-15 c as having the ability only to hop the pass band 38 between sub-bands 36, the accuracy (resolution) of the circuit can be increased by adding more switched capacitors in order for the pass band 38 to be able to move within the selected sub-band 36. Also, because the pass band 38 is centered within the sub-bands 36, no tuning elements are shown coupled to the resonant elements 18.

Reference is now made to fig. 17, which illustrates tuning of the artificial filter 10 shown in fig. 13c along a frequency range of 770MHz to 890MHz to minimize insertion loss. In this scenario, the filter 10 is tuned by adjusting the non-resonant elements 22 to hop the pass band 38 between the centers of the sub-bands 36 (as shown in fig. 16 a-16 c) and changing the frequencies of the resonant elements 18 to move the pass band 38 within the sub-bands 36 (i.e., to cover the range of frequencies between the centers of the sub-bands 36). As shown, moving the pass band 38 from 890MHz at the center of the third sub-band 36(3) (shown in fig. 15 c) to 850MHz to the left of the third sub-band 36(3), the insertion loss of the filter 10 increases from about-0.2 dB to about-1.5 dB. Once 850MHz is reached, the pass band 38 hops from the third sub-band 36(3) to the center of the second sub-band 36(2) (shown in FIG. 15 b), thereby reducing the insertion loss from about-1.5 dB to about-0.25 dB. The insertion loss of the filter 10 then increases from about-0.25 dB to about-1.5 dB by moving the pass band 38 from the center 850MHz of the second sub-band 36(2) to the left 810MHz of the second sub-band 36 (2). Once 810MHz is reached, the pass band 38 hops from the second sub-band 36(2) to the center of the first sub-band 36(1) (shown in FIG. 15 a), reducing the insertion loss from about-1.5 dB to about-0.7 dB. The insertion loss of the filter 10 then increases from about-0.7 dB to about-1.9 dB by moving the pass band 38 from the center 810MHz of the first sub-band 36(1) to the left 770MHz of the first sub-band 36 (1). Thus, it can be appreciated that by moving the pass band along the frequency range, the filter 10 can cover the full range of the 770MHz to 890MHz frequency range while hopping between sub-bands 36 to minimize insertion loss.

Using the modeling parameters shown in fig. 15, it has been demonstrated that the insertion loss is significantly reduced over a range of frequencies when the filter is tuned using the non-resonant elements 22 as compared to using only the resonant elements 18. For example, as shown in fig. 18, when the frequencies of the non-resonant elements 22 and the resonant elements 18 are adjusted to tune the filter 10 in the frequency range 770MHz to 890MHz, the worst case insertion loss of the filter 10 is approximately 8dB lower than the insertion loss of the filter 10 when the frequencies of the resonant elements are only adjusted to tune the filter 10 in the same frequency range.

It has also been demonstrated that the filter 10 as modeled according to the parameters shown in fig. 15 has significantly lower insertion loss than the prior art switched filter tuning technique. For example, as shown in fig. 19, when the frequencies of the variable non-resonant elements and resonant elements are adjusted to tune the filter 10 in the frequency range 770MHz to 890MHz, the insertion loss of the filter 10 is significantly lower than that of a switched filter tuned in the same frequency range (assuming a small insertion loss from the addition of a switch and adjusting the frequencies of the resonant elements to cover half of the total tuning range between switching).

Note that while the insertion loss of a bandpass filter is conventionally considered to increase with an increase in the number of resonant elements, it has been demonstrated that the insertion loss does not increase with an increase in the number of resonant elements used in a filter employing the design techniques described herein. For example, as shown in fig. 20, 2-, 4-, and 6-resonator filter designs and standard ground filter designs using the techniques described herein are plotted for a frequency range of 750GHz to 950 GHz. As shown, the Q of the closest resonant element-rather than the number of resonant elements-results in the most insertion loss.

It should be noted that varying the value of the non-resonant element 22 coupled in series to the resonant element 18 may vary the transmission zero somewhat. In order to provide optimal performance of the filter, it is preferable that these transmission zeros not be inadvertently shifted.

Specifically, as shown in fig. 21, the circuit is again simplified to only the constituent components necessary to reconfigure the filter 10 using the non-resonant elements 22. In this case, the tuning element 20 is not necessary for the reconfiguration of the modelling filter 10 and is therefore removed from the circuit representation of fig. 21.

In the illustrated embodiment, there are four susceptances B ^R(specifically, B)₁ ^R、B₂ ^R、B₃ ^RAnd B₄ ^R) The illustrated resonant elements 18 and fifteen non-resonant elements 22, which may be arranged to be defined by susceptances B^N(specifically, B)_S ^N、B₁ ^N、B₂ ^N、B₃ ^N、B₄ ^NAnd B_L ^N) Six non-resonant elements 22(1) (also called NRN-ground (shunt non-resonant elements)) are shown, consisting of an admittance transformer J (in particular, J, ground)₀₁、J₁₂、J₂₃、J₃₄And J₄₅) Five non-resonant elements 22(2) are shown (also referred to as NRN-NRN (series non-resonant elements)), and four admittance transformers J (specifically J)₁、 J₂、J₃And J₄) The non-resonant elements 22(3) (also referred to as NRN-resonators (resonator coupling) are shown. The non-resonant elements 22(1), 22(2) are coupled in parallel to each resonant element 18, while the non-resonant elements 22(3) are coupled in series to each resonant element 18. The selected non-resonant elements 22 can be changed while any remaining non-resonant elements 22 remain fixed. In the illustrated embodiment, the non-resonant elements 22 (i.e., non-resonant elements 22(3)) coupled in series to resonant elements 18 (which tend to "raise" the resonant frequency when implemented in a practical solution) remain fixed.

It should be noted that in designs where the resonant element 18 is implemented using an acoustic resonator such as a Surface Acoustic Wave (SAW), a thin Film Bulk Acoustic Resonator (FBAR), a microelectromechanical system (MEMS) resonator, the non-resonant element 22 may be implemented as an electrical or mechanical coupling element. In this case, it may be advantageous to implement the non-resonant elements 22(3) to act as electromechanical transducers to allow the non-resonant elements 22(3) of the circuit and the acoustic wave resonant elements 18 to remain fixed, while also allowing the use of only the non-resonant elements 22(1), 22(2) for electronic tuning.

Fig. 22 shows a coupling matrix representation of the filter 10. As shown, nodes S, 1-4, L, and 5-8 (shown in FIG. 20) are on the left side of the matrix representation, and nodes S, NRN1-NRN4 (non-resonating nodes), L, and resonating nodes R1-R4 are on the upper side of the matrix representation. Also shown in fig. 22, the coupling values between the nodes are the susceptance values and admittance transformer values of the resonant element 18 and the non-resonant element 22.

Simulation using different sets of coupling coefficients to hop the pass band 38 between the centers of the sub-bands 36The filter representation shown in fig. 21. In particular, fig. 23 a-23 c illustrate exemplary filter responses (and their corresponding coupling matrix representations) in which four reflection zeroes 34 have been displaced within the stop band 32 to selectively create the pass band 38 at the center of all three sub-bands 36. That is, with reference to fig. 23 a-23 c in sequence, the pass band 38 hops from the first sub-band 36(1) (fig. 23a) to the second sub-band 36(2) (fig. 23b), and then to the third sub-band 36(3) (fig. 23 c). Thus, the center of the pass band 38 hops between nominal frequencies-0.80, 0.0, and 0.80. As can be appreciated from the corresponding matrix representations of fig. 23 a-23 c, the series-coupled non-resonant elements 22(3) (i.e., J) ₁-J₄) Is fixed at-1 while the admittance transformer values of the parallel-coupled non-resonant elements 22(1), 22(2) are changed to hop the pass band between the sub-bands 36. The variation (and lack of variation) of these values as the pass band 38 hops between the three nominal frequencies is graphically illustrated in fig. 24. As shown, the parallel-coupled non-resonant elements 22(1), 22(2) (i.e., J)₀₁、J₁₂、J₂₃、J₃₄、J₄₅、B₁ ^N、B₂ ^N、B₃ ^NAnd B₄ ^N) The value of (c) is changed, and the series-coupled non-resonant elements 23(3) (i.e., J)₁、J₂、J₃And J₄) The value of (c) is kept constant.

As discussed above with respect to fig. 4 a-4 g, while the pass band 38 can jump between the sub-bands 36 to discretely cover the desired frequency range, the transmission zeroes can be simultaneously moved from their nominal positions (i.e., by adjusting the frequencies of the resonant elements) to displace the entire stop band 32, and thus the pass band 38, within the normalized frequency range. Thus, with respect to fig. 23 a-23 c, the pass band 38 can be moved from the center of the sub-bands 36 (i.e., -0.80, 0.0, and 0.80) to cover a continuum of the desired frequency range. Thus, if all of the transmission zeroes 30 can be displaced by +/-0.40 from their nominal positions (i.e., the resonant elements are tuned together for a frequency range of +/-0.40), each pass band 38 shown in FIGS. 23 a-23 c will cover 33% of the normalized frequency range from-1.20 to 1.20.

While the pass band 38 is positioned at the center of the sub-bands 36 as shown in fig. 23 a-23 c, the reflection zeroes 34 can be displaced within the stop band 32 (i.e., by adjusting the values of the non-resonant elements) to selectively move the pass band 38 within a selected sub-segment 36. In this case, the pass band 38 can be hopped between sub-bands 36 and moved within each sub-band 36 to reduce the amount of transmission zeroes 30 required to adjust the pass band 38 to cover a continuum of the desired frequency range. For example, FIG. 25 graphically illustrates the variation (no variation) in the value of the non-resonant elements 22 as the pass band 38 moves within a continuum of the nominal frequency range of-1.0 to 1.0.

Note that the coupling values listed in fig. 25 are quite different from those listed in fig. 24, and therefore, it should be appreciated that there is more than one coupling matrix for each filter (i.e., the coupling matrices do not have unique solutions). For example, FIG. 26 graphically illustrates another set of variations (no variation) in the values of the non-resonant elements 22 as the pass band 38 is moved within a continuum of the nominal frequency range of-1.0 to 1.0.

By further analyzing the performance characteristics of the filter, such as power handling, intermodulation, or insertion loss, selection of an ideal coupling matrix from a set of coupling matrices that perform the same filter function may be driven. As demonstrated in co-pending patent application serial No. 12/163,837 (attorney docket number STI-008), entitled "Electrical Filters with Improved interaction distribution," minor changes in the internal structure of the filter can produce an enhancement in the performance characteristics of the filter terminals without changing the filter function, as can be seen from the S parameter at the measured input/output terminals, which is expressly incorporated herein in its entirety by reference. The techniques disclosed in U.S. patent application serial No. 12/163,837, including changing the order of transmission zeros, may be applied to the filter circuits disclosed herein.

As briefly described above, the filter 10 may be tuned using parameter extraction and analysis techniques, and then changing one of the non-resonant elements 22 to selectively shift the pass band 38 within a selected sub-band 36. In particular, the filter 10 may be operated at a desired operating temperature to determine a variety of initial or pre-tuned performance characteristics. For example, the HTS filter may be operated at 77 degrees K and measurements taken. Parameter extraction may then be performed by a network analyzer, for example. For example, the measured S-parameter response (e.g., return loss) may be used to determine various parameters related to the filter (e.g., resonator frequency and/or resonator-to-resonator coupling values). Next, the filter response can be optimized, for example, by a computer. The difference between the extracted filter characteristics and the optimized filter characteristics may then be determined and used to provide a tuning scheme. The filter may then be tuned according to the tuning scheme. In various embodiments, this tuning may be performed, for example, by selecting capacitors that are open or closed to adjust the pass band 38 within the selected sub-band 36 using the electrical controller 24. Once the filter has been tuned, the filter can be checked. For example, the filter may again be operated at its operating temperature and measured to determine new performance characteristics of the filter. If the newly tuned performance characteristics, such as frequency response and/or S-parameter response, are acceptable, the filter may be packaged for operation.

Another tuning technique for high performance planar filters involves the use of one or more tuning elements capable of tuning the filter. For example, referring to fig. 27, tuning elements in the form of

tuning forks

40, 42 may be arranged on the same substrate 44 as the resonator element 18, in the case shown in the form of a half-wavelength structure in a spiral in-spiral out shape. For purposes of illustration, only one resonant element 18 is shown in fig. 27, although a complete filter may include multiple resonant elements 18, as shown in fig. 1. In a multi-resonator planar filter, each resonant element 18 may have a

tuning fork

40, 42. For example, modifying the frequency of the resonant element 18 coupled to the

tuning forks

40, 42 by scribing can remove portions of the

tuning forks

40, 42 from the substrate 44, thereby displaying transmission zeroes along the stop band 32 corresponding to the frequency of the resonant element 18 relative to the reflection zeroes 34. In the case of tuning multiple resonant elements 18, the frequencies of the resonant elements 18 may be modified to simultaneously time shift the band 32 with the pass band 38 along the frequency range. The

tuning forks

40, 42 are coupled to one end of the resonant element 18 by a series of capacitors 46 arranged at an angle to each other.

Alternatively, the

tuning forks

40, 42 may be coupled directly to the resonant element 18. However, if the tuning fork is directly connected to the resonator, the series capacitor can be designed to reduce the tuning sensitivity to about 10% of the sensitivity that can be seen. This reduced sensitivity can support tuning by hand using, for example, a mechanical device such as a diamond scribe pen. The manual scribing may be performed under a microscope with a diamond scribing pen. Alternative means of scribing the

tuning forks

40, 42 may also be used, such as a laser scribing tool, a focused ion beam, or photolithography. In any case, to change the capacitance of the filter circuit, the resonator 18 may be tuned by physically disconnecting (e.g., scribing) portions of the

tuning forks

40, 42.

For precision and ease of tuning, the

tuning forks

40, 42 may include a coarse scale 48 and a fine scale 50, respectively, to provide ease of scribing for both coarse and fine tuning. The

scales

48, 50 may be associated with a tuning scheme. Although two

tuning forks

40, 42 are shown, any number of tuning forks may be used depending on the desired tuning range and tuning resolution.

A parameter extraction based technique can be used to determine the coupling and resonant frequency of the filter and to provide a scribed tuning fork solution. In this way, a filter design is provided that enables very precise tuning without any expensive tools.

As another example, the tuning element in the form of a trim tab 52 may be arranged on the same substrate 44 as the resonator element 18, as shown in fig. 28. The trimming tab 52 is located, for example, on the edge of the resonator that is trimmed (i.e., disconnected from the circuit) to reduce the shunt capacitance of the resonating element 18. The flattening patches 52 may have discrete values that shift the resonant frequency of the filter by different known amounts, and the amounts may be configured in binary columns.

For example, on each resonant element 18, the filter may have four trim tabs 52 that may shift the resonant frequency in a binary sequence (e.g., 1500KHz, 800KHz, 400KHz, 200kHz, and 100 KHz). In the illustrated embodiment, seven trim tabs 52 having different sizes are provided. Specifically, when trimming (trimming), the trim tab 52(1) causes a 1500KHz frequency shift of the resonant element 18; when leveled, the trim tab 52(2) causes an 800KHz frequency shift of the resonant element 18; when leveled, the trim tab 52(3) causes a 400KHz frequency shift of the resonant element 18; when trimmed, plate 52(4) causes a 200KHz frequency shift of the resonant element 18; and each trim tab 52(5) -56(7) causes a 100KHz frequency shift of the resonant element 18 when trimmed. Thus, as an example, if the resonant element 18 requires a frequency shift of 670KHz according to one tuning scheme, one of the trim tabs 52(2), 52(3) (200KHz), and 52(5) - (56) (7) may be removed from the substrate 44.

Further details of Tuning a resonator using a Tuning fork and a trimming blade are described in U.S. patent application serial No. 12/330,510, entitled Systems and Methods for Tuning Filters, which is incorporated herein by reference in its entirety.

Techniques based on parameter extraction can be used to determine the coupling and resonant frequency of the filter and to provide a trim tab 52 indicating that it is necessary to disconnect or trim from the resonator edges in order to produce a properly tuned filter.

Referring now to fig. 29, another tunable RF filter constructed in accordance with the present invention will now be described. RF filter 100 can be dynamically tuned to compensate for variations in operating temperature that might otherwise cause pass band 38 to inadvertently move from its nominal design position within a frequency range in a manner similar to the movement of pass band 78 shown in fig. 11. That is, changes in operating temperature cause the coupling values of the resonant and

non-resonant elements

18 and 22 to change from their nominal values (i.e., the reactance of the elements at the temperature at which the RF filter 100 is initially tuned). For example, the reactance of the non-resonant element 22 may vary by ± 1% per 10 ° change at operating temperature. Thus, the RF filter 100 can dynamically adjust the reactances of the resonant and

non-resonant elements

18 and 22 to return the pass band 38 to its nominal position within the frequency range.

The RF filter 100 is similar to the RF filter 10 shown in fig. 13a, except that the RF filter 100 additionally includes an electrical controller 124, a temperature sensor 126, and a memory 128. Like the electrical controller 24 shown in fig. 1, the electrical controller 124 is configured to adjust the non-resonant elements 22 to introduce and displace the reflection zeroes along the stop band 32 to move the narrow pass band 38 within the desired frequency range, and the frequencies of the resonant elements 18 can be further adjusted by tuning elements (not shown) to move the transmission zeroes along the frequency range to optimize the filter response. Unlike the electrical controller 24, the electrical controller 124 is configured to dynamically adjust the resonant elements 18 and non-resonant elements 22 to compensate for changes in operating temperature.

To this end, the electrical controller 124 obtains a measurement of the current operating temperature from the temperature sensor 126, accesses the look-up table from the memory 128, and adjusts the resonant elements 18 and the non-resonant elements 22 based on the look-up table. Specifically, the look-up table contains a plurality of reference operating temperatures (which may be distributed in 10 ° increments from-20 ° K to 100 ° K, for example) and a set of adjustment settings corresponding to each reference operating temperature. Each set of adjustment settings controls the reactance of one resonant element 18 or one non-resonant element 22. A typical set of adjustment settings would include adjustment settings that control the plurality of resonant elements 18 and non-resonant elements 22.

The electrical controller 124 applies adjustment settings to the resonant and

non-resonant elements

18 and 22 via electrical signals to adjust their electrical reactance in a manner that returns the pass band 38 to its nominal position within the frequency range. In particular, the electrical controller compares the measured operating temperature to reference operating temperatures in a look-up table, selects a set of adjustment settings corresponding to the reference operating temperature that best matches the measured operating temperature, and adjusts the reactances of the resonant and

non-resonant elements

18 and 22 in accordance with the selected set of adjustment settings.

In a preferred embodiment, the resonant elements 18 are adjusted in a manner to return the selected sub-band 36 to its nominal position within the frequency range, and the non-resonant elements 22 are adjusted in a manner to return the pass band 38 to its nominal position within the selected sub-band 36, similar to the tuning technique illustrated in fig. 5 a-5 d. Alternatively, the resonant elements 18 may be adjusted in a manner that does not return the sub-bands 36 to their nominal positions within the frequency range or not at all, in which case the non-resonant elements 22 may be adjusted in a manner that does not return the pass band 38 to its nominal position within the selected sub-band 36. In any case, the pass band 38 will be returned to its nominal position within the frequency range.

The nature of the adjustment settings will depend on the mechanism used to adjust the resonant and

non-resonant elements

18 and 22. For example, if each of the resonant and

non-resonant elements

18 and 22 includes a capacitor and a switch in parallel to form a variable capacitive circuit, each set of adjustment settings may include data indicating which switch to open to include the respective capacitor in the capacitive circuit or which switch to close to exclude the respective capacitor in the circuit in order to vary the reactance of the respective resonant element 18 or non-resonant element 22 in a manner that locates the pass band 38 at its nominal position within the frequency range or as close as possible to the nominal position within the frequency range of the resolution given by the look-up table. Thus, in this case, the look-up table will have an on-off state setting for the switched capacitor of each resonant element 18 and non-resonant element 22 for each measured operating temperature. The adjustment settings in the look-up table may be determined by exposing the filter 100 to each reference operating temperature and determining the adjustment settings for the resonant elements 18 and non-resonant elements 22 using the parameter extraction and analysis techniques described above.

Note that as shown in fig. 15 a-15 c, opening or closing the parallel capacitors that compensate for the operating temperature variations of the non-resonant elements 18 may include at least some of the parallel capacitors used to move the pass band 38 between the different sub-bands 36. Further, while the look-up table has been described as including adjustment settings for only one sub-band 36, the look-up table may include adjustment settings for a plurality of sub-bands 36. In this case, the adjustment setting for the particular sub-band 36 in which the pass band 38 is currently located may be used to move the pass band 38 to its nominal position within the frequency range in response to operating temperature changes.

While particular embodiments of the present invention have been shown and described, it should be understood that the above discussion is not intended to limit the invention to these embodiments. It will be apparent to those skilled in the art that various changes and modifications may be made without departing from the spirit and scope of the invention. For example, the invention has applications far beyond filters with inputs and outputs, and embodiments of the invention can be used to form duplexers, multiplexers, channelizers, reactive switches, etc., where low-loss selective circuits are used. Accordingly, the invention is intended to cover alternatives, modifications and equivalents, which may be included within the spirit and scope of the invention as defined by the appended claims.

Claims

1. A radio frequency, RF, filter (10) comprising:

a signal transmission path (12) having an input (14) and an output (16):

a plurality of resonant elements (18) arranged along the signal transmission path (12) between the input (14) and the output (16); and

a plurality of non-resonant elements (22) coupling the resonant elements (18) together to form a stop band (32) having a plurality of transmission zeroes (30) corresponding to respective frequencies of the resonant elements (18) and at least one sub-band (36) between the transmission zeroes (30), wherein the non-resonant elements (22) comprise at least one variable non-resonant element (22) for selectively introducing at least one reflection zero (34) within the stop band (32) to create a pass band (38) in a selected one of the at least one sub-band (36), wherein the stop band (32) is within the 800MHz-900MHz or 1,800MHz-2,220MHz frequency range;

An electrical controller (124) configured to receive an operating temperature of the RF filter and adjust the at least one variable non-resonant element (22) based on the received operating temperature to selectively move the at least one reflection zero (34) along the stop band (32) to move the pass band (38) within the selected sub-band (36).

2. The RF filter (10) of claim 1, wherein the at least one sub-band (36) comprises a plurality of sub-bands (36).

3. The RF filter (10) of claim 2, wherein the at least one variable non-resonant element (22) is configured to displace the at least one reflection zero (34) along the stop band (32) to create the pass band (38) within a selected one of the sub-bands (36).

4. The RF filter (10) of claim 3 wherein the pass band (38) has a different bandwidth within the selected sub-bands (36).

5. The RF filter (10) of claim 2, wherein the at least one variable non-resonant element (22) is configured to displace at least one other reflection zero (34) within the stop band (32) to create another pass band (38) within another one of the sub-bands (36).

6. The RF filter (10) of claim 1, wherein the electrical controller is further configured to vary the reactance of each non-resonant element by varying the capacitance of the capacitive circuit by operating a switch to selectively include or exclude at least one capacitor of the capacitive circuit to thereby selectively move the reflection zero within the stop band to move the pass band within the selected sub-band.

7. The RF filter (10) of claim 1, wherein the at least one reflection zero (34) comprises a plurality of reflection zeros (34).

8. The RF filter (10) of claim 1, wherein the at least one variable non-resonant element (22) comprises a plurality of variable non-resonant elements (22).

9. The RF filter (10) of claim 1, comprising a plurality of tuning elements (20) configured to adjust the frequency of the resonant elements (18).

10. The RF filter (10) of claim 9, wherein the tuning element (20) is configured to adjust the frequency of the resonant element (18) to move the transmission zero (30) along a frequency range.

11. The RF filter (10) of claim 9, wherein the plurality of tuning elements are configured to modify the frequencies of the resonant elements (18) to simultaneously displace the stop band (32) and the pass band (38) along a range of frequencies.

12. The RF filter (10) of claim 1, wherein the at least one variable non-resonant element (22) has an adjustable susceptance.

13. The RF filter (10) of claim 1, wherein the at least one variable non-resonant element (22) comprises at least one of: variable capacitors, low loss switches, varactors, and switched capacitors.

14. The RF filter (10) of claim 1 wherein each of the resonant elements (18) comprises a thin film lumped element structure.

15. The RF filter (10) of claim 14, wherein the thin-film lumped element structure comprises a High Temperature Superconductor (HTS).

16. The RF filter (10) of claim 1, the electrical controller configured to generate an electrical signal that adjusts the non-resonant element (22).

17. The RF filter (10) of claim 1, wherein each of the resonant elements (18) comprises an acoustic wave resonator.

18. The RF filter (10) of claim 1, wherein the plurality of non-resonant elements (22) includes a first plurality of non-resonant elements (22(1)) coupled in parallel with the resonant elements (18), respectively, and a second plurality of non-resonant elements (22(3)) coupled in series with the resonant elements (18), respectively, wherein the first plurality of non-resonant elements (22(1)) includes the at least one variable non-resonant element (22).

19. The RF filter (10) of claim 1 wherein each of the non-resonant elements (22) has a plurality of capacitors (C) coupled in parallel with each other to form a capacitive circuit₁-C₃) And to at least one of said capacitors (C)₁-C₃) At least one switch (S)₁-S₂) Wherein the electric controller is configured to control the at least one switch (S)₁-S₂) To selectively make said capacitive circuit include or not include at least one of said capacitors (C)₁-C₃) To vary the capacitance of the capacitive circuit and thereby the reactance of each non-resonant element to selectively move the reflection zeroes within the stopband to move the passband in the selected subband.

20. The RF filter (10) of claim 19 wherein each of the non-resonant elements has three capacitors (C1-C3) in parallel, with the outer two capacitors (C1-C2) having respective switched capacitances and a boost switch (S)₁-S₂) Is connected in series with a resistor (R1-R2) for resistive losses by closing the switch (S)₁-S₂) Including the capacitor (C1-C2) in a circuit, and by independently opening the switches (S)₁-S₂) The capacitor (C1-C2) is excluded from the circuit.

21. The RF filter (10) of claim 1, wherein each of the resonant elements (18) has two tuning forks (40, 42), the resonant elements (18) being tuned by physically disconnecting portions of the tuning forks (40, 42), the two tuning forks (40, 42) being disposed on a substrate (44).

22. The RF filter (10) of claim 21, wherein the tuning forks (40, 42) are configured to modify the frequencies of the resonant elements (18) to simultaneously displace the stop band (32) and the pass band (38) along a range of frequencies.

23. The RF filter (10) of claim 21, wherein a portion of the tuning fork (40, 42) is configured to be removed from the substrate (44) via a laser scribing tool, a focused ion beam, or photolithography.

24. The RF filter (10) of claim 20 wherein each of the resonant elements (18) includes seven balancing tabs (52), one of which is removable to reduce the bypass capacitance of the respective resonant element (18).

25. The RF filter (10) of claim 24, wherein the trim tabs (52) are located at edges of the respective resonant elements (18), the trim tabs having discrete values that shift the resonant frequency of the filter by different known amounts and configure the known amounts into a binary sequence.

26. The RF filter (10) of claim 1, further comprising an electrical controller (124) configured to receive an operating temperature of the RF filter and adjust the at least one variable non-resonant element (22) based on the received operating temperature to selectively move the at least one reflection zero (34) along the stop band (32) to move the pass band (38) within the one sub-band (36).

27. The RF filter (10) of claim 26, further comprising a temperature sensor (126) configured to measure the operating temperature, wherein the electrical controller (124) is configured to receive the measured operating temperature from the temperature sensor (126).

28. The RF filter (10) of claim 26, further comprising a memory storing a look-up table containing a plurality of reference operating temperatures corresponding to different ones of the operating temperatures respectively and a set of adjustment settings corresponding to each reference operating temperature, wherein the electrical controller (124) is configured to compare the measured operating temperature with the plurality of reference operating temperatures in the look-up table, select a set of adjustment settings corresponding to the reference operating temperatures that best matches the measured operating temperature, and adjust the reactances of the resonant element (18) and the non-resonant element (22) according to the selected set of adjustment settings.

29. The RF filter (10) of claim 26, wherein the electrical controller (124) is configured to adjust the at least one variable non-resonant element to selectively move the at least one reflection zero (34) along the stop band (32) to return the pass band (38) to a nominal design position within a frequency range.

30. The RF filter (10) of claim 29, wherein the electrical controller is configured to adjust at least one of the resonant elements (18) based on the received operating temperature to selectively move the transmission zeroes (30) corresponding to each frequency of the at least one resonant element (18) along the stop band (32) to return the pass band (38) to the nominal design position within the range of frequencies.

31. A method of tuning the RF filter (10) of claim 21, removing a portion of the tuning fork (40, 42) from the substrate (44) by scribing, thereby modifying the frequency of the resonant element (18).

32. The method of claim 31, wherein portions of the tuning forks (40, 42) are removed from the substrate (44) via a laser scribing tool, a focused ion beam, or photolithography.

33. The method of claim 31, wherein each of the resonant elements (18) includes seven trimming tabs (52), one of the trimming tabs being removed to reduce a bypass capacitance of the respective resonant element (18).

34. The method of claim 33, wherein the trim tabs (52) are located at edges of the respective resonant elements (18), the trim tabs having discrete values that shift the resonant frequency of the filter by different known amounts and configure the known amounts into a binary sequence.