MXPA00009380A - Dual stator winding induction machine drive - Google Patents

Abstract

An induction machine (20) adapted for variable speed operation, including low and zero speed, comprising:a polyphase stator having two separate stator windings (abc, xyz) with separate input terminals for each winding by which drive power may be applied separately to the two windings, wherein the two windings have different number of poles that are in the ratio 1:3 and a squirrel cage rotor mounted within the stator.

Description

MECHANICAL CONTROL OF INDUCTION OF DUAL STATEROIC WRAPPING FIELD OF THE INVENTION This invention is generally related to the field of electric motors and with controls for said motors, and particularly with mechanical induction controls.

BACKGROUND OF THE INVENTION The use of a common magnetic structure that is shared by two sets of stator windings was first introduced in the late 1920s as a way to increase the power capacity of large synchronous generators. See, P.L. Alger, et al., "Double Windings for Turbine Alternators," AIEE Transactions, Vol. 49, January, 1930, pp. 226-244. Since then, dual stator machines have been used in many other applications. These include synchronous machines with AC and DC outputs. P.W. Franklin, "A Theoretical Study of the Three-Phase Salient Pole Type Generator with Simultaneous AC and Bridge Rectified DC Output," IEEE Transactions on Power App. And Systems, Vol. PAS-92, No. 2, March / April 1973, pp. . 543-557. The dual stator machines have also been used as current source converters for large pumps, compressors and laminators activated by induction machines. T. Kataoka, et al., "Dynamic Control of a Current-Source Inverter / Double-Wound Synchronous Machine System for AC Power Supply," IEEE Transactions on Industry Applcations, Vol. IA-17, No. 3, May / June 1981, pp. 314-320. Another purpose for the use of dual stators has been to improve the reliability at the system level. See, for example, J.R. Fu, and others, "Disturbance Free Operation of a Multiphase Current Regulated Motor Drive with an Open Phase," IEEE Transactions on Industry Applications, Vol. 30, Sep./Oct. 1994, pp. 1267-1274; J.C. Salmon, and others, "A Split-Wound Induction Motor Design to Improve the Reliability of PWM Inverter Drives," IEEE Transactions on Industry Applications, Vol. IA-26, No. 1, Jan.Jef. 1990, pp. 143-150. The dual stator machines are usually constructed by "splitting" the stator winding into two displaced but identical windings. See, for example, E.F. Fuchs, et al., "Analysis of an Alternative with Two Displaced Stator Windings," IEEE Transactions on Power App. And Systems, Vol. PAS-93, No. 6 NovJDic. 1974, pp. 1776-1786. However, dividing the stator winding in this way results in a mutual coupling between the stators, causing circulating harmonic currents. K. Gopakumar, et al., "Split-Phase Induction Motor Operation Form PWM Voltage Source Inverter," IEEE Transactions on Industry Applications, Vol. 29, No. 5, Sep./Oct. 1993 pp. 927-932. Said stator winding machines divided in this way have a great disadvantage because the circulating currents increase the stator losses and demand a higher semiconductor device value. In addition, there is coupling between the electromagnetic torques produced by each stator winding. See, T.A. Lipo, "A d-q Model for Six Phase Induction Machines," International Conference on Electric Machines, Athens, Greece, 1980, p. 860-867.

COMPENDIUM OF THE INVENTION A dual stator winding induction machine according to the invention has two polyphase windings with input terminals available to be supplied separately with control power. The two stator windings are wound with a different number of poles to essentially eliminate the magnetic coupling between the two stator windings and to decouple the twists produced by each set of stator windings. In addition, the circulating harmonic currents found in conventional dual stator winding machines due to the so-called mutual leakage coupling are completely eliminated. Since the output torque corresponds to the algebraic sum of two independent torsions, the stator frequency is no longer determined solely by the rotor speed and the slip frequency, but by the added variable of a second torsion component, adding an additional degree of freedom to the system for greater control flexibility. The dual stator winding machine supplied with power from two separate variable frequency drive controls according to the invention provides two independently controllable torsion components, thereby enabling the low frequency operation of the machine (still at rest) to be improved. This characteristic is particularly important for the operation of constant control of volts per hertz at zero speed, where the influence of stator resistance becomes dominant. In the present invention, the operation at zero speed does not require a zero excitation frequency for the two power controls, thereby significantly reducing the effect of the resistance voltage drop. The dual stator machine of the present invention can be constructed with minimal modifications to standard winding configurations, without requiring any structural modification in the stator structure. By providing two stator windings the reliability of the machine is also increased compared to machines with a single stator winding, while improving the use of the magnetic material for normal operation. A particular advantage of the present invention is the ability to operate the machine in a zero-speed, low-speed operation without the need for a rotor position encoder to provide rotor position and velocity feedback. The speed operation at zero can be obtained by applying control power to the two windings at a frequency and energy level to provide balanced torques opposite the rotor. The implementation of a vector control without detectors, in this way, is facilitated since the control energy supplied to the two stator windings will always be above the zero frequency. The stator of the machine of the present invention is constructed by dividing the polyphase (typically three-phase) simple normal winding into two separate (eg, three-phase) windings wound by a different number of poles. Although any combination of different number of poles can be used, to better utilize the magnetic material, and to avoid localized saturation and additional stator losses, according to the invention, it is preferred that a combination of 2 poles and 6 poles be used. For better utilization of magnetic material, this pole number configuration provides an almost trapezoidal magnetomotive force (MMF) distribution while limiting the maximum number of poles to provide a good power factor and efficiency. Other objects, features and advantages of the invention will be apparent from the following detailed description when taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 is a schematic diagram showing the distributions of the dual stator windings in the machine of the present invention. Figure 2 is a graph illustrating the 6-pole peak of the MMF for a constant total peak MMF. Figure 3 is a graph showing velocity-torsion curves for the two stator windings in a first mode of operation where torsions are added. Figure 4 is a graph showing velocity-torsion curves for stator windings for a low speed operation where two torques applied by two stator windings oppose each other. Figure 5 is another illustrative view of a dual poles-6 pole stator winding induction machine according to the invention. Figure 6 is a diagram illustrating the distribution of windings in the machine of dual stator windings for a fractional pitch and a variable shift angle,, with a phase width of 60 °. Figure 7 is a diagram like the one in Figure 6 illustrating the distribution of windings for a fractional step and a variable displacement angle? with a phase width of 30 °. Figure 8 is an illustrative view of a short circuit rotor that is used in an induction machine of the present invention, illustrating rotor currents. Figure 9 is a schematic circuit diagram equivalent for the 2-pole winding of the dual stator winding machine of the invention. Figure 10 is a schematic circuit diagram equivalent for a "P" pole winding for the dual stator machine of the invention. Figure 11 is a schematic diagram and a control system incorporating the dual stator winding machine of the invention that implements a constant operation V / f. Figure 12 is a schematic diagram of a control system incorporating the dual stator winding machine of the invention, which implements the field-oriented operation. Figure 13 are graphs illustrating simulation results of complex vector model and total matrix for the dual-pole, 6-pole, 6-pole winding machine of the invention. Figure 14 is a schematic diagram illustrating the use of two separate converters to activate a dual stator winding induction motor according to the invention. Figure 15 is a schematic diagram illustrating a converter having separate sections for activating a dual stator winding induction motor according to the invention. Figure 16 are diagrams illustrating the winding distribution for a 4 pole winding in a dual stator winding induction motor according to the invention. Figure 17 is a diagram illustrating the distribution of windings for a 12-pole winding of a dual stator winding motor according to the invention.

DETAILED DESCRIPTION OF THE INVENTION The stator of the machine of the invention is constructed by dividing the normal polyphase (for example, three-phase) stator winding into two separate polyphase windings, windings to obtain a different number of poles. Three-phase motors are by far the most common type, but it should be understood that the present invention can be used with machines having two or more phases. Any combination of different number of poles can be used; however, to better utilize the magnetic material, and to avoid localized saturation and additional stator losses, it is found that the most advantageous combination is a combination of 2-6 poles. Said arrangement is incorporated in the machine shown generally at 20 in Figure 1. The machine 20 according to the invention has a short-circuited rotor 21 with rotor rod conductors 22 around its periphery in a conventional manner, and is separate by an air gap 24 of the stator 25. The rotor is mounted to rotate within the stator in a conventional manner. Figure 1 illustrates the physical arrangement of the dual stator windings, a first bipolar winding abe and a second winding of 6 poles xyz. For simplicity of illustration, the metal structure and magnetic material of the stator, which are entirely conventional, are not shown in Figure 1. Each of the windings abe and xyz extends to the three external terminals (not shown in the Figure). 1) by which energy is supplied independently to each of the two windings. From the perspective of using magnetic material it is convenient to choose a combination of number of poles that, in their uniform state, tend to produce an almost trapezoidal MMF distribution. This type of distribution can be obtained more efficiently by choosing the number of poles in a ratio of 1: 3, for example, 2 and 6 poles, 4 and 12 poles, etc. On the other hand, the magnetization inductance varies inversely proportional to the square of the number of poles, thus a machine with a large number of poles results in a low energy and low efficiency factor. In addition, to achieve a sinusoidal winding distribution, the stator winding must be distributed among several slots, and, for a given internal stator diameter, the number of slots per pole decreases in proportion to the number of poles. Also, for a given rotor speed, the stator frequency increases directly proportional to the number of poles. This results in additional losses, in the machine and in the energy converter, further reducing efficiency. All these factors suggest that the maximum number of poles should be kept to a minimum, and thus the best combination is 2 and 6 poles. However, other combinations of poles can be used, which are within the scope of the invention. The total MMF distribution in the air space corresponds to the sum of MMFs produced by each stator winding. In order to avoid the presence of highly saturated points and, at the same time, to use the magnetic core in its entirety, it is desirable to maintain the total peak flow density distribution equal to that created by a two-pole winding acting on its own. Figure 2 shows a peak magnitude of a 2-pole MMF as a function of a 6-pole peak MMF, for a constant peak MMF. The optimal distribution corresponds to choosing a 6-pole MMF equal to approximately 40% of that of the 2-pole distribution. In this case the total MMF and the 2-pole MMF have the same peak amplitude, thus preserving the saturation level. The rotor 21 of the machine 20, preferably, corresponds to that of a short-circuit type of rotor. This construction ensures that both stator current constructions are simultaneously coupled with the rotor flow to produce the desired torque. Because the decoupling effect produced by the difference in the number of poles, the dual stator machine 20 behaves like two independent induction machines that are mechanically coupled through the rotor shaft. Therefore, all the known control techniques used in the controls of induction machines can also be applied to the dual stator winding machine. These include both controls of volts per hertz (V / f) scalar constants as vector controls or field orientation (FO). The basic control method involves generating two torsion commands that, when combined, produce the required resultant torque. By choosing appropriate current commands, the two individual torsions can be added or subtracted, thus providing the ability to control the excitation frequency. The two different modes of operation are possible: low speed (below a selected minimum speed) where two twists produced by winding abe and winding xyz are subtracted (opposite one another) as shown in Figure 4, and medium to high velocity (above the minimum velocity) where the twists are added as shown in Figure 3. A detailed but simple dynamic modelThe machine can be developed using the following general assumptions: negligible saturation, uniform air gap, sinusoidal distributed stator windings, no electrical interconnection between the stators, and negligible interbar current. It is also assumed that the two stator windings are wound to obtain 2 and 6 poles, respectively, and that one stator is displaced with respect to the other by an angle? but arbitrary. The main stator (2 poles) is denoted as the abe windings and the second stator, having 6 poles, like the xyz windings, as illustrated in Figure 1. The rotor of the machine is a standard short-circuit rotor. A simplified diagram showing the relative placement of the windings and their angular relationships is given in Figure 5. Since both stators are sinusoidally distributed in space but windings to obtain a different number of poles (and electrically isolated) there is no mutual coupling due to the main flow between them. However, since both windings share common grooves and are close to each other, there is a common leakage flow that binds them. Hence the so-called mutual leakage coupling. The total flow linked by the stator windings and due only to the stator currents is abc and isxyz can be written, in matrix form as: where: Ls1 and Ls2 represent the self inductance matrices of the windings abe and xyz, respectively. They have the form of: (3) The magnetization inductance Lmsl is known as: 2 3 where Nsi is the total number of turns per phase of each set 4 of windings and P is the number of poles. / s, represents the 5 auto-leakage inductance per total phase of each winding and can be calculated using traditional methods. 7 The sub-matrices Ls12 and s2i justify the mutual leakage coupling 8 between the two stator windings. In general, the leakage flow can be divided into slot components, end winding, band and zigzag and each of them will contribute to the auto 11 leakage inductance and the mutual leakage inductance. However, 12 for simplicity, the mutual leak, due to the 13-gauge and zigzag leakage components, will not be taken into account ^ and it will be assumed that 14 only contribute to the auto-leak. Therefore, it is assumed that only the end winding and slot 16 components contribute to the mutual leakage. In addition, it is assumed that the leakage of 17 end windings varies as does the slot leak. 18 The winding distributions of Figures 6 and 7 can be used to study the mutual leak. The illustrative winding distribution 20 shown in Figure 6 corresponds to the fractional step 21 due to the displacement? between the stators, a 22 phase width of 60 °, and in the distribution of Figure 7, a 23 phase width of 30 ° corresponds to the fractional step ,. Since i í 24 the two sets of windings have 2 and 6 poles respectively, their angles of passage a, and a2 are in a 26 ratio of 6/2. In Figure 6, define p, and p2 as the step of 27 the windings abe and xyz respectively, for a variation of? 28 between 0 and 20 °, the corresponding step factors vary as 1 8/9 < p! < 1 and 2/3 < p2 < 1. 2 Slit leak can be divided into auto leak and leak 3 mutual. The auto drain represents that part of the flow produced by the 4-phase current component (ie, 5-bovine phase slots representing the same phase). The mutual leak 6 corresponds to the leakage flow because the conductors of 7 different phases share common grooves. In general, for a self-winding 8, (Lsis), and a mutual winding, (Ls / m) two-layer, 9 the components of the slot leakage inductance can be express, as a function of step or, by: 11 12 L *. = K p) L nm (6) 13 14 where L? T and LIB are the inductance of associated slot leaks with the bovines in the upper and lower halves of the grooves. 16 They are calculated for the unit pass case and do not depend on the 17 winding step. The term L | TB represents the mutual inductance 18 between the bovines in the upper and lower halves of the groove. 19 The quantities ks and km are called slot factors and correspond to the constants of proportionality that depend 21 of the step. 22 For the dual statoric machine of the invention, it can be 23 demonstrate that both stator windings are fully uncoupled and that the total flow linked by the windings statorics can be written as 26 27? Wér ™ (7) for the primary winding and • * HJT ~ i tlltxa jLtu-il, (8) for the secondary winding. The matrices Lsr1 and Lsr2 describe the mutual coupling between the stator and the rotor circuits and can be determined using winding functions. When using complex vector representations, the stator flux associated with the abe winding can be written as 1 -tT * J 3 r \ i. 2 n "ssimn < "5 t? #, *« N .. 2 '(9) *, where n is the number of rotor bars, d is half the angle between the rotor bars, and the complex vector currents ir1 are defined by with a = ej2p 3 and b = ej2p n. The vector [irl) ir2 ... im] t represents instantaneous rotor currents, defined according to Figure 8, which illustrate the currents flowing in the rotor bars 22 and in the end rings 27 connecting the 1 bars 22. 2 A similar analysis can be performed for the winding x and z. 3 Furthermore, it can be shown that the stator current is1 4 depends only on the applied voltage y_s1 and the rotor current ir1. Similarly, the stator current is2 depends only on the applied voltage vs2 and the rotor current ir2. 7 This result is consistent with the fact that, for a sinusoidal distributed 8 winding, there are only 9 couplings between the current distributions of the same number of poles. Although the instantaneous rotor current distribution 12 simultaneously contains two components of different frequencies and of pole number, each stator field is able to interact only with that part of the rotor field with the "correct" number of poles. This is true not only on an average basis but also instantaneously. It is well known 17 that the sinusoidally distributed windings are only coupled with winding fields for the same number of poles; however, the short circuit rotor is clearly not a sinusoidal winding and one can expect that the presence of two superimposed flow distributions would accommodate pulses 22. However, this is not the case for a dual stator winding machine. An equivalent circuit using a 24 connotation d-q, is shown in Figures 9 and 10 for the i i 25 windings of two poles and pole P respectively. 26 By not taking into account saturation, torsion 27 electromagnetic can be expressed as the partial variation of Ja 28 co-energy with respect to the position 29 2 which can be described as the separate sum of the torsions 3 produced by each set of stator currents 4 6 7 substituting the corresponding matrices and carrying out the 8 differentiation results in the torsion as: 9 11 12 where P = 6 is used. Since I go? and L-2 are orthogonal vectors, the two torsion components can be controlled independently by means of the stator currents. As mentioned above, because the machine of the invention behaves like two independent induction machines 17, mechanically coupled through the shaft, all known control techniques used in the controls of the induction machine are also can apply to the dual stator winding machine. 21 In general, there are two distinct modes of operation, the ii 22 low speed range (ie, frequencies below a minimum frequency, eg, a few hertz, and the range of 24 medium to high speed. low speed range, objective 25 is to keep the winding frequency of two poles above a minimum level (typically around 3 hertz) and the torque is controlled by adjusting the winding frequency of 6 poles. of this preset limit, the influence of the stator resistance is minimized, with this simplifying the control In this mode the two MMFs rotate asynchronously, but due to the reduced frequency, the additional losses caused by the saturation are minimal. medium to high speed, the negative effect of the stator resistance is not a problem and the frequencies are kept in the same proportion as the number of poles, for example, proportional ion of 1: 3. This restriction guarantees an almost trapezoidal flow restriction and the torsion is controlled by adjusting the magnitude of the voltages applied. The trapezoidal shape, in turn, allows a slightly greater flow of 2 poles than when only the 2-pole winding is excited, with this producing a slightly greater torque for each ampere. The operation and control can be explained with reference to Figures 3 and 4. For high speed, the stators are fed with voltages with frequencies in a ratio of 1: 3 to produce the torsion / speed curves of Figure 3. Torsion The resultant for a given rotor speed corresponds to the algebraic sum of the torsions T, and T2 produced by each of the stators. The torsion produced by each winding can be controlled by adjusting the magnitude of the stator voltages supplied to each winding. When both stators are fed with different effective frequencies, the result is that shown in Figure 4. When setting the frequency f-, of the energy supplied to one of the stator windings say abe, the resulting 1 total torque can be adjusted by controlling the frequency f2 (and the voltage), 2 supplied to the xyz winding. As shown in Figure 4, a 3 increase in torsion requires an increase in / 2t and vice versa. 4 In this case, the first of the stator windings (abe) operates in the motor region while the other (xyz) operates as a 6 generator. Note that this mode of operation corresponds to 7 that required to operate at zero speed, and that the torsion 8 can be controlled from zero to a graduated value. 9 A block diagram is shown in Figure 11 Simplified control scheme for constant operation 11 V / f. As shown in this figure, the abe winding receives 12 three-phase power at the input terminals 30 from a 13 PWM 31 three-phase voltage source converter, while the 14 xyz windings receive power at terminals 32 from a separate PWM three-phase voltage source converter 34. The 16 speed reached? ", Where it is provided to a union 17 additioner 35 where it is compared with the calculated speed BJ, and 18 the difference is provided to a speed controller 37, 19 for example, a proportional - integral controller (Pl). The result of the speed controller is provided to a circuit of 21 frequency limit detection 38, which is provided 22 with a minimum frequency that can be selected, / min. He 23 circuit 38 is connected by lines 39 to provide signals 24 control to switches 40, 41 and 42. If the frequency of l. { 25 entry is greater than / m? N, the exit of circuit 38 on a line 43 26 is the same as the input and the switches 40, 41 and 42 are in the 27 positions shown in Figure 11 (40 open, 41 and 42 28 closed). Yes The frequency of entry to circuit 38 is less than 29 / min, the output frequency of the circuit is set to the value / min, and 1 the switches are activated so that the switch 40 is closed 2 and the switches 41 and 42 open. The result fC of the circuit 38 3 is used as the reference frequency for a function of 4 modulation V / f 43 and a carrier frequency function cos (? 1t) 5 44, whose results are multiplied and applied as the input to the 6 converter of voltage source 31 to provide power of the abe winding command 7 (eg, 2-pole) at terminals 8 30. The result of speed controller 37 is also provided to switch 40 and thence to connection 10 adding 46, which also receives, from a gain unit 47 11 through switch 41, an input equal to 3 /, *. The result 12/2 * of the add-on junction 46 is equal to 3 // when the switch 13 41 is closed, and is equal to the result of the speed controller 37 when the switch 40 is closed and the switch 15 41 is open. The signal f2 'is provided to a circuit 16 composed of a V / f function 50 and a cos function (? 2t +?) 51, 17 whose results are multiplied and applied to the voltage source converter 34 to provide the energy command of 19 xyz winding (for example, 6-pole) to the terminals 32. The energy applied to the machine 20 at the input terminals 30 and 21 32 are fed back via lines 55 and 56 for the 22 calculators of speed and flow 57 and 58, respectively. The 23 calculating circuit 58, of a conventional design, provides a 24 calculation XÜ of motor speed on a line 60 back to the i i 25 add-on junction 35. The calculating circuits 57 and 58 also 26 provide indicative signals of the flow applied by the 27 stator windings, whose signals pass through the 28 phase angle measurement circuits 61 and 62 and add up in the 29 add-on junction 63. The calculated flux of the winding xyz (6 poles) has three times the flow frequency of the abe winding; for this reason a frequency divider block 64 is used in calculating the flow of the calculator 58. The phase difference of the junction 63 is provided via the switch 42 to a control circuit 66 (eg, Pl), whose result is provided as a difference from phase 0 to circuit 51. A simplified block diagram of a vector controller is provided in Figure 12. As in the V / f constant method, the vector control operation is divided into two regions of operation: a high-speed range defined by frequencies above a minimum frequency / m? n and a low-speed range for frequencies below m? n- For the high-speed region, the controller divides the resulting portion between the two windings to produce similar stator currents and an almost trapezoidal flow distribution. In the low speed range, a negative torsion command is provided to the secondary winding (eg, xyz), thereby increasing the torque produced by the primary winding that occurs at an increased stator frequency. The objective is to maintain the primary stator frequency at a constant value equal to / m? N- The torsion command input T * and the minimum frequency / m? N are provided to a torsion-divider control circuit 70, which provides output command signals for the two windings T and T2 \ for the windings abe and xyz, respectively. The torsion command T ^ is provided to an add-on joint 71, which also receives a torsion feedback calculation T, * on line 72, and the difference is provided to a torque controller 74. A flow command signal and a flow feedback signal is provided to an add-on junction 75, the result of which is provided to a flow controller 76. The results of the flow controller 76 and the torque controller 74 are supplied to a transformation circuit 77, which also receives a signal which is a calculation of the rotor flux angle of a calculator circuit 80. The transformer circuit 77 provides current command signals to a three-phase PWM converter controlled by current 81, which provides output power on lines 30 to winding abe The energy signals on the lines 30 are also supplied to a torsion controller feedback circuit 83, the result of which is provided to a torque calculator 84 to provide the torsional calculation on the line 72. Similar components, designated by similar numbers with a prime notation, "'", are used in the control circuit to provide the command power in lines 32 to the winding x and z. The torsional divider used in the control loop for the field-oriented control strategy of Figure 12 works as follows: given the external torsion command and the limit frequency fmin the torsion command T ^ and T2 * is set so that the two frequencies supplied are in a ratio of 1: 3 and the lowest frequency (2-pole winding) is above fmin. If the required frequency is less than the minimum value, the frequency reached in the 2-pole winding is set to the value fmin and the torque command T2"is adjusted so that the resulting torque corresponds to the torsion reached externally. The results obtained from a machine space vector model 20 and those obtained from a full machine model of the machine are shown in Figure 13 to run at a free acceleration of 60Hz (f2 = 180Hz). Both the complex and the whole matrix model are superimposed, both simulations provide essentially identical results, demonstrating the validity of the complex vector model.As shown in Figure 13, the rotor currents contain two different frequencies dictated by the frequency of each the stator currents and the mechanical speed of the rotor, although the rotor currents simultaneously produce two field distributions that rotate At different speeds, due to the different number of poles and the sinusoidal characteristics of the stator windings, they do not provide harmonic torques. The present invention can be implemented using two separate converters 31 and 34 to provide control power to the induction motor windings 20, as illustrated in Figure 14. The first converter 31, provides control power to the input lines 30 of the abe winding, receives power through busbar lines CD 100 and 101, with an illustrative CD busbar capacitor shown at 102. The semiconductor switches S1-S3 and S7-S9 are appropriately controlled to provide the desired control power to the dual stator induction motor. The second converter 34 receives command power from the separate CD busbar lines 104 and 105 with a CD busbar capacitor illustrated illustratively at 106. The semiconductor switches S4-S6 and S10 and S12 operate to provide power to the xyz windings in the entry lines 32. Although not necessary according to the invention, it is convenient to use the so-called modulated amplitude drives of current-regulated pulsation (CRPWM). Alternatively, as shown in Figure 15, converters 31 and 34 can be implemented using a single set of busbar lines CD 110 and 111 with a CD bus capacitor shown in a manner illustrated at 112. Switches S1-S3 and S7-S9 operate through the busbar lines CD 110 and 111 to provide the control power to the input lines 30, while the switches S4-S6 and S10-S12 operate to provide command power in the bus lines. entry 32 to the windings xyz. The two converter sections 31 and 34 can then operate independently of one another although they use the same CD bus lines. The present invention can be implemented using combinations of windings different from those of 2 poles to 6 poles, for example, 4 poles to 12 poles or more. Figure 16 illustrates at 120 the 4-pole winding distribution for a 4-pole machine at 12 poles, and in the upper diagram labeled 121, the corresponding distribution of the abe windings at the top of the slots (space side of air). Figure 17 illustrates the 12-pole winding distribution at 130, and diagram 131 illustrates the distribution of the windings x and z at the bottom of the groove (fork side) of the machine. It should be understood that the invention is not confined to the particular embodiments set forth herein as illustrative, but encompasses all forms thereof as soon as they fall within the scope of the following claims.

Claims (16)

1. An induction machine adapted for a variable speed operation, including low and zero speed, comprising: a) a polyphase stator having two separate stator windings with separate input terminals for each winding by which the control power can be applied by separated to the two windings, where the two windings have a different number of poles which is in the ratio of 1: 3; and b) a short-circuited rotor mounted within the stator.

2. The machine according to claim 1, wherein one of the stator windings has 2 poles and the other stator winding has 6 poles. The machine according to claim 1, wherein one of the stator windings has 4 poles and the other stator winding has 12 poles. The machine according to claim 1, wherein one stator winding is displaced with respect to the other by a selected angle. 5. An engine control system comprising: a) a polyphase induction motor comprising a stator having two separate stator windings with separate input terminals for each winding by which the control power can be applied separately to the two windings, where the two windings have a different number of poles which is in the ratio of 1: 3, and a short circuit rotor mounted within the stator; b) a first power converter connected to the input terminals of a first stator winding having the lowest number of poles, the first power converter provides output power at a frequency that can be selected; and c) a second power converter connected to the input terminals of a second stator winding that has the highest number of poles, the second power converter provides output power at a frequency that can be selected separately from the power frequency output provided by the first power converter. 6. The motor control system according to claim 5, wherein one of the stator windings has 2 poles and the other stator winding has 6 poles. The motor control system according to claim 5, wherein one of the stator windings has 4 poles and the other stator winding has 12 poles. The motor control system according to claim 5, wherein the frequency of the control power of the second converter is three times the frequency of the control power of the first converter. The motor control system according to claim 5, further comprising control means connected to the first and second converter to provide control signals thereto for controlling the converters for an operation of volts per constant hertz. The motor control system according to claim 9, wherein the control means operates in a first mode at a low engine speed below a selected minimum speed and in a second mode above the minimum speed selected, wherein the control means in the first mode controls the second converter to provide control power to the second winding that applies twisting that opposes the torsion applied by the first winding activated by the energy of the first converter under the control of the Control means; and in the second mode the control means control both converters to apply power to the first and second windings to apply twist in the same direction. The motor control system according to claim 10, wherein in the second mode the control means controls the converters so that the second converter provides power at three times the frequency of the energy provided by the first converter. The motor control system according to claim 5, which includes control means connected to the first and second converters for controlling the converters in a field-oriented operation. 1

3. A method for controlling a polyphase induction motor having two separate stator windings having a different number of poles and a short-circuited rotor within the stator, comprising the steps of: a) applying energy at a first frequency to a first winding that has the smallest number of poles; and b) separately applying energy at a second frequency to the second winding having the highest number of poles. The method according to claim 14, wherein in the step of applying energy to the second winding, the energy is applied at a frequency such that the torsion applied by the second winding opposes the torsion applied by the first winding. The method according to claim 13, wherein in the step of applying energy to the second winding, the energy is applied so that the torque applied by the second winding is added to the torsion applied by the first winding. The method according to claim 15, wherein the second winding has three times the number of poles as the first winding, and the frequency of the energy applied to the second winding is three times the frequency of the energy applied to the first winding . SUMMARY An induction machine (20) adapted for a variable speed operation, including low and zero speed, comprising: a polyphase stator having two separate stator windings (abe, xyz) with separate input terminals for each winding by which the Command energy can be applied separately to the two windings, where the windings have different number of poles that are in the ratio of 1: 3 and a short circuit rotor mounted inside the stator.