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16 pages, 5705 KiB  
Article
Performance and Characterization of Additively Manufactured BST Varactor Enhanced by Photonic Thermal Processing
by Carlos Molina, Ugur Guneroglu, Adnan Zaman, Liguan Li and Jing Wang
Crystals 2024, 14(11), 990; https://doi.org/10.3390/cryst14110990 - 16 Nov 2024
Viewed by 824
Abstract
The demand for reconfigurable devices for emerging RF and microwave applications has been growing in recent years, with additive manufacturing and photonic thermal treatment presenting new possibilities to supplement conventional fabrication processes to meet this demand. In this paper, we present the realization [...] Read more.
The demand for reconfigurable devices for emerging RF and microwave applications has been growing in recent years, with additive manufacturing and photonic thermal treatment presenting new possibilities to supplement conventional fabrication processes to meet this demand. In this paper, we present the realization and analysis of barium–strontium–titanate-(Ba0.5Sr0.5TiO3)-based ferroelectric variable capacitors (varactors), which are additively deposited on top of conventionally fabricated interdigitated capacitors and enhanced by photonic thermal processing. The ferroelectric solution with suspended BST nanoparticles is deposited on the device using an ambient spray pyrolysis method and is sintered at low temperatures using photonic thermal processing by leveraging the high surface-to-volume ratio of the BST nanoparticles. The deposited film is qualitatively characterized using SEM imaging and XRD measurements, while the varactor devices are quantitatively characterized by using high-frequency RF measurements from 300 MHz to 10 GHz under an applied DC bias voltage ranging from 0 V to 50 V. We observe a maximum tunability of 60.6% at 1 GHz under an applied electric field of 25 kV/mm (25 V/μm). These results show promise for the implementation of photonic thermal processing and additive manufacturing as a means to integrate reconfigurable ferroelectric varactors in flexible electronics or tightly packaged on-chip applications, where a limited thermal budget hinders the conventional thermal processing. Full article
(This article belongs to the Special Issue Ceramics: Processes, Microstructures, and Properties)
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Figure 1

Figure 1
<p>(<b>a</b>) Diagram of IDC device fabricated with key dimensions labeled; (<b>b</b>) diagram depicting how BST is deposited to only cover the finger area; (<b>c</b>) microscope image of fabricated electrode layer; (<b>d</b>) SEM image of a device after BST deposition.</p>
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<p>Diagram of ambient spray pyrolysis setup.</p>
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<p>(<b>a</b>) Magnitude of the reflection coefficient (<span class="html-italic">S</span><sub>11</sub>) of the BST IDC device under a varied bias voltage (0–50 V), and (<b>b</b>) phase of the reflection coefficient of the BST IDC device under a varied bias voltage (0–50 V).</p>
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<p>(<b>a</b>) Real part of the calculated <span class="html-italic">Z</span><sub>11</sub> (impedance) for different bias conditions, (<b>b</b>) imaginary part of the calculated <span class="html-italic">Z</span><sub>11</sub> (reactance) for different bias conditions, (<b>c</b>) calculated capacitance for the IDC device under different bias conditions, and (<b>d</b>) zoomed-in region of the calculated capacitance versus the frequency below 3 GHz.</p>
Full article ">Figure 4 Cont.
<p>(<b>a</b>) Real part of the calculated <span class="html-italic">Z</span><sub>11</sub> (impedance) for different bias conditions, (<b>b</b>) imaginary part of the calculated <span class="html-italic">Z</span><sub>11</sub> (reactance) for different bias conditions, (<b>c</b>) calculated capacitance for the IDC device under different bias conditions, and (<b>d</b>) zoomed-in region of the calculated capacitance versus the frequency below 3 GHz.</p>
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<p>(<b>a</b>) Tunability vs. frequency of a measured IDC device at various bias voltages, and (<b>b</b>) zoomed-in tunability vs. frequency for a reduced frequency range up to 3 GHz, over which the tunability exhibits quasi-constant values.</p>
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<p>(<b>a</b>) Graph of the derived dielectric constant, and (<b>b</b>) zoomed-in dielectric constant graph.</p>
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<p>Calculated <span class="html-italic">Q</span> Factor of the IDC device as a function of frequency under various bias voltages.</p>
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<p>(<b>a</b>) Calculated loss tangent vs. frequency at a varied bias voltage (0–50 V), and (<b>b</b>) zoomed-in loss tangent vs. frequency at a reduced frequency range up to 3 GHz under a varied bias voltage (0–50 V).</p>
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<p>SEM images of the BST film deposited on the IDC device at different magnifications: (<b>a</b>) 240× magnification, (<b>b</b>) 1000× magnification, (<b>c</b>) 4000× magnification, and (<b>d</b>) 8000× magnification.</p>
Full article ">Figure 9 Cont.
<p>SEM images of the BST film deposited on the IDC device at different magnifications: (<b>a</b>) 240× magnification, (<b>b</b>) 1000× magnification, (<b>c</b>) 4000× magnification, and (<b>d</b>) 8000× magnification.</p>
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<p>(<b>a</b>) Full span two-theta XRD scan of the BST film after the photonic thermal processing (unlabeled peaks correspond to Kβ), and (<b>b</b>) a comparison of XRD responses of as-deposited and thermally processed BST films.</p>
Full article ">Figure 10 Cont.
<p>(<b>a</b>) Full span two-theta XRD scan of the BST film after the photonic thermal processing (unlabeled peaks correspond to Kβ), and (<b>b</b>) a comparison of XRD responses of as-deposited and thermally processed BST films.</p>
Full article ">
27 pages, 5970 KiB  
Article
Machine Learning-Aided Dual-Function Microfluidic SIW Sensor Antenna for Frost and Wildfire Detection Applications
by Amjaad T. Altakhaineh, Rula Alrawashdeh and Jiafeng Zhou
Energies 2024, 17(20), 5208; https://doi.org/10.3390/en17205208 - 19 Oct 2024
Cited by 1 | Viewed by 1216
Abstract
In this paper, which represents a fundamental step in ongoing research, a new smart low-energy dual-function half-mode substrate integrated waveguide cavity-interdigital capacitor (HMSIWC-DIC) antenna-based sensor is developed and investigated for remote frost and wildfire detection applications at 5.7 GHz. The proposed methodology exploits [...] Read more.
In this paper, which represents a fundamental step in ongoing research, a new smart low-energy dual-function half-mode substrate integrated waveguide cavity-interdigital capacitor (HMSIWC-DIC) antenna-based sensor is developed and investigated for remote frost and wildfire detection applications at 5.7 GHz. The proposed methodology exploits the HMSIW antenna-based sensor, a microfluidic channel (microliter water channel (50 μL)), interdigital capacitor technologies, and the resonance frequency parameters combined with machine learning algorithms. This allows for superior interaction between the water channel and the TE101 mode, resulting in high sensitivity (∆f/∆ε = 5.5 MHz/ε (F/m) and ∆f/∆°C = 1.83 MHz/°C) within the sensing range. Additionally, it exhibits high decision-making ability and immunity to interference, demonstrating a best-in-class sensory response to weather temperature across two ranges: positive (≥0 °C, including frost and wildfire) and negative (<0 °C, including ice accumulation). To address the challenges posed by the non-linear, unpredictable behavior of resonance frequency results, even when dealing with weak sensor antenna responses, an innovative sensory intelligent system was proposed. This system utilizes resonance frequency results as features to classify and predict weather temperature ranges into three environmental states: Early Frost, Normal, and Early Wildfire, achieving an accuracy of 96.4%. Several machine learning techniques are employed, including artificial neural networks (ANNs), random forests (RF), decision trees (DT), support vector machines (SVMs), and Gaussian processes (GPs). This sensor serves as an ideal solution for energy management through its utilization in RF-based weather temperature sensing applications. It boasts stable performance, minimal energy consumption, and real-time sensitivity, eliminating the necessity for manual data recording. Full article
(This article belongs to the Section F: Electrical Engineering)
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Figure 1
<p>Configuration of the proposed HMSIW sensor: (<b>a</b>) top view, (<b>b</b>) layout view. Filling hole diameter r = 3 mm, channel width c = 5 mm, SIW diminutions (L = 17.7 mm, W = 20 mm), (vias diameter d = 0.77 mm, vias space S = 1.52), interdigital capacitor dimensions (n = 0.7 mm, m = 2 mm).</p>
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<p>Top view of the proposed FMSIW cavity, (<b>a</b>) FMSIW, (<b>b</b>) HMSIW (REF1). The A–a and B–b lines mark the cutting edges of the magnetic walls.</p>
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<p>Simulated electric field distributions of the fundamental mode TE101 at 6.2 GHz over (<b>a</b>) FMSIW cavity and (<b>b</b>) HMSIW cavity. The electric field distributions are minimal near the via walls (blue color) and strongly concentrated within the SIW cavity (red color).</p>
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<p>Antenna structure evolution steps.</p>
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<p>Fundamental equivalent circuit model of unit-section.</p>
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<p>Simulated reflection coefficient of the REF 1 and REF 2 antennas, showing the electric field magnitude. The electric field distribution is represented using an intensity scale (heat map). The upper and lower rows indicate the minimum (blue) and maximum (red) magnitudes of the electric fields.</p>
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<p>Simulated reflection coefficient of the loaded and unloaded antennas (REF 2 and REF 3).</p>
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<p>Effective permittivity as a function of water temperature.</p>
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<p>Simulated S<sub>11</sub> of the sensor: (<b>a</b>) as a function of water temperature for different temperatures, (<b>b</b>) presence of ice, frost, and water.</p>
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<p>The simulated far-field parameters of the HMSIW (unloaded): (<b>a</b>) total efficiency, (<b>b</b>) peak gain at 5.7 GHz.</p>
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<p>Simulated far-field parameters of the antenna sensor for the sensing range: (<b>a</b>) total efficiency, (<b>b</b>) peak gain at 5.72 GHz.</p>
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<p>Simulated normalized radiation patterns of sensor antenna at 5.72 GHz: (<b>a</b>) XY plane, (<b>b</b>) XZ plane.</p>
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<p>The proposed ANN architecture. FC (resonance frequency), and the lower and upper frequencies (FL, Fu), hidden layers (HL), numbers of units (256, 128, 64, 16, etc.), and Early Frost (EF), Normal (N), and Early Wildfire (EW).</p>
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<p>ANN Model accuracy for training and validation sets.</p>
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13 pages, 3334 KiB  
Article
Gelatin-Coated High-Sensitivity Microwave Sensor for Humidity-Sensing Applications
by Junho Yeo and Younghwan Kwon
Sensors 2024, 24(19), 6286; https://doi.org/10.3390/s24196286 - 28 Sep 2024
Cited by 1 | Viewed by 793
Abstract
In this paper, the humidity-sensing characteristics of gelatin were compared with those of poly(vinyl alcohol) (PVA) at L-band (1 ~ 2 GHz) microwave frequencies. A capacitive microwave sensor based on a defected ground structure with a modified interdigital capacitor (DGS-MIDC) in a microstrip [...] Read more.
In this paper, the humidity-sensing characteristics of gelatin were compared with those of poly(vinyl alcohol) (PVA) at L-band (1 ~ 2 GHz) microwave frequencies. A capacitive microwave sensor based on a defected ground structure with a modified interdigital capacitor (DGS-MIDC) in a microstrip transmission line operating at 1.5 GHz without any coating was used. Gelatin is a natural polymer based on protein sourced from animal collagen, whereas PVA is a high-sensitivity hydrophilic polymer that is widely used for humidity sensors and has a good film-forming property. Two DGS-MIDC-based microwave sensors coated with type A gelatin and PVA, respectively, with a thickness of 0.02 mm were fabricated. The percent relative frequency shift (PRFS) and percent relative magnitude shift (PRMS) based on the changes in the resonant frequency and magnitude level of the transmission coefficient for the microwave sensor were used to compare the humidity-sensing characteristics. The relative humidity (RH) was varied from 50% to 80% with a step of 10% at a fixed temperature of around 25 °C using a low-reflective temperature and humidity chamber manufactured with Styrofoam. The experiment’s results show that the capacitive humidity sensitivity of the gelatin-coated microwave sensor in terms of the PRFS and PRMS was higher compared to that of the PVA-coated one. In particular, the sensitivity of the gelatin-coated microwave sensor at a low RH from 50% to 60% was much greater compared to that of the PVA-coated one. In addition, the relative permittivity of the fabricated microwave sensors coated with PVA and gelatin was extracted by using the measured PRFS and the equation was derived by curve-fitting the simulated results. The change in the extracted relative permittivity for the gelatin-coated microwave sensor was larger than that of the PVA-coated one for varying the RH. Full article
(This article belongs to the Special Issue RF and IoT Sensors: Design, Optimization and Applications)
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Figure 1
<p>Chemical structures of (<b>a</b>) PVA and (<b>b</b>) gelatin.</p>
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<p>DGS-MIDC-based microwave sensor: (<b>a</b>) geometry, (<b>b</b>) electric-field distribution at 1.5 GHz, and (<b>c</b>) S-parameter characteristics and simplified equivalent circuit model.</p>
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<p>Performance characteristics of the DGS-MIDC-based microwave sensor for varying relative permittivity of the coated polymer with tan <span class="html-italic">δ</span> = 0: (<b>a</b>) S<sub>21</sub>, (<b>b</b>) <span class="html-italic">f</span><sub>r</sub>, and (<b>c</b>) PRFS.</p>
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<p>Extracted equivalent circuit parameters of the DGS-MIDC-based microwave sensor for varying relative permittivity of the coated polymer with tan <span class="html-italic">δ</span> = 0: (<b>a</b>) <span class="html-italic">C</span><sub>1</sub>; (<b>b</b>) <span class="html-italic">L</span><sub>1</sub>; (<b>c</b>) Δ<span class="html-italic">C</span><sub>1</sub>/<span class="html-italic">C</span><sub>1</sub>(%) and (<b>d</b>) Δ<span class="html-italic">L</span><sub>1</sub>/<span class="html-italic">L</span><sub>1</sub>(%).</p>
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<p>Photographs of the fabricated microwave sensors coated with (<b>a</b>) PVA and (<b>b</b>) gelatin.</p>
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<p>Block diagram and photographs of the experiment setup for the humidity-sensing measurements: (<b>a</b>) block diagram, (<b>b</b>) experiment setup with an open-top cover, and (<b>c</b>) experiment setup with closed-top cover.</p>
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<p>Measured S<sub>21</sub> characteristics of the fabricated microwave sensors coated with the polymers for varying RH. (<b>a</b>) PVA and (<b>b</b>) gelatin.</p>
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<p>Performance comparison of the fabricated microwave sensors coated with the polymers for varying RH. (<b>a</b>) <span class="html-italic">f</span><sub>r</sub>, (<b>b</b>) PRFS, (<b>c</b>) S<sub>21</sub> magnitude, and (<b>d</b>) PRMS.</p>
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<p>Comparison of the extracted relative permittivity from measured PRFSs of PVA- and gelatin-coated microwave sensors for varying RH.</p>
Full article ">
16 pages, 6756 KiB  
Article
Aging Characterization of Modified Insulating Paper Based on the Transmission Characteristics of Microstrip Resonant Sensors
by Mi Xiao, Gaoyan Yang and Wei Zhang
Energies 2024, 17(11), 2499; https://doi.org/10.3390/en17112499 - 23 May 2024
Viewed by 756
Abstract
In this paper, the aging characterization of a kind of insulating paper modified by magnetron sputtering MgO particles based on a microstrip resonant sensor was presented. Firstly, the modified insulating paper with 0, 15 and 30 min MgO particle sputtering times was prepared [...] Read more.
In this paper, the aging characterization of a kind of insulating paper modified by magnetron sputtering MgO particles based on a microstrip resonant sensor was presented. Firstly, the modified insulating paper with 0, 15 and 30 min MgO particle sputtering times was prepared by a magnetron sputtering device. After that, the properties of the modified insulating paper with different sputtering times were analyzed through microscopic characterization, infrared spectrum, polymerization degree, dielectric constant, AC breakdown strength and thermal aging experiments. The results show that the dielectric constant of the modified insulating paper decreased obviously, the AC breakdown strength increased and the thermal aging resistance was better after 15 min of sputtering. The overall performance of the modified insulating paper after 30 min of sputtering is reduced due to excessive sputtering. In addition, microstrip resonant sensors are introduced to characterize the thermal aging degree of the modified insulating paper, and two microstrip resonant sensors are prepared: a complementary split ring resonator (CSRR) and an interdigital-capacitor-shaped defected ground structure resonator (IDCS-DGS). The resonance frequency deviation of the modified insulating paper samples after aging was measured by microstrip resonance sensors to show the influence of aging temperature on aging degree. The experimental results show that the test results of the microstrip resonance sensors are in good agreement with the traditional characterization methods and can characterize the various aging stages of the modified insulating paper to a certain extent, which proves the feasibility of the characterization method. Full article
(This article belongs to the Section F6: High Voltage)
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Figure 1

Figure 1
<p>Magnetron sputtering equipment.</p>
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<p>RF magnetron sputtering schematic.</p>
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<p>Thermal aging experiment arrangement.</p>
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<p>Microstrip resonant sensors and test environment. (<b>a</b>) CSRR sensor and its CSRR-etched structure; (<b>b</b>) CSRR sensor and its CSRR-etched structure; (<b>c</b>) Test fixture; (<b>d</b>) Test environment.</p>
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<p>Microstrip sensor transmission characteristics curve. (<b>a</b>) CSRR sensor; (<b>b</b>) IDCS-DGS sensor.</p>
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<p>Modified insulating paper FESEM and EDS. (<b>a</b>) Sputter-modified insulating paper for 0 min; (<b>b</b>) Sputter modified insulating paper for 15 min; (<b>c</b>) Sputter-modified insulating paper for 30 min.</p>
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<p>ATR-IR spectra of modified insulating paper with different sputtering times.</p>
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<p>FESEM of modified insulating paper aged 90 days. (<b>a</b>) Sputter-modified insulating paper for 0 min; (<b>b</b>) Sputter-modified insulating paper for 15 min; (<b>c</b>) Sputter-modified insulating paper for 30 min.</p>
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<p>Polymerization degree of modified insulating paper at different aging stages.</p>
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<p>Dielectric constants of modified insulating paper at different aging stages. (<b>a</b>) Sputter-modified insulating paper for 0 min; (<b>b</b>) Sputter-modified insulating paper for 15 min; (<b>c</b>) Sputter-modified insulating paper for 30 min.</p>
Full article ">Figure 11
<p>Ac breakdown strength of modified insulating paper at different aging stages under Weibull distribution (<b>a</b>) Sputter-modified insulating paper for 0 min; (<b>b</b>) Sputter-modified insulating paper for 15 min; (<b>c</b>) Sputter-modified insulating paper for 30 min.</p>
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<p>Transmission characteristics of CSRR loaded with samples before and after aging.</p>
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<p>Resonant frequency deviation. (<b>a</b>) CSRR; (<b>b</b>) IDCS-DGS.</p>
Full article ">
23 pages, 10230 KiB  
Article
Compact and Hybrid Dual-Band Bandpass Filter Using Folded Multimode Resonators and Second-Mode Suppression
by Nicolas Claus, Kamil Yavuz Kapusuz, Jo Verhaevert and Hendrik Rogier
Electronics 2024, 13(10), 1921; https://doi.org/10.3390/electronics13101921 - 14 May 2024
Cited by 2 | Viewed by 1312
Abstract
The proliferation of the Internet of Things (IoT) propels the continuous demand for compact, low-cost, and high-performance multiband filters. This paper introduces a novel low-profile dual-band bandpass filter (BPF) constructed with a back-to-back coupled pair of shielded folded quarter-mode substrate integrated waveguide (SF-QMSIW) [...] Read more.
The proliferation of the Internet of Things (IoT) propels the continuous demand for compact, low-cost, and high-performance multiband filters. This paper introduces a novel low-profile dual-band bandpass filter (BPF) constructed with a back-to-back coupled pair of shielded folded quarter-mode substrate integrated waveguide (SF-QMSIW) multimode cavities. A hybrid structure is obtained by etching a coplanar waveguide (CPW) coupling line in the folded cavity’s septum layer. It serves multiple functions: generating an additional resonance, providing a separate coupling mechanism for the upper passband, and offering the flexibility to control the passbands’ center frequency ratio. Additionally, the unused second higher-order mode is suppressed by integrating embedded split-ring resonators (ESRRs) with an inter-digital capacitor (IDC) structure into the feed lines. A filter prototype has been fabricated and experimentally tested. The measurements confirmed reliable operation in two passbands having center frequencies 3.6 GHz and 5.8 GHz, and exhibiting 3 dB fractional bandwidths (FBWs) of 6.4% and 5.3%, respectively. Furthermore, the group delay variation within both passbands equals only 0.62 ns and 1.00 ns, respectively. Owing to the second higher-order mode suppression, the filter demonstrated an inter-band rejection exceeding 38 dB, within a compact footprint of 0.71λg2 (with λg being the guided wavelength at the lower passband’s center frequency). Full article
(This article belongs to the Section Microwave and Wireless Communications)
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Figure 1
<p>Evolution towards the SF-QMSIW: (<b>a</b>) square cavity with the electric sidewalls synthesized by rows of vias with diameter <span class="html-italic">d</span> and spacing <span class="html-italic">s</span>, (<b>b</b>) conventional QMSIW cavity, (<b>c</b>) F-QMSIW cavity, and (<b>d</b>) SF-QMSIW cavity with added guard trace and guard gap. The top and bottom figures represent the top view and cross-sectional side view along <math display="inline"><semantics> <mrow> <mi>A</mi> <msup> <mi>A</mi> <mo>′</mo> </msup> </mrow> </semantics></math>, respectively. For the folded structures, the septum gap in the septum layer is indicated by a white hatched pattern.</p>
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<p>Overview of the simulated electric (E-field) and magnetic (H-field) mode profiles for the first three resonant modes in an unfolded QMSIW (<a href="#electronics-13-01921-f001" class="html-fig">Figure 1</a>b) and an SF-QMSIW cavity (<a href="#electronics-13-01921-f001" class="html-fig">Figure 1</a>d). A red color represents the maximal field strength. A deep blue color indicates the minimal value. For the SF-QMSIW, the septum gap is indicated by a white hatched pattern. The inductive window’s initial center location in the design process is situated at a distance <math display="inline"><semantics> <msub> <mi>x</mi> <mi>w</mi> </msub> </semantics></math> from the cavity’s back electric wall. This is further explained in <a href="#sec3dot3dot1-electronics-13-01921" class="html-sec">Section 3.3.1</a>.</p>
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<p>Simulated electric field vectors within the cross-section along <math display="inline"><semantics> <mrow> <mi>B</mi> <msup> <mi>B</mi> <mo>′</mo> </msup> </mrow> </semantics></math> in <a href="#electronics-13-01921-f002" class="html-fig">Figure 2</a>, with (<b>a</b>) Mode 1, (<b>b</b>) Mode 2, and (<b>c</b>) Mode 3. Large red arrows represent maximal field strengths. Small blue arrows indicate minimal field strengths. The region of the septum gap is indicated by a dashed rectangle.</p>
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<p>Conceptual evolution from two SF-QMSIW building blocks (Cavity 1 and Cavity 2) to the proposed dual-band filter topology, where both cavities are positioned back-to-back. Cavity 2 is an identical copy of Cavity 1, but mirrored along the indicated mirror line. They are separated by a row of metallic vias with diameter <math display="inline"><semantics> <mi>δ</mi> </semantics></math> and spacing <math display="inline"><semantics> <mi>σ</mi> </semantics></math>. These vias constitute the shared folding wall for both cavities. Similar to <a href="#electronics-13-01921-f001" class="html-fig">Figure 1</a>, the septum gaps are indicated by white hatched patterns.</p>
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<p>Exploded view of the proposed dual-band bandpass filter, with semitransparent metal layers (indicated by M1, M2, M3, and M4). For better visualization, not all brass nuts and bolts are shown. In addition, the grounding vias are made invisible. Inter-resonator coupling is obtained through the circled inductive coupling window and the CPW coupling trace. The former is exploited to control the coupling for the lower passband, whereas the latter is applied to control the coupling for the upper passband.</p>
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<p>Topology of the proposed dual-band bandpass filter. It shows the metal layers of <a href="#electronics-13-01921-f005" class="html-fig">Figure 5</a> and some detailed insets: (<b>a</b>) M1, (<b>b</b>) M2, (<b>c</b>) M3, and (<b>d</b>) M4. (<b>e</b>) Detail A is the feed entering the cavity. (<b>f</b>) Detail B is the inductive coupling window with the CPW coupling trace. (<b>g</b>) Detail C is the section of ESRRs in the feed lines. All layers are depicted from the top perspective, and the grounding vias are displayed.</p>
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<p>Flowchart illustrating the design process. The depicted steps are further explained below.</p>
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<p>The starting topology for the design process: (<b>a</b>) geometry, with the feeds on opposite metal layers (i.e., Feed 1 on M1 and Feed 2 on M4, or vice versa), and (<b>b</b>) its simulated transmission characteristic <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (in dB). As indicated by the dashed lines, the feed offset is initially zero (<math display="inline"><semantics> <mrow> <msub> <mi>o</mi> <mi>f</mi> </msub> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math>). The inductive coupling window with width <math display="inline"><semantics> <msub> <mi>w</mi> <mi>w</mi> </msub> </semantics></math> is represented by a hatched pattern. Its center is initially located at the distance <math display="inline"><semantics> <msub> <mi>x</mi> <mi>w</mi> </msub> </semantics></math> from the cavity’s back electric wall (<a href="#electronics-13-01921-f002" class="html-fig">Figure 2</a>), i.e., <math display="inline"><semantics> <mrow> <msub> <mi>c</mi> <mi>w</mi> </msub> <mo>=</mo> <msup> <mi>a</mi> <mo>′</mo> </msup> <mo>/</mo> <mn>4</mn> <mo>=</mo> <mi>a</mi> <mo>/</mo> <mrow> <mo>(</mo> <mn>4</mn> <msqrt> <mn>2</mn> </msqrt> <mo>)</mo> </mrow> </mrow> </semantics></math>.</p>
Full article ">Figure 9
<p>Simulated effect of the feed offset <math display="inline"><semantics> <msub> <mi>o</mi> <mi>f</mi> </msub> </semantics></math> on (<b>a</b>) the transmission characteristic <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (in dB) of the starting topology in <a href="#electronics-13-01921-f008" class="html-fig">Figure 8</a>a, and (<b>b</b>) the external quality factor for each mode.</p>
Full article ">Figure 10
<p>Simulated effect of the window width <math display="inline"><semantics> <msub> <mi>w</mi> <mi>w</mi> </msub> </semantics></math> on (<b>a</b>) the reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (dot-dashed lines) and transmission coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (solid lines) in dB, and (<b>b</b>) the coupling coefficient for Mode 1.</p>
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<p>Simulated reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (dot-dashed lines) and transmission coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (solid lines), in dB, for variations of the coupling trace parameters: (<b>a</b>) variation of the coupling trace length <math display="inline"><semantics> <msub> <mi>l</mi> <mi>t</mi> </msub> </semantics></math>, (<b>b</b>) variation of the window’s center location <math display="inline"><semantics> <msub> <mi>c</mi> <mi>w</mi> </msub> </semantics></math>, (<b>c</b>) variation of the coupling trace’s slot width <math display="inline"><semantics> <msub> <mi>w</mi> <mi>t</mi> </msub> </semantics></math> in the frequency range from 2 GHz to 7 GHz, (<b>d</b>) zoomed in to the frequency range from <math display="inline"><semantics> <mrow> <mn>5.5</mn> </mrow> </semantics></math> GHz to <math display="inline"><semantics> <mrow> <mn>6.5</mn> </mrow> </semantics></math> GHz, (<b>e</b>) variation of the coupling trace separation <math display="inline"><semantics> <msub> <mi>s</mi> <mi>t</mi> </msub> </semantics></math> in the frequency range from 2 GHz to 7 GHz, and (<b>f</b>) zoomed in to the frequency range from <math display="inline"><semantics> <mrow> <mn>5.5</mn> </mrow> </semantics></math> GHz to <math display="inline"><semantics> <mrow> <mn>6.5</mn> </mrow> </semantics></math> GHz.</p>
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<p>Simulated outcomes for the intermediate result in the design process (Filter I in <a href="#electronics-13-01921-f007" class="html-fig">Figure 7</a>): (<b>a</b>) reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (solid line) and transmission coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (dot-dashed line), in dB. E-fields at the indicated frequency points, i.e., at (<b>b</b>) frequency A, (<b>c</b>) frequency B, (<b>d</b>) frequency C, and (<b>e</b>) frequency D. H-fields at the indicated frequency points, i.e., at (<b>f</b>) frequency A, (<b>g</b>) frequency B, (<b>h</b>) frequency C, and (<b>i</b>) frequency D. A red color represents the maximal field strength. A deep blue color indicates the minimal value.</p>
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<p>Simulated reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (solid line) and transmission coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (dot-dashed line), in dB, for (<b>a</b>) the filter without ESRRs, (<b>b</b>) the stand-alone ESRR section, and (<b>c</b>) the filter with ESRRs (Filter II in <a href="#electronics-13-01921-f007" class="html-fig">Figure 7</a>). The gray area indicates the region outside of which the reflection coefficient of the ESRR section is below −10 dB. The blue areas denote the lower and upper passbands, which are nearly unaffected by the application of the ESRRs. In this simulation, the filter is assumed to be lossless, and rows of vias are replaced by solid sheets of PEC.</p>
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<p>Photographs of the assembled prototype: (<b>a</b>) top view, (<b>b</b>) bottom view, and (<b>c</b>) the prototype connected to the VNA.</p>
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<p>Simulated (solid lines) and measured (dashed lines) reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (red lines) and transmission coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>21</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (blue lines), in dB. The relevant frequency bands, namely, 5G NR n48 (3.55–3.70 GHz), U-NII-3 (5.725–5.850 GHz), and U-NII-4 (5.850–5.925 GHz), are indicated by blue, orange, and yellow rectangles, respectively. In this simulation, all losses are included as well as the actual rows of vias.</p>
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<p>Simulated loss contributions to the transmission characteristic of Filter III, for (<b>a</b>) the lower passband, and (<b>b</b>) the upper passband. The borders of both passbands are indicated by dashed lines.</p>
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<p>Simulated (solid lines) and measured (dot-dashed lines) group delays (in ns), for (<b>a</b>) the lower passband, and (<b>b</b>) the upper passband. The relevant frequency bands, namely, 5G NR n48 (3.55–3.70 GHz), U-NII-3 (5.725–5.850 GHz), and U-NII-4 (5.850–5.925 GHz), are indicated by blue, orange, and yellow rectangles, respectively.</p>
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15 pages, 4879 KiB  
Article
Real-Time Detection of Yeast Growth on Solid Medium through Passive Microresonator Biosensor
by Bo-Wen Shi, Jun-Ming Zhao, Yi-Ke Wang, Yan-Xiong Wang, Yan-Feng Jiang, Gang-Long Yang, Jicheng Wang and Tian Qiang
Biosensors 2024, 14(5), 216; https://doi.org/10.3390/bios14050216 - 26 Apr 2024
Viewed by 1902
Abstract
This study presents a biosensor fabricated based on integrated passive device (IPD) technology to measure microbial growth on solid media in real-time. Yeast (Pichia pastoris, strain GS115) is used as a model organism to demonstrate biosensor performance. The biosensor comprises an [...] Read more.
This study presents a biosensor fabricated based on integrated passive device (IPD) technology to measure microbial growth on solid media in real-time. Yeast (Pichia pastoris, strain GS115) is used as a model organism to demonstrate biosensor performance. The biosensor comprises an interdigital capacitor in the center with a helical inductive structure surrounding it. Additionally, 12 air bridges are added to the capacitor to increase the strength of the electric field radiated by the biosensor at the same height. Feasibility is verified by using a capacitive biosensor, and the change in capacitance values during the capacitance detection process with the growth of yeast indicates that the growth of yeast can induce changes in electrical parameters. The proposed IPD-based biosensor is used to measure yeast drop-added on a 3 mm medium for 100 h at an operating frequency of 1.84 GHz. The resonant amplitude of the biosensor varies continuously from 24 to 72 h due to the change in colony height during vertical growth of the yeast, with a maximum change of 0.21 dB. The overall measurement results also fit well with the Gompertz curve. The change in resonant amplitude between 24 and 72 h is then analyzed and reveals a linear relationship with time with a coefficient of determination of 0.9844, indicating that the biosensor is suitable for monitoring yeast growth. Thus, the proposed biosensor is proved to have potential in the field of microbial proliferation detection. Full article
(This article belongs to the Section Biosensor and Bioelectronic Devices)
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<p>3D structure of the proposed biosensor with its equivalent circuit model and structure of air bridge section.</p>
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<p>Performance of different design of biosensors: (<b>a</b>) different electric field density of (<b>a.i</b>) Design 1, (<b>a.ii</b>) Design 2, (<b>a.iii</b>) Design 3, and (<b>a.iv</b>) Design 4; (<b>b</b>) different electric field density of (<b>b.i</b>) Design 1, (<b>b.ii</b>) Design 2, (<b>b.iii</b>) Design 3, and (<b>b.iv</b>) Design 4 at the height of 0.4 mm; and (<b>c</b>) the variation of electric field density with height for Design 1 to Design 4.</p>
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<p>Proposed biosensors: (<b>a</b>) capacitive biosensor and (<b>b</b>) microwave biosensor.</p>
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<p>Experimental environment for pre-experiment and used capacitive biosensor: (<b>a</b>) experimental environment setup; (<b>b</b>) bonded petri dish of capacitive biosensor; and (<b>c</b>) structure and equivalent circuits of capacitive biosensors.</p>
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<p>Record of growth of yeast: Growth situation of yeast on (<b>a</b>) biosensor and (<b>b</b>) petri dish with scale bar. (<b>c</b>) Measured ΔCp for 1 µL of yeast growth and the fitted Gompertz model curve at constant time intervals. (<b>d</b>) ΔCp measurements of yeasts every 4 h during the growth period with linear fit results with error bar. Note: error bars generated by fitting multiple measurement data using standard deviation (SD &lt; 3.8%).</p>
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<p>Experimental environment and used biosensor: (<b>a</b>) experimental environment setup; (<b>b</b>) microscope photo of IPD-based biosensor; (<b>c</b>) top view and (<b>d</b>) side view of the final test fixture; and (<b>e</b>) measured and simulated S11 parameters of biosensor.</p>
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<p>Weight of the biosensor at different stages: (<b>a</b>) weight of capacitive biosensor (<b>a.i</b>) before adding medium, (<b>a.ii</b>) after adding medium, and (<b>a.iii</b>) after 100 h of yeast growth; (<b>b</b>) weight of IPD-based biosensor (<b>b.i</b>) before adding medium, (<b>b.ii</b>) after adding medium, and (<b>b.iii</b>) after 100 h of yeast growth; and (<b>c</b>) weight changes and percentage of weight loss summarized in a bar chart.</p>
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<p>Record of experimental data and growth of yeast: (<b>a</b>) growth situation of yeast on biosensor with scale bar; (<b>b</b>) measured ΔAmplitude for 1 µL of yeast growth and the fitted Gompertz model curve at constant time intervals; and (<b>c</b>) measured ΔAmplitude of yeast every 4 h during the growth period with linear fit results with error bar. Note: error bars generated by fitting multiple measurement data using standard deviation (SD &lt; 3.8%).</p>
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<p>Mechanism diagrams for real-time monitoring of yeast growth: (<b>a</b>) 3D view of yeast growth on the biosensor; side view of yeast (<b>b</b>) before and (<b>c</b>) after growth; and (<b>d</b>) S11 parameters for yeast colony heights of 0.05, 0.10, 0.15, 0.20 mm, and without yeast colony on medium in simulation.</p>
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37 pages, 3374 KiB  
Review
The Role of Interdigitated Electrodes in Printed and Flexible Electronics
by Shayma Habboush, Sara Rojas, Noel Rodríguez and Almudena Rivadeneyra
Sensors 2024, 24(9), 2717; https://doi.org/10.3390/s24092717 - 24 Apr 2024
Cited by 3 | Viewed by 3655
Abstract
Flexible electronics, also referred to as printable electronics, represent an interesting technology for implementing electronic circuits via depositing electronic devices onto flexible substrates, boosting their possible applications. Among all flexible electronics, interdigitated electrodes (IDEs) are currently being used for different sensor applications since [...] Read more.
Flexible electronics, also referred to as printable electronics, represent an interesting technology for implementing electronic circuits via depositing electronic devices onto flexible substrates, boosting their possible applications. Among all flexible electronics, interdigitated electrodes (IDEs) are currently being used for different sensor applications since they offer significant benefits beyond their functionality as capacitors, like the generation of high output voltage, fewer fabrication steps, convenience of application of sensitive coatings, material imaging capability and a potential of spectroscopy measurements via electrical excitation frequency variation. This review examines the role of IDEs in printed and flexible electronics since they are progressively being incorporated into a myriad of applications, envisaging that the growth pattern will continue in the next generations of flexible circuits to come. Full article
(This article belongs to the Section Electronic Sensors)
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<p>Configuration of IDE structure wherein (<b>a</b>) geometric parameters of interdigitated sensors; (<b>b</b>) electric current displacement between electrodes; and (<b>c</b>) electrical model of an interdigitated sensor and sample (an ionic solution) (Source: [<a href="#B3-sensors-24-02717" class="html-bibr">3</a>]).</p>
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<p>Technology schematics of the main (PE) techniques: (<b>a</b>) gravure printing (schemed modified from [<a href="#B31-sensors-24-02717" class="html-bibr">31</a>]; (<b>b</b>) screen printing; (<b>c</b>) inkjet printing; (<b>d</b>) 3D printing, (<b>e</b>) flexographic printing (schemed modified from [<a href="#B31-sensors-24-02717" class="html-bibr">31</a>] and (<b>f</b>) laser scribbling (reprint with permission from reference [<a href="#B32-sensors-24-02717" class="html-bibr">32</a>]).</p>
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<p>Technology schematics of the main (PE) techniques: (<b>a</b>) gravure printing (schemed modified from [<a href="#B31-sensors-24-02717" class="html-bibr">31</a>]; (<b>b</b>) screen printing; (<b>c</b>) inkjet printing; (<b>d</b>) 3D printing, (<b>e</b>) flexographic printing (schemed modified from [<a href="#B31-sensors-24-02717" class="html-bibr">31</a>] and (<b>f</b>) laser scribbling (reprint with permission from reference [<a href="#B32-sensors-24-02717" class="html-bibr">32</a>]).</p>
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<p>Frequency spectrum for acoustic and electromagnetic digital sensors. OPD: organic photo-diodes, SAW: surface acoustic wave, APM: acoustic plate mode, NDE: node detection emitter, AC: alternating current, FPW: flexural plate-wave, SONAR: sound navigation and ranging.</p>
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<p>Design of capacitive devices (1—oscillator, 2—trigger circuit, and 3—output switching device).</p>
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<p>Mapping to the unit disk.</p>
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<p>(<b>a</b>) Responses of the sensors to 1–10 ppm of NO<sub>2</sub> at room temperature measured under near-infrared light using ZnO/PbS nanocomposites with varying PbS loading. (<b>b</b>) Photoluminescence spectra of ZnO/PbS-2 and PbS excited at 831 nm. Reproduced with permission. (<b>c</b>) Reactions of ZnO/g-C<sub>3</sub>N<sub>4</sub> composites to 7 ppm NO<sub>2</sub> under varied light illumination wavelengths with varying g-C<sub>3</sub>N<sub>4</sub> content. (<b>d</b>) Dynamic resistance curves for ZnO/g-C<sub>3</sub>N<sub>4</sub>-10 weight percent to 1%#x2013;6 ppm NO<sub>2</sub> at room temperature under 460 nm light irradiation. Licensed reproduction (reprint with permission from reference [<a href="#B131-sensors-24-02717" class="html-bibr">131</a>]).</p>
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<p>Resistance vs. temperature at 55%RH for electrodes with a width of 150 μm and spacing of 200 μm was studied, showing the temperature-dependent behavior of the electrode material [<a href="#B97-sensors-24-02717" class="html-bibr">97</a>].</p>
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<p>Capacitance strain measurement for the capacitive strain gauge shown in <a href="#sensors-24-02717-f008" class="html-fig">Figure 8</a> undergoing a strain of 125%. The fit was calculated using the equation = + ϵ + C × c c [<a href="#B135-sensors-24-02717" class="html-bibr">135</a>].</p>
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3 pages, 942 KiB  
Abstract
A Phenylalanine Ammonia Lyase Capacitive Sensor for Phenylalanine Detection
by Bruno Andò, Salvatore Castorina, Ludovica Maugeri, Salvatore Petralia, Maria Anna Messina, Martino Ruggieri, Giovanni Neri, Angelo Ferlazzo, Emilio Sardini and Mauro Serpelloni
Proceedings 2024, 97(1), 51; https://doi.org/10.3390/proceedings2024097051 - 19 Mar 2024
Cited by 1 | Viewed by 810
Abstract
In this paper, an easy-to-use and fast biosensor for phenylalanine quantification in patients affected by phenylketonuria is investigated. The phenylalanine concentration was indirectly estimated through the ammonia released as a by-product of an enzymatic reaction, which was then detected by exploiting an yttria-stabilized [...] Read more.
In this paper, an easy-to-use and fast biosensor for phenylalanine quantification in patients affected by phenylketonuria is investigated. The phenylalanine concentration was indirectly estimated through the ammonia released as a by-product of an enzymatic reaction, which was then detected by exploiting an yttria-stabilized zirconia layer deposited over an interdigitated capacitive sensor. The latter was manufactured by rapid-prototyping technologies. A sensor limit of detection higher than 1.25 µM was estimated, along with an accuracy better than 18.31 µM. Full article
(This article belongs to the Proceedings of XXXV EUROSENSORS Conference)
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<p>The sensor: (<b>a</b>) top view (layout), (<b>b</b>) cross section and real view, (<b>c</b>) calibration diagram.</p>
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10 pages, 2360 KiB  
Article
Development of a Battery-Free, Chipless, and Highly Sensitive Radio Frequency Glucose Biosensor
by Md. Rajibur Rahaman Khan
Micromachines 2024, 15(2), 272; https://doi.org/10.3390/mi15020272 - 14 Feb 2024
Viewed by 1303
Abstract
In our study, we designed and developed a glucose biosensor that operates without a battery or chip. This biosensor utilizes the principles of radio frequency (RF) to operate. For the construction of a glucose-sensitive interdigitated capacitor (IDC), a famous glucose-sensitive substance called phenylboronic [...] Read more.
In our study, we designed and developed a glucose biosensor that operates without a battery or chip. This biosensor utilizes the principles of radio frequency (RF) to operate. For the construction of a glucose-sensitive interdigitated capacitor (IDC), a famous glucose-sensitive substance called phenylboronic acid (PBA) is combined with a polyvinyl chloride (PVC) and n,n-dimethylacetamide (DMAC) solution. According to the theory of radio frequency sensing, the resonance frequency shifts whenever there is a change in the capacitance of the glucose-sensitive IDC. This change is caused by the fluctuations in glucose concentrations. As far as we are aware, this is the first glucose sensor that employs the RF principle to detect changes in glucose solution concentrations using PBA as the principal glucose-sensitive material. The sensor can detect glucose levels with remarkable sensitivity, around 40.89 kHz/decade, and a broad dynamic range covering 10 μM to 1 M. Additionally, the designed biosensor has excellent linearity performance, with a value of around 0.988. The proposed glucose biosensor has several benefits: lightweight, inexpensive, easy to build, and an acceptable selectivity response. Our study concludes by comparing the proposed RF sensor’s effectiveness to that of existing glucose sensors, which it outperforms. Full article
(This article belongs to the Special Issue Recent Advances in Sensors and Sensing System Design)
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<p>(<b>a</b>) Schematic of the RF glucose sensor without a sensing membrane in the IDE; (<b>b</b>) schematic of the RF glucose biosensor with a sensing membrane in the IDE to form the IDC; and (<b>c</b>) the molecular structure of phenylboronic acid and its reaction mechanism with glucose.</p>
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<p>Photographs of the fabricated glucose biosensor. (<b>a</b>) A top view of the sensor before depositing the sensing membrane in the IDE; (<b>b</b>) a top view of the sensor after depositing the sensing membrane in the IDE to create the IDC glucose biosensor; and (<b>c</b>) a bottom view of the fabricated glucose biosensor.</p>
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<p>Schematic of the experimental setup to detect glucose.</p>
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<p>Performance of the sensor. (<b>a</b>) Frequency response of the sensor; and (<b>b</b>) enlarged view (indicated in yellow rectangular in (<b>a</b>)) of the frequency response of the sensor.</p>
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<p>Performance of the sensor. (<b>a</b>) Resonance frequency vs. different concentrations of glucose solutions; and (<b>b</b>) capacitance vs. glucose concentration.</p>
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<p>Performance of the sensor. (<b>a</b>) The relationship between capacitance and the resonance frequency under different concentrations of glucose; and (<b>b</b>) the selectivity response of the sensor under 1 mM of different solutions.</p>
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<p>(<b>a</b>) The repeatability performance of the proposed glucose biosensor, and (<b>b</b>) the reproducibility performance of the proposed glucose biosensor.</p>
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17 pages, 7785 KiB  
Article
Design for SAW Antenna-Plexers with Improved Matching Inductance Circuits
by Min-Yuan Yang and Ruey-Beei Wu
Micromachines 2024, 15(1), 89; https://doi.org/10.3390/mi15010089 - 30 Dec 2023
Cited by 1 | Viewed by 1900
Abstract
This study designs antenna-plexers, including a surface acoustic wave (SAW) extractor and an upper- and mid-high band (UHB + MHB) diplexer, for LTE 4G and 5G bands using carrier aggregation. The SAW extractor combines a bandpass filter (BPF) and a band-stop filter (BSF) [...] Read more.
This study designs antenna-plexers, including a surface acoustic wave (SAW) extractor and an upper- and mid-high band (UHB + MHB) diplexer, for LTE 4G and 5G bands using carrier aggregation. The SAW extractor combines a bandpass filter (BPF) and a band-stop filter (BSF) in a single unit that consists of eight modified Butterworth–van Dyke (mBVD) resonators that resonate in parallel with an inductor and SAW resonators. This BSF behaves as a high-pass filter at frequencies lower than the designed WIFI band and as a capacitor at higher frequencies. The SAW extractor meets product specifications in the frequency range 0.7 to 2.7 GHz. The UHB + MHB diplexer, which is composed of a microwave filter, a SAW filter, and a simple matching inductor, uses frequency response methods to create an RF component for 2.4 GHz + WIFI 6E applications. The design uses a SAW’s interdigital transducer (IDT) structure, and the experimental results are in agreement with the simulation results, so the design is feasible. Full article
(This article belongs to the Special Issue Novel Surface and Bulk Acoustic Wave Devices)
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<p>Various frequency bands that are used for handheld devices [<a href="#B1-micromachines-15-00089" class="html-bibr">1</a>]. The blue colors denote the bands used for cellular communications.</p>
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<p>SAW extractor topology. Black boxes with numerical insets represent SAW resonators. Resonators 1 to 5 are used for bandpass filter at WiFi and resonators 6 to 8 for band-stop filter.</p>
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<p>Equivalent mBVD model for a SAW resonator [<a href="#B18-micromachines-15-00089" class="html-bibr">18</a>].</p>
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<p>UHB + MHB diplexer topology. Black boxes with numerical insets represent SAW resonators. Resonators 1 to 7 are used for bandpass filter at mid-high band.</p>
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<p>Flowchart for the antenna-plexer design.</p>
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<p>Frequency responses for a SAW extractor using the mBVD model. (<b>a</b>) Insertion loss and (<b>b</b>) return loss and isolation.</p>
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<p>Frequency responses for a UHB + MHB diplexer using the mBVD model. (<b>a</b>) Insertion loss and (<b>b</b>) isolation.</p>
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<p>The effects of a parallel inductor on the frequency response for the band-stop filter. Black boxes represent SAW resonators and the numerical insets correspond to those in <a href="#micromachines-15-00089-f002" class="html-fig">Figure 2</a>.</p>
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<p>The Smith chart for the band-stop filter <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math> (<b>a</b>) without a series inductor and (<b>b</b>) with a series inductor.</p>
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<p>The frequency response for the band-stop filter if a series inductor is added for matching. Black boxes represent SAW resonators and the numerical insets correspond to those in <a href="#micromachines-15-00089-f002" class="html-fig">Figure 2</a>.</p>
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<p>The Smith chart for <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math> for (<b>a</b>) a 7th-order SAW filter and (<b>b</b>) a WIFI 6E filter.</p>
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<p>Structure of a single-port resonator.</p>
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<p>The relationship between pitch width (<span class="html-italic">p</span>) and <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>f</mi> </mrow> <mrow> <mi>s</mi> </mrow> </msub> </mrow> </semantics></math>.</p>
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<p>Circuit layout of (<b>a</b>) SAW extractor and (<b>b</b>) UHB + MHB diplexer. The unit is µm. The black boxes represent SAW resonators and the numerical insets correspond to those in <a href="#micromachines-15-00089-f002" class="html-fig">Figure 2</a> and <a href="#micromachines-15-00089-f004" class="html-fig">Figure 4</a>, respectively.</p>
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<p>Realized circuit for (<b>a</b>) SAW extractor and (<b>b</b>) UHB + MHB diplexer.</p>
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<p>Realized circuit for (<b>a</b>) SAW extractor and (<b>b</b>) UHB + MHB diplexer.</p>
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<p>Comparison of experimental and simulated results for SAW extractor: (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>21</mn> </mrow> </msub> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>31</mn> </mrow> </msub> </mrow> </semantics></math> and (<b>b</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>23</mn> </mrow> </msub> </mrow> </semantics></math>.</p>
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<p>Comparison of experimental and simulated results for UHB + MHB diplexer: (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>21</mn> </mrow> </msub> </mrow> </semantics></math>, (<b>b</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>31</mn> </mrow> </msub> </mrow> </semantics></math>, (<b>c</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math>, and (<b>d</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>23</mn> </mrow> </msub> </mrow> </semantics></math>.</p>
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<p>Comparison of experimental and simulated results for UHB + MHB diplexer: (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>21</mn> </mrow> </msub> </mrow> </semantics></math>, (<b>b</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>31</mn> </mrow> </msub> </mrow> </semantics></math>, (<b>c</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math>, and (<b>d</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>S</mi> </mrow> <mrow> <mn>23</mn> </mrow> </msub> </mrow> </semantics></math>.</p>
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15 pages, 3154 KiB  
Article
Inkjet-Printed Interdigitated Capacitors for Sensing Applications: Temperature-Dependent Electrical Characterization at Cryogenic Temperatures down to 20 K
by Giovanni Gugliandolo, Andrea Alimenti, Mariangela Latino, Giovanni Crupi, Kostiantyn Torokhtii, Enrico Silva and Nicola Donato
Instruments 2023, 7(3), 20; https://doi.org/10.3390/instruments7030020 - 19 Jul 2023
Cited by 2 | Viewed by 1833
Abstract
Microwave transducers are widely used for sensing applications in areas such as gas sensing and microfluidics. Inkjet printing technology has been proposed as a promising method for fabricating such devices due to its capability to produce complex patterns and geometries with high precision. [...] Read more.
Microwave transducers are widely used for sensing applications in areas such as gas sensing and microfluidics. Inkjet printing technology has been proposed as a promising method for fabricating such devices due to its capability to produce complex patterns and geometries with high precision. In this work, the temperature-dependent electrical properties of an inkjet-printed single-port interdigitated capacitor (IDC) were investigated at cryogenic temperatures down to 20 K. The IDC was designed and fabricated using inkjet printing technology, while its reflection coefficient was measured using a vector network analyzer in a cryogenic measurement setup and then transformed into the corresponding admittance. The resonant frequency and quality factor (Q-factor) of the IDC were extracted as functions of the temperature and their sensitivity was evaluated. The results showed that the resonant frequency shifted to higher frequencies as the temperature was reduced, while the Q-factor increased as the temperature decreased. The trends and observations in the temperature-dependent electrical properties of the IDC are discussed and analyzed in this paper, and are expected to be useful in future advancement of the design and optimization of inkjet-printed microwave transducers for sensing applications and cryogenic electronics. Full article
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Figure 1

Figure 1
<p>Sketch of the microwave dielectric-loaded resonator used to measure the dielectric constant of the FR4 substrate. The lower and upper bases are made of brass, as is the sample holder. The dielectric crystal employed in the measurement is a sapphire cylinder of 5 mm height and 8 mm diameter. The TE<sub>011</sub> mode is generated by means of coaxial cables that are ended with magnetic loops and operate at approximately 12.5 GHz. The dielectric sample is loaded beneath the upper base, and the resonator is closed by placing a 500 g weight on it. For further details, please refer to [<a href="#B42-instruments-07-00020" class="html-bibr">42</a>].</p>
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<p>Calibration curves <math display="inline"><semantics><mrow><msup><mi>ε</mi><mo>′</mo></msup><mrow><mo>(</mo><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>/</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub><mo>)</mo></mrow></mrow></semantics></math> (black dots—left scale) and <math display="inline"><semantics><mrow><mi>η</mi><mo>(</mo><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>/</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub><mo>)</mo></mrow></semantics></math> (red triangles—right scale), with <math display="inline"><semantics><mrow><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>=</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>s</mi><mi>a</mi><mi>m</mi><mi>p</mi><mi>l</mi><mi>e</mi></mrow></msub><mo>−</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub></mrow></semantics></math>, i.e., the difference between the resonance frequency <math display="inline"><semantics><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>s</mi><mi>a</mi><mi>m</mi><mi>p</mi><mi>l</mi><mi>e</mi></mrow></msub></semantics></math> measured when the sample iswas loaded into the resonator and that of <math display="inline"><semantics><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi>i</mi><mi>r</mi></mrow></msub></semantics></math> measured with the reference (air). The continuous curves were obtained by a second-order polynomial fit: <math display="inline"><semantics><mrow><msup><mi>ε</mi><mo>′</mo></msup><mo>=</mo><mo>−</mo><mn>2.81</mn><mrow><mo>(</mo><mn>3</mn><mo>)</mo></mrow><mo>×</mo><msup><mn>10</mn><mn>4</mn></msup><msup><mrow><mo>(</mo><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>/</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub><mo>)</mo></mrow><mn>2</mn></msup><mo>−</mo><mn>1.002</mn><mrow><mo>(</mo><mn>2</mn><mo>)</mo></mrow><mo>×</mo><msup><mn>10</mn><mn>3</mn></msup><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>/</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub><mo>+</mo><mn>1.002</mn><mrow><mo>(</mo><mn>2</mn><mo>)</mo></mrow></mrow></semantics></math> with <math display="inline"><semantics><mrow><mn>1</mn><mo>−</mo><msup><mi>R</mi><mn>2</mn></msup><mo>=</mo><mn>2.86</mn><mo>×</mo><msup><mn>10</mn><mrow><mo>−</mo><mn>6</mn></mrow></msup></mrow></semantics></math> and <math display="inline"><semantics><mrow><mi>η</mi><mo>=</mo><mn>72.95</mn><mrow><mo>(</mo><mn>7</mn><mo>)</mo></mrow><msup><mrow><mo>(</mo><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>/</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub><mo>)</mo></mrow><mn>2</mn></msup><mo>−</mo><mn>2.1400</mn><mrow><mo>(</mo><mn>3</mn><mo>)</mo></mrow><mo>Δ</mo><msub><mi>f</mi><mn>0</mn></msub><mo>/</mo><msub><mi>f</mi><mrow><mn>0</mn><mo>,</mo><mi>a</mi><mi mathvariant="normal">i</mi><mi>r</mi></mrow></msub><mo>+</mo><mn>1.9720</mn><mrow><mo>(</mo><mn>4</mn><mo>)</mo></mrow><mo>×</mo><msup><mn>10</mn><mrow><mo>−</mo><mn>3</mn></mrow></msup></mrow></semantics></math> with <math display="inline"><semantics><mrow><mn>1</mn><mo>−</mo><msup><mi>R</mi><mn>2</mn></msup><mo>=</mo><mn>1.38</mn><mo>×</mo><msup><mn>10</mn><mrow><mo>−</mo><mn>8</mn></mrow></msup></mrow></semantics></math>. The uncertainty bars are within the dimensions of the symbols.</p>
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<p>A 2D sketch of the fabricated prototype. The IDC consists of a coplanar structure with nine fingers in a parallel configuration, and is printed on a 1.6-mm-thick FR4 substrate. All of the prototype’s nominal dimensions are reported in the figure.</p>
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<p>Photos of the IDC prototype: (<b>a</b>) Voltera V-One during the printing process and (<b>b</b>) fabricated prototype after curing and SMA connector soldering.</p>
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<p>Measurement (blue) and simulation (orange) of the admittance of the IDC in the frequency range from 100 MHz to 6 GHz, depicting (<b>a</b>) real and (<b>b</b>) imaginary parts. The measured <span class="html-italic">Y</span> is calculated from the IDC reflection coefficient acquired with the VNA in the same frequency range.</p>
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<p>Study of the IDC fabrication method’s reproducibility. Three samples were compared in terms of of the admittance parameter for both the real and imaginary parts, with the respective results shown in (<b>a</b>) and (<b>b</b>).</p>
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<p>Photo of the IDC prototype inside the stainless-steel chamber placed on the cryogenic thermal chuck. The power resistor and cryogenic resistive extensometer used for temperature actuation and control are depicted as well.</p>
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<p>Schematic representation of the cryogenic measurement system.</p>
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<p>Temperature dependence of the resonance at 932 MHz: (<b>a</b>) depicts the relationship between the resonant frequency and the temperature; the trend is modeled with a fourth-order polynomial function with a <math display="inline"><semantics><mrow><msup><mi>R</mi><mn>2</mn></msup><mo>=</mo><mn>0.984</mn></mrow></semantics></math> (the red dashed line), while the sensitivity towards the temperature is shown in (<b>b</b>).</p>
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<p>Temperature dependence of the resonance at 932 MHz: (<b>a</b>) depicts the relationship between the Q-factor and the temperature; the trend can be considered linear with a good approximation (<math display="inline"><semantics><mrow><msup><mi>R</mi><mn>2</mn></msup><mo>&gt;</mo><mn>0.98</mn></mrow></semantics></math>) (the red dashed line), while The sensitivity towards the temperature is shown in (<b>b</b>) and is about <math display="inline"><semantics><mrow><mo>−</mo><mn>0.2</mn></mrow></semantics></math> K<math display="inline"><semantics><msup><mrow/><mrow><mo>−</mo><mn>1</mn></mrow></msup></semantics></math>.</p>
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<p>Temperature dependence of the resonance at 2.388 GHz: (<b>a</b>) depits the relationship between the resonant frequency and the temperature; the trend is modeled with a third-order polynomial function with a <math display="inline"><semantics><mrow><msup><mi>R</mi><mn>2</mn></msup><mo>=</mo><mn>0.996</mn></mrow></semantics></math> (the red dashed line), while the sensitivity towards the temperature is shown in (<b>b</b>).</p>
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<p>Temperature dependence of the resonance at 2.388 GHz: (<b>a</b>) depicts the relationship between the Q-factor and the temperature; The trend can be considered linear with a good approximation (<math display="inline"><semantics><mrow><msup><mi>R</mi><mn>2</mn></msup><mo>&gt;</mo><mn>0.98</mn></mrow></semantics></math>) (the red dashed line), while the sensitivity towards the temperature is shown in (<b>b</b>) and is about <math display="inline"><semantics><mrow><mo>−</mo><mn>0.2</mn></mrow></semantics></math> K<math display="inline"><semantics><msup><mrow/><mrow><mo>−</mo><mn>1</mn></mrow></msup></semantics></math>.</p>
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16 pages, 3997 KiB  
Article
Microwave Sensor for the Determination of DMSO Concentration in Water–DMSO Binary Mixture
by Supakorn Harnsoongnoen and Benjaporn Buranrat
Micromachines 2023, 14(7), 1378; https://doi.org/10.3390/mi14071378 - 5 Jul 2023
Cited by 1 | Viewed by 1932
Abstract
This research aims to develop a microwave sensor to accurately measure the concentration of dimethyl sulfoxide (DMSO) in water–DMSO binary mixtures. The proposed sensor will utilize microwave frequency measurements to determine the DMSO concentration, providing a non-invasive and efficient method for analyzing DMSO [...] Read more.
This research aims to develop a microwave sensor to accurately measure the concentration of dimethyl sulfoxide (DMSO) in water–DMSO binary mixtures. The proposed sensor will utilize microwave frequency measurements to determine the DMSO concentration, providing a non-invasive and efficient method for analyzing DMSO solutions. The research will involve the design, fabrication, and testing of the sensor, as well as the development of an appropriate calibration model. The outcomes of this study will contribute to improved monitoring and quality control in various fields, including pharmaceuticals, chemical synthesis, and biomedical research. The binary mixtures of dimethyl sulfoxide (DMSO) and water with varying concentrations were investigated in the frequency range of 1 GHz to 5 GHz at room temperature using a microwave sensor. The proposed microwave sensor design was based on an interdigital capacitor (IDC) microstrip antenna loaded with a hexagonal complementary ring resonator (HCRR). The performance of the sensor, fabricated using the print circuit board (PCB) technique, was validated through simulations and experiments. The reflection coefficient (S11) and resonance frequency (Fr) of binary mixtures of DMSO and water solutions were recorded and analyzed for DMSO concentrations ranging from 0% v/v to 75% v/v. Mathematical models were developed to analyze the data, and laboratory tests showed that the sensor can detect levels of DMSO/water binary mixtures. The sensor is capable of detecting DMSO concentrations ranging from 0% v/v to 75% v/v, with a maximum sensitivity of 0.138 dB/% for S11 and ΔS11 and 0.2 MHz/% for Fr and ΔFr at a concentration of 50% v/v. The developed microwave sensor can serve as an alternative for detecting DMSO concentrations in water using a simple and cost-effective technique. This method can effectively analyze a wide range of concentrations, including highly concentrated solutions, quickly and easily. Full article
(This article belongs to the Special Issue Recent Advances in Microwave Components and Devices)
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<p>The proposed planar microwave sensor (<b>a</b>) layout and (<b>b</b>) sensor fabrication.</p>
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<p>Modeling the proposed sensor using an equivalent circuit.</p>
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<p>Comparison of simulated and measured S<sub>11</sub> spectra.</p>
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<p>Measurement setup.</p>
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<p>The S<sub>11</sub> spectra in frequency range of 1 GHz–5 GHz for free space, empty tube, DI water and different concentrations of DMSO/water binary mixture.</p>
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<p>The S<sub>11</sub> spectra and smoothed data spectra were obtained from measurements of DI water and different concentrations of DMSO/water binary mixture samples versus the 3.5 GHz–3.75 GHz frequency range.</p>
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<p>The linearity of (<b>a</b>) S<sub>11</sub> and (<b>b</b>) F<sub>r</sub> with the different concentrations of DMSO.</p>
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<p>The linearity of (<b>a</b>) S<sub>11</sub> and (<b>b</b>) F<sub>r</sub> with the different concentrations of DMSO.</p>
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<p>The linearity of (<b>a</b>) ΔS<sub>11</sub> and (<b>b</b>) ΔF<sub>r</sub> with the different concentrations of DMSO.</p>
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<p>The linearity of (<b>a</b>) ΔS<sub>11</sub> and (<b>b</b>) ΔF<sub>r</sub> with the different concentrations of DMSO.</p>
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<p>The sensitivity of the sensor and the parameter sensing of (<b>a</b>) S<sub>11</sub> and ΔS<sub>11</sub> and (<b>b</b>) F<sub>r</sub> and ΔF vary with different concentrations of DMSO.</p>
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21 pages, 6141 KiB  
Article
Miniaturized Antenna Array-Based Novel Metamaterial Technology for Reconfigurable MIMO Systems
by Humam Hussein, Ferhat Atasoy and Taha A. Elwi
Sensors 2023, 23(13), 5871; https://doi.org/10.3390/s23135871 - 25 Jun 2023
Cited by 3 | Viewed by 2045
Abstract
In this work, a highly miniaturized microstrip antenna array based on two elements is proposed for multiple inputs multiple outputs (MIMO) application systems at sub-6 GHz frequency bands. The antenna is structured from a meander line in conjugate with an interdigital capacitor when [...] Read more.
In this work, a highly miniaturized microstrip antenna array based on two elements is proposed for multiple inputs multiple outputs (MIMO) application systems at sub-6 GHz frequency bands. The antenna is structured from a meander line in conjugate with an interdigital capacitor when excited through the monopole basic antenna. The proposed antenna elements are separated with a Minkowski factor-shaped metamaterial (MTM) column to achieve a separation distance (D) of 0.08λ at 3 GHz when printed on an FR-4 substrate. Later on, the antenna performance in terms of bandwidth and gain is controlled using a photonic process based on optical active switches based on light-dependent resistances (LDR). Therefore, the reconfiguration complexity with such a technique can be eliminated significantly without the need for a biasing circuit. The antenna design was conducted through several parametric studies to arrive at the optimal design that realizes the frequency bandwidth between 3 and 5.5 GHz with a maximum gain of about 4.5 dBi when all LDR terminals are off. For a wireless channel performance study-based massive MIMO environment, the proposed antenna is suitable to be configured in arrays of 64 × 64 elements. From this study, it was found the maximum bit error rate (BER) does not exceed 0.15 with a channel capacity (CC) of 2 Gbps. For validation, the antenna was fabricated based on two elements and tested experimentally. Finally, it was revealed that the measured results agree very well with simulations after comparing the theoretical calculations with the measured data. Full article
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<p>Antenna geometrical details in mm scale: (<b>a</b>) front view and (<b>b</b>) back view.</p>
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<p>The proposed MTM structure circuit model: (<b>a</b>) equivalent circuit, (<b>b</b>) S-parameters results, and (<b>c</b>) retrieved electromagnetic properties in terms of ε<sub>r</sub> and µ<sub>r</sub>.</p>
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<p>The obtained S<sub>11</sub> spectra variation for the proposed antenna with changing Xg.</p>
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<p>The obtained S<sub>11</sub> spectra variation for the proposed antenna with changing Yg.</p>
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<p>The obtained S<sub>11</sub> spectra variation for the proposed antenna with changing Xm.</p>
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<p>Obtained antenna performance variation with changing IDC iteration: (<b>a</b>) S<sub>11</sub> and (<b>b</b>) gain spectra.</p>
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<p>The obtained S<sub>11</sub> and gain spectra for the proposed antenna with changing IDC iteration: (<b>a</b>) S11 and (<b>b</b>) gain spectra..</p>
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<p>The obtained S-parameters spectra for the proposed antenna array with changing D: (<b>a</b>) S<sub>11</sub> and (<b>b</b>) S<sub>12</sub> spectra.</p>
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<p>The obtained S-parameters spectra for the proposed antenna array with and without MTM defects.</p>
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<p>The obtained S-parameters spectra for the proposed antenna array with changing MTM defects number: (<b>a</b>) S<sub>11</sub> and (<b>b</b>) S<sub>12</sub> spectra.</p>
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<p>Array performance in terms of (<b>a</b>) the obtained TRAC spectra at different signal phase excitations and (<b>b</b>) radiation efficiency spectra.</p>
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<p>The obtained S-parameters spectra for the proposed antenna array with changing switching scenarios: (<b>a</b>) S<sub>11</sub> and (<b>b</b>) S<sub>12</sub> spectra.</p>
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<p>The proposed prototype: (<b>a</b>) front view and (<b>b</b>) back view.</p>
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<p>The measured S-parameters spectra for the proposed antenna array with changing switching scenarios: (<b>a</b>) S<sub>11</sub> and (<b>b</b>) S<sub>12</sub> spectra.</p>
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<p>Measured antenna radiation patterns at different frequency bands: (<b>a</b>) 3.5 GHz, (<b>b</b>) 4 GHz, (<b>c</b>) 4.5 GHz, (<b>d</b>) 5 GHz, and (<b>e</b>) 5.5 GHz.</p>
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<p>Antenna array performance in terms of ECC, DG, MEG, CCL, and TRAC.</p>
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<p>Channel performance calculations at different frequency bands: (<b>a</b>) BER and (<b>b</b>) CC.</p>
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8 pages, 1246 KiB  
Proceeding Paper
Current Density-Voltage (J-V) Characterization of Monolithic Nanolaminate Capacitors
by Zeinab Mousavi Karimi and Jeffrey A. Davis
Mater. Proc. 2023, 14(1), 54; https://doi.org/10.3390/IOCN2023-14590 - 12 Jun 2023
Viewed by 1481
Abstract
In a world of miniaturized electronics, there is a rapidly increasing need for reliable, efficient, and compact energy storage systems with low-loss dielectrics. To address this need, this work proposes the development of compact, micro-capacitive energy storage devices compatible with IC processing so [...] Read more.
In a world of miniaturized electronics, there is a rapidly increasing need for reliable, efficient, and compact energy storage systems with low-loss dielectrics. To address this need, this work proposes the development of compact, micro-capacitive energy storage devices compatible with IC processing so that they can be integrated monolithically on-chip. There are two main approaches to the fabrication of integrated on-chip micro-supercapacitor energy storage devices: interdigitated electrode (IDE) devices and parallel plate electrode (PPE) devices. As part of the design of such systems, this study aims to investigate the behavior of current density-voltage (J-V) in homogeneous and heterogeneous IDE and PPE devices to determine whether the anomalies between the interfaces of dielectric materials in such structures affect their leakage current. The ultimate goal is to design a solid-state capacitor energy storage module with low-loss dielectrics, high energy densities, and improved areal capacitance density that can offer a high number of charge/discharge cycles for portable power electronics. An understanding of J-V characteristics is crucial in achieving this objective. Specifically, this paper will explore and investigate nanolaminate, solid-state PPE, and IDE capacitive energy storage “modules” fabricated using nanolithographic techniques. The dielectric layers in these structures are composed of alternating nanolaminate layers of thin higher-k Al2O3 and lower-k SiO2. Recent findings have shown that capacitive energy storage devices made from a large number of these on-chip multilayer nanolaminate energy storage PPE (MNES-PPE) structures that utilize the interfacial anomalies of thin high-k/SiO2 nanolaminates could have the potential to overcome many of the limitations of current compact energy storage technologies. Preliminary projections indicate that these high-density nanolaminate capacitors with laminate thicknesses around 5 nm could produce devices with high volumetric energy densities (290 J/cm3) that are significantly higher than conventional supercapacitors (20 J/cm3). Full article
(This article belongs to the Proceedings of The 4th International Online Conference on Nanomaterials)
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Figure 1
<p>Parallel plate electrode (PPE) devices with (<b>a</b>) SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, (<b>b</b>) Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math> and (<b>c</b>) Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math> dielectrics.</p>
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<p>Interdigitated electrode (IDE) devices with (<b>a</b>) SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, (<b>b</b>) Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>/Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>, (<b>c</b>) Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>, (<b>d</b>) Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, (<b>e</b>) Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math> and (<b>f</b>) Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math> dielectrics.</p>
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<p>(<b>a</b>) J-V characteristic plots for the SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>, and Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math> homogeneous IDE and PPE devices. Solid blue line represents IDE SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>; blue dashed line represents IDE Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>/Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>; blue dotted line represents IDE Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>; solid red line represents PPE SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>; red dashed line represents PPE Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>; red dotted line represents PPE Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>. (<b>b</b>) Plot of electric field vs. electrode spacing characteristics between two electrodes for homogeneous (<b>c</b>) PPE SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, and (<b>d</b>) IDE SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math> devices with 200 nm electrode spacing at 0.1 V.</p>
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<p>J-V characteristic plots for the heterogeneous and homogeneous IDE devices with (<b>a</b>) Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, and Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>, (<b>b</b>) Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>, and Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>, and (<b>c</b>) Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math>, Si<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>N<math display="inline"><semantics><msub><mrow/><mn>4</mn></msub></semantics></math> dielectrics.</p>
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<p>A zoomed FEM model of an multilayer nanolaminate energy storage (MNES) PPE structure composed of 18 alternating layers of Al<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math>O<math display="inline"><semantics><msub><mrow/><mn>3</mn></msub></semantics></math>/SiO<math display="inline"><semantics><msub><mrow/><mn>2</mn></msub></semantics></math> with 1 nm high-k interface.</p>
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17 pages, 6425 KiB  
Article
Advances in a Microwave Sensor-Type Interdigital Capacitor with a Hexagonal Complementary Split-Ring Resonator for Glucose Level Measurement
by Supakorn Harnsoongnoen and Benjaporn Buranrat
Chemosensors 2023, 11(4), 257; https://doi.org/10.3390/chemosensors11040257 - 20 Apr 2023
Cited by 13 | Viewed by 2859
Abstract
This study involved the creation and assessment of a microwave sensor to measure glucose levels in aqueous solutions without invasiveness. The sensor design utilized a planar interdigital capacitor (IDC) loaded with a hexagonal complementary split-ring resonator (HCSRR). The HCSRR was chosen for its [...] Read more.
This study involved the creation and assessment of a microwave sensor to measure glucose levels in aqueous solutions without invasiveness. The sensor design utilized a planar interdigital capacitor (IDC) loaded with a hexagonal complementary split-ring resonator (HCSRR). The HCSRR was chosen for its ability to generate a highly intense electric field that is capable of detecting variations in the dielectric characteristics of the specimen. A chamber tube was used to fill glucose solutions at the sensor’s sensitive area, and changes in the device’s resonance frequency (Fr) and reflection coefficient (S11) were used to measure glucose levels. Fitting formulas were developed to analyze the data, and laboratory tests showed that the sensor could accurately measure glucose levels within a range of 0–150 mg/dL. At a concentration of 37.5 mg/dL, the sensitivity based on S11 and Fr reached maximum values of 10.023 dB per mg/dL and 1.73 MHz per mg/dL, respectively. This implies that the sensor put forward has the possibility of being utilized in medical settings for the monitoring of glucose levels. Full article
Show Figures

Figure 1

Figure 1
<p>The proposed sensor: (<b>a</b>) geometry, (<b>b</b>) an electric field distribution, (<b>c</b>) a cross-sectional view at sensing area, and (<b>d</b>) an equivalent circuit model.</p>
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<p>The proposed sensor: (<b>a</b>) geometry, (<b>b</b>) an electric field distribution, (<b>c</b>) a cross-sectional view at sensing area, and (<b>d</b>) an equivalent circuit model.</p>
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<p>The fabricated hexagonal CSRR microwave sensor is shown in (<b>a</b>) the top view and (<b>b</b>) the bottom view.</p>
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<p>Measurement setup.</p>
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<p>Comparison of simulated and measured S<sub>11</sub> spectra.</p>
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<p>The reflection coefficients of the microwave sensor were measured for free space, an empty tube, DI water, and glucose concentrations of 37.5, 50, 75, and 150 mg/dL in the solution.</p>
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<p>Sensor response signal in peak 1 (<b>a</b>) reflection coefficient spectra, and (<b>b</b>) S<sub>11</sub> and (<b>c</b>) F<sub>r</sub> of the microwave sensor, as measured by the reflection coefficient for glucose concentrations of 0, 37.5, 50, 75, and 150 mg/dL in the solution.</p>
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<p>Sensor response signal in peak 2. (<b>a</b>) Smooth data and effect of glucose concentration on (<b>b</b>) S<sub>11</sub> and (<b>c</b>) resonant frequency on of the microwave sensor, as measured by the reflection coefficient for glucose concentrations of 0, 37.5, 50, 75, and 150 mg/dL in the solution.</p>
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<p>Sensor response signal in peak 2. (<b>a</b>) Smooth data and effect of glucose concentration on (<b>b</b>) S<sub>11</sub> and (<b>c</b>) resonant frequency on of the microwave sensor, as measured by the reflection coefficient for glucose concentrations of 0, 37.5, 50, 75, and 150 mg/dL in the solution.</p>
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<p>Plot of glucose concentration versus (<b>a</b>) ΔS<sub>11</sub> and (<b>b</b>) ΔF<sub>r</sub>.</p>
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<p>Sensitivity of proposed microwave sensor.</p>
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