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WO2022042463A1 - 滤波器带外抑制优化方法和滤波器、多工器、通信设备 - Google Patents

滤波器带外抑制优化方法和滤波器、多工器、通信设备 Download PDF

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Publication number
WO2022042463A1
WO2022042463A1 PCT/CN2021/114004 CN2021114004W WO2022042463A1 WO 2022042463 A1 WO2022042463 A1 WO 2022042463A1 CN 2021114004 W CN2021114004 W CN 2021114004W WO 2022042463 A1 WO2022042463 A1 WO 2022042463A1
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Prior art keywords
resonator
filter
series
parallel
electromechanical coupling
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PCT/CN2021/114004
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English (en)
French (fr)
Inventor
徐利军
庞慰
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诺思(天津)微系统有限责任公司
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Publication of WO2022042463A1 publication Critical patent/WO2022042463A1/zh

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/48—Coupling means therefor
    • H03H9/50—Mechanical coupling means
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02007—Details of bulk acoustic wave devices
    • H03H9/02086—Means for compensation or elimination of undesirable effects
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/48—Coupling means therefor
    • H03H9/52—Electric coupling means
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/54—Filters comprising resonators of piezoelectric or electrostrictive material
    • H03H9/542—Filters comprising resonators of piezoelectric or electrostrictive material including passive elements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/54—Filters comprising resonators of piezoelectric or electrostrictive material
    • H03H9/58—Multiple crystal filters
    • H03H9/60—Electric coupling means therefor
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/70—Multiple-port networks for connecting several sources or loads, working on different frequencies or frequency bands, to a common load or source
    • H03H9/703—Networks using bulk acoustic wave devices
    • H03H9/706—Duplexers

Definitions

  • the present invention relates to the technical field of filters, and in particular, to a filter out-of-band suppression optimization method, a filter, a multiplexer, and a communication device.
  • filters, duplexers and multiplexers which are key components of RF front-end, have received extensive attention, especially in the fastest growing field of personal mobile communications. application.
  • filters and duplexers that are widely used in the field of personal mobile communications are mostly made of surface acoustic wave resonators or bulk acoustic wave resonators.
  • BAW resonators Compared with surface acoustic wave resonators, BAW resonators have better performance.
  • BAW resonators have the characteristics of high Q value, wide frequency coverage, and good heat dissipation performance, which are more suitable for the development needs of 5G communication.
  • the resonance of BAW resonators is generated by mechanical waves, not electromagnetic waves.
  • the wavelength of mechanical waves is shorter than that of electromagnetic waves. Therefore, the volume of BAW resonators and the filters they consist of is greatly reduced compared to traditional electromagnetic filters. In addition, due to The crystal growth of piezoelectric crystals can be well controlled, the loss of the resonator is extremely small, and the quality factor is high, which can cope with complex design requirements such as steep transition band and low insertion loss.
  • BAW resonators are suitable for frequency bands above 1.2GHz, but not suitable for frequency bands below 1.2GHz.
  • the first reason is that when the frequency is low, the piezoelectric layer is thicker. , resulting in a large resonator area, which is not conducive to miniaturization.
  • scandium-doped aluminum nitride technology and technology this problem has been solved.
  • the second reason is that when the frequency is low, the high order of the resonator The resonance amplitude is very strong.
  • the invention provides an optimization method for out-of-band suppression of a filter, a filter, a multiplexer, and a communication device, which can not only keep the passband coverage of the filter unchanged, but also solve the problem of poor harmonic suppression of the low-frequency bulk acoustic wave filter. At the same time, it can also ensure the suppression balance in the harmonic suppression region.
  • a method for optimizing out-of-band suppression of a filter includes a plurality of series resonators and a plurality of parallel resonators, the method includes: adjusting the piezoelectricity of the series resonators and the parallel resonators The thickness of the layer, so that the effective electromechanical coupling coefficient of the parallel resonator is larger than the initial value of the effective electromechanical coupling coefficient of the series resonator, and the sum of the two said initial values is a fixed value, and the series resonance of the harmonics of the parallel resonator The frequency point is located between the series resonance frequency point and the parallel resonance frequency point of the harmonics of the series resonator; when the fundamental frequency of the filter meets the requirements of the index and the low frequency suppression amplitude and high frequency suppression amplitude of the filter harmonic region are not equal , perform the following steps A or B until the low frequency suppression amplitude and high frequency suppression amplitude in the harmonic region of the filter are equal and greater than the
  • each parallel resonator is connected with a grounding inductor, and the inductance value of the grounding inductor is smaller than a preset value.
  • the step of adjusting the thicknesses of the piezoelectric layers of the series resonators and the parallel resonators includes: fabricating the series resonators and the parallel resonators on different wafers, and adjusting the thicknesses of the piezoelectric layers on the two wafers respectively. , so that the thicknesses of the piezoelectric layers of the series and parallel resonators are different.
  • the initial value of the effective electromechanical coupling coefficient of the parallel resonator is 1% to 2% larger than the initial value of the effective electromechanical coupling coefficient of the series resonator, and the sum of the two is 4-5 times the relative bandwidth of the filter.
  • the specified value is 30dB.
  • step A or step B the initial value of the effective electromechanical coupling coefficient of the parallel resonator and the initial value of the effective electromechanical coupling coefficient of the series resonator are increased or decreased by 0.5%.
  • the preset value is 0.5nH.
  • a filter comprising an upper wafer, a lower wafer, multiple series resonators and multiple parallel resonators, all parallel resonators are arranged on the first surface of the upper wafer, and all are connected in series
  • the resonator is arranged on the first surface of the lower wafer; the upper wafer and the lower wafer are superimposed to form a package structure; inside the package structure, the first surface of the upper wafer and the first surface of the lower wafer are arranged in parallel and opposite to each other , the series resonator and the parallel resonator are bonded by butt pins to form a multi-stage series-parallel filter circuit; wherein the thickness of the piezoelectric layer of the multiple series resonators is different from the thickness of the piezoelectric layer of the parallel resonator, and , the effective electromechanical coupling coefficient of the parallel resonator is greater than the effective electromechanical coupling coefficient of the series resonator, and the low frequency and high frequency
  • the filter circuit further includes a grounding inductor, the first end of the grounding inductor is connected to the parallel resonator, and the second end is grounded; the inductance value of the grounding inductor is less than a preset value.
  • a duplexer including the above filter.
  • Fig. 1 is the impedance curve schematic diagram of the low frequency resonator in the filter
  • Fig. 2 is the impedance curve schematic diagram of two resonators in the filter
  • FIG. 3 is a schematic diagram of a passband curve of a filter
  • Figure 4 is a schematic diagram of the comparison of resonator impedance curves
  • FIG. 5 is a schematic diagram of a passband curve of a filter
  • FIG. 6 is a schematic diagram of a passband curve of a filter
  • Figure 7 is a schematic diagram showing the comparison of passband curves of parallel resonators with different piezoelectric layer thicknesses
  • FIG. 8 is a schematic flowchart of a filter out-of-band suppression optimization method provided by an embodiment of the present invention.
  • Fig. 9 is the topology structure schematic diagram of filter
  • FIG. 10 is a schematic diagram of a passband curve of a simulated filter
  • Figure 11 is a schematic diagram of the passband curve after filter optimization
  • Figure 12 is a schematic diagram showing the comparison of the change curve of the series resonance frequency point after the parallel resonator in the filter is connected to the ground inductance;
  • Figure 13 is a schematic diagram showing the comparison of the pass-band curves after the parallel resonator is connected to the ground inductance
  • FIG. 14 is a cross-sectional view of a filter package structure according to an embodiment of the present invention.
  • FIG. 15 is a front view of an upper wafer in a filter package structure provided by an embodiment of the present invention.
  • FIG. 16 is a front view of the lower wafer in the filter package structure provided by the embodiment of the present invention.
  • the technical solutions in the embodiments of the present invention can keep the passband coverage of the filter unchanged, and can solve the problem of poor harmonic suppression of low-frequency bulk acoustic wave filters. At the same time, it can also ensure the suppression balance in the harmonic suppression region.
  • it can also ensure the suppression balance in the harmonic suppression region.
  • Figure 1 is a schematic diagram of the impedance curve of the low frequency resonator in the filter.
  • the resonator is a typical resonator structure, which includes superimposed upper electrode, piezoelectric layer and lower electrode.
  • the curve has two resonance regions, namely the fundamental frequency resonance region and the harmonic resonance region.
  • the fundamental frequency resonance region has a lower frequency.
  • the resonance is about 900MHz, including the series resonance frequency and the parallel resonance frequency.
  • the impedance Rp of the parallel resonance frequency is about 6500 ohms, while the frequency of the harmonic resonance region is higher, and the resonance is about 3000MHz, including the series resonance frequency. and the parallel resonance frequency point, wherein the Rp of the parallel resonance frequency point is 800 ohms, which has a higher impedance value.
  • FIG. 2 is a schematic diagram of impedance curves of two resonators in the filter.
  • the solid line in the figure is the impedance curve of the series resonator, which is exactly the same as the impedance curve shown in Figure 1, and the dotted line is the impedance curve of the parallel resonator, which adopts the mass-loaded parallel resonator.
  • the method realizes frequency shifting.
  • This curve is similar to the impedance curve of the series resonator. It also includes two resonance regions, namely the fundamental frequency resonance region and the harmonic resonance region.
  • the fundamental frequency resonance region has a lower frequency, and the resonance is around 865MHz.
  • the area includes the series resonance frequency point and the parallel resonance frequency point.
  • the impedance Rp of the parallel resonance frequency point is about 6500 ohms, while the frequency of the harmonic resonance area is higher, and the resonance is about 2900MHz.
  • the harmonic resonance area includes the series resonance frequency point and The parallel resonance frequency point, wherein the Rp of the parallel resonance frequency point is 800 ohms, which has a higher impedance value. Comparing the two curves, it can be seen that the parallel resonance frequency of the fundamental frequency of the parallel resonator is located near the series resonance frequency of the fundamental frequency of the series resonator. form a passband.
  • FIG. 3 is a schematic diagram of the passband curve of the filter.
  • a passband is formed near 900MHz, and a pseudo passband is formed near 2900MHz.
  • the existence of the pseudo passband deteriorates the out-of-band rejection near this frequency.
  • the frequency of the parallel resonator is similar, that is, the series and parallel resonators have the same stack, and only when the mass load is loaded on the parallel resonator, the parallel resonance frequency of the harmonic of the parallel resonator will be located near the series resonance frequency of the harmonic of the series resonator. , thus forming a pseudo-passband, the existence of the pseudo-passband deteriorates the out-of-band suppression of the frequency band, and seriously affects the promotion and use of the BAW filter in the low frequency band. Therefore, it needs to be improved.
  • the series resonator and the parallel resonator of the filter are respectively used with different piezoelectric layer thicknesses, and the thickness of the piezoelectric layer of the parallel resonator is larger than that of the series resonator.
  • thickness that is, the effective electromechanical coupling coefficient of the parallel resonator is greater than the effective electromechanical coupling coefficient of the series resonator, so that in the fundamental frequency band, the parallel resonance frequency of the fundamental frequency of the parallel resonator is located at the series resonance frequency of the fundamental frequency of the series resonator.
  • FIG. 4 is a schematic diagram showing the comparison of the impedance curves of the resonators.
  • the solid line is the impedance curve of the series resonator, which includes two resonance regions, namely the fundamental frequency resonance region and the harmonic resonance region, while the dotted line is the impedance curve of the parallel resonator.
  • the stack is different from the series resonator, and the piezoelectric layer thickness of the parallel resonator is larger than that of the series resonator, that is, the frequency shift is realized by using the piezoelectric layer as a loading mass load.
  • This curve is similar to the impedance curve of the series resonator. , also includes two resonance regions, namely the fundamental frequency resonance region and the harmonic resonance region, the parallel resonance frequency of the fundamental frequency of the parallel resonator is located near the series resonance frequency of the fundamental frequency of the series resonator, and a plurality of the above series and parallel resonators are composed of filter ladder topology, which creates a passband at the fundamental frequency.
  • the piezoelectric layers of the series resonator and the parallel resonator are set as follows: first, the thickness of the piezoelectric layer of the series resonator is set to a certain value (that is, the effective electromechanical coupling coefficient of the series resonator is a constant value), and then the thickness of the piezoelectric layer of the series resonator is set to a certain value.
  • Optimize the thickness of the piezoelectric layer of the parallel resonator ie, the effective electromechanical coupling coefficient of the parallel resonator
  • the suppression difference between the two positions is 16dB, and the suppression amplitude of the two positions is unbalanced;
  • the series resonance frequency of the harmonic deviates from the parallel resonance frequency of the harmonic of the series resonator, and is larger than the parallel resonance frequency of the harmonic of the series resonator, which will lead to the deterioration of high frequency suppression in the harmonic region, indicated by circle 1 in Figure 6.
  • the position suppression can reach 41dB, and the position indicated by circle 2 has only 21dB suppression.
  • the suppression difference between these two positions is 20dB, and the two sides are unbalanced.
  • FIG. 7 is a schematic diagram showing the comparison of passband curves of parallel resonators with different piezoelectric layer thicknesses.
  • the dotted line is the curve when the piezoelectric layer of the parallel resonator is thin
  • the solid line is the curve when the piezoelectric layer of the parallel resonator is thick.
  • the range of the filter varies. If there is suppression on the left and right sides of the passband, during the optimization process, the passband coverage of the filter changes, and the filter will cause the adjacent band suppression to become worse because the passband becomes wider, or because the passband becomes narrower. , resulting in worse sideband insertion loss.
  • the embodiments of the present invention provide an optimization method for filter out-of-band suppression, which can not only maintain the filter passband coverage almost constant during the optimization process, but also solve the problem of poor harmonic suppression of low-frequency bulk acoustic wave filters problem, as well as ensuring the suppression balance in the harmonic suppression region.
  • FIG. 8 is a schematic flowchart of a method for optimizing out-of-band suppression of a filter provided by an embodiment of the present invention.
  • step S81 adjust the thicknesses of the piezoelectric layers of the series resonator and the parallel resonator, so that the thicknesses of the piezoelectric layers of the two are different, so that the initial values of the effective electromechanical coupling coefficients of the two are different, wherein, The two initial values should meet the following two conditions: 1. The initial value of the effective electromechanical coupling coefficient of the parallel resonator is greater than the initial value of the effective electromechanical coupling coefficient of the series resonator; 2.
  • Step S82 determine Whether the series resonance frequency of the harmonics of the parallel resonator is located between the series resonance frequency and the parallel resonance frequency of the harmonics of the series resonator, if so, go to step S83, otherwise return to step S81; step S83: check the filter topology Perform simulation to determine whether the fundamental frequency of the filter meets the index requirements, if so, go to step S84, otherwise return to step S81; Step S84: determine whether the low-frequency suppression amplitude and high-frequency suppression amplitude in the harmonic region are equal, and greater than the specified value, specify The value is generally 30dB; if so, the optimization is over, otherwise, go to step S85; step S85: determine whether the low frequency
  • FIG. 9 is a schematic diagram of the topology structure of the filter.
  • the topology is a 5-4 structure (of course not limited to the 5-4 structure, it can be an MN structure, M and N are natural numbers, here only the 5-4 structure is used as an example), the topology includes 1 series branch and 4 parallel branches.
  • the series branch is composed of series resonators S11, S12, S13, S14 and S15 connected in series between port 1 and port 2.
  • the parallel branch includes parallel resonators and For the grounding inductor, one end of the parallel resonator is connected to the node between two adjacent series resonators, and the other end is connected to the grounding inductor.
  • the first parallel branch includes a parallel resonator P11 and a grounded inductor L11
  • the second parallel branch includes a parallel resonator P12 and a grounded inductor L12
  • the third parallel branch includes a parallel resonator P13 and a grounded inductor L13
  • the fourth parallel branch includes a parallel resonator P13 and a grounded inductor L13.
  • the branch includes a parallel resonator P14 and a grounded inductor L14.
  • the sum of the effective electromechanical coupling coefficients of the resonator and the parallel resonator is 16.3%; the series-parallel resonance frequency point analysis of the harmonics of the series-parallel resonator is carried out, and the thickness of the piezoelectric layer selected above is determined, so that the series-parallel resonance of the harmonics of the parallel resonator is determined.
  • the resonance frequency is just between the series resonance frequency and the parallel resonance frequency of the harmonics of the series resonator; then the filter topology can be simulated and optimized.
  • the above parameters show that the passband insertion loss of the entire filter is less than 1.8dB, which is basically If the fundamental frequency index requirements are met, the next step can be performed to analyze the harmonic suppression of the filter.
  • FIG. 10 is a schematic diagram of the passband curve of the simulated filter. From the curve shown in Figure 10, it can be seen that the worst point of harmonic suppression is only 25dB, which does not meet the requirements. At the same time, it is found that the worst point of harmonic suppression is the high frequency part of the harmonic region, and the low frequency part of the harmonic region is better. up to 40dB.
  • the effective electromechanical coupling coefficient of the parallel resonator is reduced by 0.5%, while the effective electromechanical coupling coefficient of the series resonator is increased by 0.5%, the effective electromechanical coupling coefficient of the parallel resonator is changed to 8.8%, and its piezoelectric layer is changed to 0.87 micron, the effective electromechanical coupling coefficient of the series resonator is changed to 7.5%, and its piezoelectric layer is 0.68 ⁇ m, keeping the sum of the effective electromechanical coupling coefficient of the series resonator and the parallel resonator unchanged at 16.3%.
  • Figure 11 is a schematic diagram of the passband curve after filter optimization.
  • the insertion loss of the fundamental frequency meets the requirements of the index
  • the insertion loss of the entire passband is less than 1.8dB
  • the entire out-of-band suppression of the filter is greater than 40dB, especially in the harmonic region, where the suppression is greater than 40dB, and the low-frequency suppression in the harmonic region Amplitude and high frequency rejection are more balanced.
  • the grounding inductance of the parallel branch in the filter also plays a key role in harmonic suppression.
  • the main reason is that when a parallel resonator is connected in series with a grounding inductance, it will change the position of the fundamental frequency of the resonator and the series resonance frequency in the harmonic region.
  • the position of the series resonance frequency is generally moved to the low frequency, so when the inductance value of the parallel resonator series is large, the series resonance frequency of the harmonics of the parallel resonator may be smaller than the series resonance frequency of the harmonics of the series resonator.
  • FIG. 12 is a schematic diagram showing the comparison of the change curve of the series resonance frequency point after the parallel resonator in the filter is connected to the grounding inductor.
  • the harmonic resonance of the series resonator is marked with a thin solid line in the figure.
  • the harmonic series resonance frequency is at 2.88GHz
  • the parallel resonance frequency is at 2.93GHz.
  • the inductance connected to the parallel resonator When the inductance connected to the parallel resonator is At 0.3nH, its harmonic series resonance frequency is at 2.925GHz, and its parallel resonance frequency is at 2.96GHz. At this time, the series resonance frequency of the parallel resonator harmonic is just at the series resonance frequency of the series resonator harmonic and the parallel resonance frequency. Between the frequency points, with the increase of the series inductance, the harmonic series resonance frequency point moves to the low frequency, that is, when the inductance increases to 0.5nH, the series resonance frequency point of the harmonics of the parallel resonator moves to 2.84GHz. It is located between the series resonance frequency and the parallel resonance frequency of the harmonics of the series resonator, so the suppression of the low frequency band in the harmonic region will be deteriorated. FIG.
  • FIG. 13 is a schematic diagram showing the comparison of pass-band curves after the parallel resonator is connected to the grounded inductor.
  • the solid line in the figure is the corresponding curve when the grounding inductance value is 0nH, the harmonic suppression in this curve is better, and the dotted line is the corresponding curve when the grounding inductance value is 0.5nH, the harmonics in the curve are Rejection deteriorated by 15dB.
  • FIG. 14 is a cross-sectional view of a filter package structure according to an embodiment of the present invention. As shown in FIG. 14 , in the package structure of the filter, all parallel resonators are fabricated on the upper wafer, and all series resonators are fabricated on the lower wafer.
  • Fig. 15 is a front view of the upper wafer in the filter package structure provided by the embodiment of the present invention;
  • Fig. 16 is the front view of the lower wafer in the filter package structure provided by the embodiment of the present invention.
  • the upper wafer includes parallel resonators P11, P12, P13 and P14, as well as ground pins G1, G2, G3, G4 and transfer bonding pins J1, J2, J3, J4;
  • the lower wafer includes series resonators S11, S12, S13, S14 and S15, as well as ground pins G1, G2, G3, G4, transfer bonding pins J1, J2, J3, J4, input pins IN and output pins pin OUT.
  • the upper wafer and the lower wafer are superimposed on top of each other, and the bonding pins J1, J2, J3, J4 are bonded, and the ground pins G1, G2, G3, and G4 are bonded; Through holes, the signal terminals and the ground terminals of the filters manufactured by the upper wafer and the lower wafer are connected to the pads under the lower wafer through vias, and the pads under the lower wafer can be connected to the package through metal solder balls substrate to form a package structure.
  • the thicknesses of the piezoelectric layers of the plurality of series resonators are different from the thicknesses of the piezoelectric layers of the parallel resonators, and the effective electromechanical coupling coefficient of the parallel resonators is larger than the effective electromechanical coupling coefficient of the series resonators.
  • the low frequency suppression amplitude and high frequency suppression amplitude of the harmonic region are equal to and greater than the specified value, such as greater than 30dB. Since the series resonator and the parallel resonator are separately provided on two wafers, the piezoelectric layer can be provided with different thicknesses, and the thickness can be easily adjusted.
  • the filter can not only maintain the filter passband coverage unchanged, but also solve the problem of poor harmonic suppression of the low-frequency bulk acoustic wave filter, and at the same time, it can also ensure the suppression balance in the harmonic suppression region.
  • the embodiment of the present invention also provides a duplexer, which includes the above-mentioned filter. Therefore, the duplexer can also maintain the filter passband coverage unchanged, and can solve the harmonics of the low-frequency bulk acoustic wave filter. The problem of poor suppression and the effect of ensuring the balance of suppression in the harmonic suppression area.
  • Embodiments of the present invention also provide a communication device, which includes the above-mentioned filter. Therefore, the communication device can also maintain the filter passband coverage unchanged, and can solve the problem of poor harmonic suppression of the low-frequency bulk acoustic wave filter. problem, and the effect of ensuring the suppression balance of the harmonic suppression region.

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Abstract

本发明涉及滤波器技术领域，特别地涉及一种滤波器带外抑制优化方法和滤波器、多工器、通信设备。在该方法中，串联谐振器和并联谐振器的有效机电耦合系数可灵活调节，不仅能够维持滤波器通带覆盖范围不变，而且能够解决低频体声波滤波器谐波抑制较差的问题，同时，还能保障谐波抑制区域的抑制平衡。

Description

滤波器带外抑制优化方法和滤波器、多工器、通信设备 技术领域
本发明涉及滤波器技术领域，特别地涉及一种滤波器带外抑制优化方法和滤波器、多工器、通信设备。
背景技术
随着无线通信技术向着多频段、多模方向快速发展，作为射频前端关键部件的滤波器、双工器以及多工器得到广泛关注，特别是在发展最快的个人移动通信领域更是得到广泛应用。目前，在个人移动通信领域应用广泛的滤波器、双工器多是由表面声波谐振器或体声波谐振器制造而成。相较于表面声波谐振器，体声波谐振器性能更胜一筹，体声波谐振器具有Q值高、频率覆盖范围广、散热性能好等特性，更适合5G通信的发展需要。体声波谐振器其谐振由机械波产生，而非电磁波，机械波的波长比电磁波波长短，因此，体声波谐振器及其组成的滤波器体积相对传统的电磁滤波器尺寸大幅度减小；另外，由于压电晶体的晶向生长能够良好控制，谐振器的损耗极小，品质因数高，能够应对陡峭过渡带和低插入损耗等复杂设计要求。
一般情况下，体声波谐振器适合1.2GHz频率以上的频段，其并不适合1.2GHz频率以下的频段，主要有两个方面的原因，第一个原因是频率较低时，压电层较厚，导致谐振器面积较大，不利于小型化，但是，目前随着掺钪氮化铝技术和工艺的出现，已经解决了此问题，第二个原因是频率较低时，谐振器的高次谐振幅度很强，当此类谐振器组成梯型滤波器时，除了形成一个基频通带外，还会在高频段附近形成一个插损较差的滤波器通带，所以会导致滤波器的带外抑制，特别是高频带外抑制恶化，影响滤波器的正常使用。
因此，为了使体声波谐振器能够在低频滤波器中得到应用，如何利用体声波谐振器技术，降低高次谐波对滤波器带外抑制的影响，仍是待解决的技术问题。
发明内容
本发明提供了一种滤波器带外抑制优化方法和滤波器、多工器、通信设备，不仅能够维持滤波器通带覆盖范围不变，而且能够解决低频体声波滤波器谐波抑制较差的问题，同时，还能保障谐波抑制区域的抑制平衡。
本发明的一个方面，提供了一种滤波器带外抑制优化方法，所述滤波器包括多个串联谐振器和多个并联谐振器，该方法包括：调整串联谐振器和并联谐振器的压电层的厚度，使并联谐振器的有效机电耦合系数大于串联谐振器的有效机电耦合系数的初值，并且使两个所述初值之和为固定值，以及使并联谐振器谐波的串联谐振频点位于串联谐振器谐波的串联谐振频点与并联谐振频点之间；在滤波器的基频满足指标要求并且滤波器的谐波区域低频抑制幅度和高频抑制幅度不相等的情况下，执行如下步骤A或步骤B，直至滤波器的谐波区域低频抑制幅度和高频抑制幅度相等且大于指定值，其中：步骤A：若滤波器的谐波区域低频抑制幅度大于高频抑制幅度，则减小并联谐振器的有效机电耦合系数的初值，增大串联谐振器的有效机电耦合系数的初值，并且保持两个所述初值之和为固定值；步骤B：若滤波器的谐波区域低频抑制幅度小于高频抑制幅度，则增大并联谐振器的有效机电耦合系数的初值，减小串联谐振器的有效机电耦合系数的初值，并且保持两个所述初值之和为固定值。
可选地，所述滤波器中，每个并联谐振器均连有接地电感，所述接地电感的电感值小于预设值。
可选地，调整串联谐振器和并联谐振器的压电层的厚度的步骤包括：将串联谐振器和并联谐振器制造在不同的晶圆上，分别调整两块晶圆上压电层的厚度，以使串联谐振器和并联谐振器的压电层的厚度不同。
可选地，并联谐振器的有效机电耦合系数的初值比串联谐振器的有效机电耦合系数的初值大1％～2％，两者之和为滤波器相对带宽的4-5倍。
可选地，所述指定值为30dB。
可选地，所述步骤A或步骤B中，并联谐振器的有效机电耦合系数的初值和串联谐振器的有效机电耦合系数的初值增大或减小0.5％。
可选地，预设值为0.5nH。
本发明另一个方面，还提供了一种滤波器，包括上晶圆、下晶圆、多个串联谐振器和多个并联谐振器，全部并联谐振器设于上晶圆第一表面，全部串联谐振器设于下晶圆的第一表面；上晶圆和下晶圆叠加形成封装结构；在所述封装结构的内部，上晶圆的第一表面和下晶圆的第一表面平行相对设置，串联谐振器和并联谐振器通过对接管脚键合形成多级串并联的滤波器电路；其中，多个串联谐振器的压电层的厚度与并联谐振器的压电层的厚度不同，而且，并联谐振器的有效机电耦合系数大于串联谐振器的有效机电耦合系数，滤波器的谐波区域低频抑制幅度和高频抑制幅度相等且大于指定值。
可选地，所述滤波器电路还包括接地电感，接地电感的第一端连接并联谐振器，第二端接地；该接地电感的电感值小于预设值。
本发明的又一个方面，还提供了一种双工器，包括上述滤波器。
本发明的又一个方面，还提供了一种通信设备，包括上述滤波器。
附图说明
为了说明而非限制的目的，现在将根据本发明的优选实施例、特别是 参考附图来描述本发明，其中：
图1为滤波器中低频谐振器的阻抗曲线示意图；
图2为滤波器中两个谐振器的阻抗曲线示意图；
图3为滤波器的通带曲线示意图；
图4为谐振器阻抗曲线对比示意图；
图5为滤波器的通带曲线示意图；
图6为滤波器的通带曲线示意图；
图7为并联谐振器不同压电层厚度时的通带曲线对比示意图；
图8为本发明实施方式提供的滤波器带外抑制优化方法的流程示意图；
图9为滤波器的拓扑结构示意图；
图10为仿真滤波器的通带曲线示意图；
图11为滤波器优化后的通带曲线示意图；
图12为滤波器中并联谐振器连接接地电感后串联谐振频点的变化曲线对比示意图；
图13为并联谐振器连接接地电感后的通带曲线对比示意图；
图14为本发明实施方式提供的一种滤波器封装结构的剖面图；
图15为本发明实施方式提供的滤波器封装结构中上晶圆的主视图；
图16为本发明实施方式提供的滤波器封装结构中下晶圆的主视图。
具体实施方式
本发明实施方式中的技术方案，能够维持滤波器通带覆盖范围不变，以及能够解决低频体声波滤波器谐波抑制较差的问题，同时，还能保障谐波抑制区域的抑制平衡，以下具体加以说明。
图1为滤波器中低频谐振器的阻抗曲线示意图。该谐振器为典型的谐振器结构，即包括叠加的上电极、压电层和下电极，该曲线有2个谐振区域，即基频谐振区和谐波谐振区，基频谐振区频率较低，谐振在900MHz左右，包括串联谐振频点和并联谐振频点，其中，并联谐振频点的阻抗Rp为6500欧姆左右，而谐波谐振区频率较高，谐振在3000MHz左右，包括串联谐振频点和并联谐振频点，其中并联谐振频点的Rp为800欧姆， 具有较高的阻抗值。
图2为滤波器中两个谐振器的阻抗曲线示意图。如图2所示，图中实线为串联谐振器的阻抗曲线，该曲线与图1所示的阻抗曲线完全相同，而虚线为并联谐振器的阻抗曲线，该并联谐振器采用加载质量负载的方法实现移频，该曲线和串联谐振器的阻抗曲线类似，同样包括2个谐振区域，即基频谐振区和谐波谐振区，基频谐振区频率较低，谐振在865MHz左右，基频谐振区包括串联谐振频点和并联谐振频点，其中，并联谐振频点的阻抗Rp为6500欧姆左右，而谐波谐振区频率较高，谐振在2900MHz左右，谐波谐振区包括串联谐振频点和并联谐振频点，其中并联谐振频点的Rp为800欧姆，具有较高的阻抗值。对比两条曲线可知，并联谐振器基频的并联谐振频点位于串联谐振器基频的串联谐振频点附近，多个串并联谐振器组成的一个滤波器梯型拓扑结构，其会在基频形成一个通带。图3为滤波器的通带曲线示意图。如图3所示，该曲线中，在900MHz附近形成一个通带，在2900MHz附近形成一个伪通带，该伪通带的存在，恶化了这个频率附近的带外抑制，其产生的原因和基频类似，即串并联谐振器具有相同的叠层，只在并联谐振器上加载质量负载时，会使并联谐振器的谐波的并联谐振频点位于串联谐振器谐波的串联谐振频点附近，由此形成了伪通带，该伪通带的存在恶化了该频段的带外抑制，严重影响体声波滤波器在低频段的推广和使用，因此，需要对其进行改善。
为解决上述问题，可以采用如下方法：对滤波器的串联谐振器、并联谐振器分别采用不同的压电层厚度，并且并联谐振器的压电层的厚度要大于串联谐振器的压电层的厚度(即并联谐振器的有效机电耦合系数大于串联谐振器的有效机电耦合系数)，从而使在基频频段，并联谐振器基频的并联谐振频点位于串联谐振器基频的串联谐振频点附近，从而形成通带，而由于并联谐振器的压电层厚度大于串联谐振器的压电层的厚度，所以在谐波频段，并联谐振器的谐波谐振区域要高于串联谐振器的谐波谐振区域。图4为谐振器阻抗曲线对比示意图。如图4所示，实线为串联谐振器的阻抗曲线，该曲线包括2个谐振区域，即基频谐振区和谐波谐振区，而虚线 为并联谐振器的阻抗曲线，该并联谐振器的叠层与串联谐振器不同，并且并联谐振器的压电层厚度大于串联谐振器的压电层厚度，即利用压电层作为加载质量负载实现移频，该曲线和串联谐振器的阻抗曲线类似，同样包括2个谐振区域，即基频谐振区和谐波谐振区，并联谐振器基频的并联谐振频点位于串联谐振器基频的串联谐振频点附近，多个上述串并联谐振器组成的滤波器梯型拓扑结构，其会在基频形成一个通带。如图5所示，在谐波区域，如果并联谐振器谐波的串联谐振频点刚好位于串联谐振器谐波的并联谐振频点附近，则会在谐波区域形成一个类似阻带的区域，改善此区域的带外抑制。
上述方法中，串联谐振器和并联谐振器的压电层的设置方式为：首先设定串联谐振器的压电层厚度为一定值(即串联谐振器的有效机电耦合系数为定值)，然后优化并联谐振器压电层厚度(即并联谐振器的有效机电耦合系数)，从而达到改善谐波区域抑制的目的。该方法由于限定了串联谐振器压电层的厚度，其无法再兼顾基频通带插损、临带抑制以及谐波区域抑制，即如果并联谐振器压电层厚度较小，会导致并联谐振器谐波的串联谐振频点偏离串联谐振器谐波的并联谐振频点，并且靠近串联谐振器谐波的串联谐振频点，会导致谐波区域的低频段抑制恶化，图5中的圆圈1指示的位置抑制只有22dB，圆圈2指示的位置抑制可达37dB，这两个位置的抑制差16dB，两位置的抑制幅度不平衡；如果并联谐振器压电层厚度较大，会导致并联谐振器谐波的串联谐振频点偏离串联谐振器谐波的并联谐振频点，并且大于串联谐振器谐波的并联谐振频点，会导致谐波区域的高频段抑制恶化，图6中的圆圈1指示的位置抑制可达41dB，圆圈2指示的位置抑制只有21dB，这两个位置的抑制差20dB，两边不平衡。
图7为并联谐振器不同压电层厚度时的通带曲线对比示意图。如图7所示，虚线为并联谐振器压电层较薄时的曲线，实线为并联谐振器压电层较厚时的曲线，由图中所示的曲线对比可知，滤波器通带覆盖的范围是变化的，如果通带左右都有抑制的话，在优化过程中，滤波器通带覆盖范围存在变化，滤波器会因为通带变宽导致临带抑制变差，或者因为通带变窄， 导致边频插损恶化。
由此可知，以上方法对于串联谐振器和并联谐振器的压电层的厚度调节方式灵活性较差，而且无法兼顾基频通带插损、临带抑制以及谐波区域抑制。
本发明实施方式提供一种滤波器带外抑制优化方法，该方法在优化的过程中，既可以维持滤波器通带覆盖范围几乎恒定不变，还能解决低频体声波滤波器谐波抑制较差的问题，以及保障谐波抑制区域的抑制平衡。
图8为本发明实施方式提供的滤波器带外抑制优化方法的流程示意图。如图8所示，步骤S81：调整串联谐振器和并联谐振器的压电层的厚度，使两者压电层的厚度不同，进而使两者的有效机电耦合系数的初值不同，其中，两个初值应该满足以下两个条件：一、并联谐振器的有效机电耦合系数的初值大于串联谐振器的有效机电耦合系数的初值，二、两者之和为固定值；一般情况下，并联谐振器的有效机电耦合系数的初值比串联谐振器的有效机电耦合系数的初值大1％～2％，两者之和为滤波器相对带宽的4～5倍；步骤S82：判断并联谐振器谐波的串联谐振频点是否位于串联谐振器谐波的串联谐振频点与并联谐振频点之间，若是，则进行步骤S83，否则返回步骤S81；步骤S83：对滤波器拓扑结构进行仿真，判断滤波器的基频是否满足指标要求，若是，则进入步骤S84，否则返回步骤S81；步骤S84：判断谐波区域低频抑制幅度和高频抑制幅度是否相等，且大于指定值，指定值一般为30dB；若是，则优化结束，否则进入步骤S85；步骤S85：判断谐波区域低频抑制幅度是否大于高频抑制幅度；若是，则将并联谐振器的有效机电耦合系数的初值减小m％，以及将串联谐振器的有效机电耦合系数的初值增大m％后，返回步骤S83；否则将并联谐振器的有效机电耦合系数的初值增大n％，以及将串联谐振器的有效机电耦合系数的初值减小n％后，返回步骤S83。如此多次循环优化设计，直到既满足基频的指标要求，又满足谐波抑制要求，同时要保证谐波区域抑制的平衡性，以及谐波区域低频抑制幅度和高频抑制幅度基本相等，才设计完成。
以下通过具体实例验证上述方法的有效性。利用体声波谐振器设计一个低频滤波器，其频率范围覆盖880～915MHz，谐波抑制大于35dB。图9为滤波器的拓扑结构示意图。如图9所示，该拓扑结构为5-4结构(当然不限于5-4结构，可以是M-N结构，M和N为自然数，此处仅以5-4结构为例)，该拓扑结构包含1个串联支路和4个并联支路，串联支路由串联谐振器S11、S12、S13、S14和S15串接组成，串接于端口1和端口2之间，并联支路包括并联谐振器和接地电感，并联谐振器的一端与相邻的两个串联谐振器之间的节点连接，另一端与接地电感连接。其中，第一并联支路包括并联谐振器P11和接地电感L11，第二并联支路包括并联谐振器P12和接地电感L12，第三并联支路包括并联谐振器P13和接地电感L13，第四并联支路包括并联谐振器P14和接地电感L14。
为了使串联谐振器和并联谐振器的有效机电耦合系数不同，将全部串联谐振器制作在一片晶圆上，全部并联谐振器制作在另外一片晶圆上，按照上述的优化步骤：首先，选定串联谐振器的其压电层厚度初始值为0.65微米，有效机电耦合系数初始值为7％，并联谐振器的压电层厚度初始值为0.9微米，有效机电耦合系数初始值为9.3％，串联谐振器和并联谐振器的有效机电耦合系数之和为16.3％；进行串并联谐振器谐波的串并联谐振频点分析，确定以上选定的压电层厚度，使得并联谐振器谐波的串联谐振频点刚好位于串联谐振器谐波的串联谐振频点与并联谐振频点之间；然后可进行滤波器拓扑结构仿真优化，以上参数得到整个滤波器的通带插损小于1.8dB，其基本满足基频指标要求，可进行下一步操作，进行滤波器谐波抑制进行分析。图10为仿真滤波器的通带曲线示意图。由图10所示的曲线可知，谐波抑制最差点仅为25dB，并没有达到要求，同时发现谐波抑制最差点是谐波区的高频部分，谐波区的低频抑制部分较好，可达40dB。根据上述方法将并联谐振器的有效机电耦合系数减小0.5％，同时串联谐振器的有效机电耦合系数增大0.5％，并联谐振器有效机电耦合系数改为8.8％，其压电层改为0.87微米，串联谐振器有效机电耦合系数改为7.5％，其压电层为0.68微米，保持串联谐振器和并联谐振器有效机电耦合系数之 和为16.3％不变。更新有效机电耦合系数后，重新代入原设计，再进行仿真。图11为滤波器优化后的通带曲线示意图。如图11所示，基频插损满足指标要求，整个通带插损小于1.8dB，滤波器整个带外抑制都大于40dB，特别是谐波区域，抑制都大于40dB，谐波区域的低频抑制幅度和高频抑制幅度更加平衡。
滤波器中并联支路的接地电感对谐波抑制也起关键性作用，主要原因是并联谐振器串联一个接地电感后，会改变谐振器基频和谐波区域的串联谐振频点的位置，具体来说，一般会使串联谐振频点的位置向低频移动，所以当并联谐振器串联的电感值较大时，有可能会使并联谐振器谐波的串联谐振频率小于串联谐振器谐波的串联谐振频率，不再满足其必须位于串联谐振器谐波的串联谐振频点与并联谐振频点之间的要求，导致谐波区域低频部分的抑制恶化。图12为滤波器中并联谐振器连接接地电感后串联谐振频点的变化曲线对比示意图。如图12所示，串联谐振器的谐波谐振情况如图中所标的细实线，其谐波串联谐振频点在2.88GHz，并联谐振频点在2.93GHz，当并联谐振器接的电感为0.3nH时，其谐波串联谐振频点在2.925GHz，并联谐振频点在2.96GHz，此时并联谐振器谐波的串联谐振频点刚好位于串联谐振器谐波的串联谐振频点与并联谐振频点之间，随着串联电感的增大，其谐波串联谐振频点向低频移动，即当电感增大到0.5nH时，并联谐振器谐波的串联谐振频点移到了2.84GHz，不再位于串联谐振器谐波的串联谐振频点与并联谐振频点之间，所以会导致谐波区域低频段的抑制度恶化。图13为并联谐振器连接接地电感后的通带曲线对比示意图。如图13所示，图中实线是接地电感值为0nH时对应的曲线图，该曲线中谐波抑制较好，虚线是接地电感值为0.5nH时对应的曲线图，该曲线中谐波抑制恶化了15dB。
图14为本发明实施方式提供的一种滤波器封装结构的剖面图。如图14所示，该滤波器的封装结构中，将全部并联谐振器制作在上晶圆，全部串联谐振器制作在下晶圆。图15为本发明实施方式提供的滤波器封装结构中上晶圆的主视图；图16为本发明实施方式提供的滤波器封装结构中 下晶圆的主视图。如图15和图16所示，上晶圆包括并联谐振器P11、P12、P13和P14，以及接地管脚G1、G2、G3、G4和转接键合管脚J1、J2、J3、J4；下晶圆包括串联谐振器S11、S12、S13、S14和S15，以及接地管脚G1、G2、G3、G4，转接键合管脚J1、J2、J3、J4，输入管脚IN和输出管脚OUT。上晶圆和下晶圆上下叠加设置，转接键合管脚J1、J2、J3、J4键合、接地管脚G1、G2、G3、G4键合；其中，下晶圆上还设有过孔，通过过孔将上晶圆、下晶圆制造的滤波器的信号端和对地端连接到下晶圆下方的焊盘上，下晶圆下方的焊盘可以通过金属焊球连接到封装基板，从而形成封装结构。
该滤波器中，多个串联谐振器的压电层的厚度与并联谐振器的压电层的厚度不同，而且，并联谐振器的有效机电耦合系数大于串联谐振器的有效机电耦合系数，滤波器的谐波区域低频抑制幅度和高频抑制幅度相等且大于指定值，如大于30dB。由于串联谐振器和并联谐振器分设在两个晶圆上，因此，压电层可设置为不同的厚度，而且，该厚度易于调节。该滤波器不仅能够维持滤波器通带覆盖范围不变，而且能够解决低频体声波滤波器谐波抑制较差的问题，同时，还能保障谐波抑制区域的抑制平衡。
本发明实施方式还提供一种双工器，该双工器包括上述滤波器，因此，使得该双工器也能达到维持滤波器通带覆盖范围不变，能够解决低频体声波滤波器谐波抑制较差的问题，以及保障谐波抑制区域的抑制平衡的效果。
本发明实施方式还提供一种通信设备，该通信设备包括上述滤波器，因此，使得该通信设备也能达到维持滤波器通带覆盖范围不变，能够解决低频体声波滤波器谐波抑制较差的问题，以及保障谐波抑制区域的抑制平衡的效果。
上述具体实施方式，并不构成对本发明保护范围的限制。本领域技术人员应该明白的是，取决于设计要求和其他因素，可以发生各种各样的修改、组合、子组合和替代。任何在本发明的精神和原则之内所作的修改、 等同替换和改进等，均应包含在本发明保护范围之内。

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  1. 一种滤波器带外抑制优化方法，所述滤波器包括多个串联谐振器和多个并联谐振器，其特征在于，该方法包括：
    调整串联谐振器和并联谐振器的压电层的厚度，使并联谐振器的有效机电耦合系数大于串联谐振器的有效机电耦合系数的初值，并且使两个所述初值之和为固定值，以及使并联谐振器谐波的串联谐振频点位于串联谐振器谐波的串联谐振频点与串联谐振器谐波的并联谐振频点之间；
    在滤波器的基频满足指标要求并且滤波器的谐波区域低频抑制幅度和高频抑制幅度不相等的情况下，执行如下步骤A或步骤B，直至滤波器的谐波区域低频抑制幅度和高频抑制幅度相等且大于指定值，其中：
    步骤A：若滤波器的谐波区域低频抑制幅度大于高频抑制幅度，则减小并联谐振器的有效机电耦合系数的初值，增大串联谐振器的有效机电耦合系数的初值，并且保持两个所述初值之和为固定值；
    步骤B：若滤波器的谐波区域低频抑制幅度小于高频抑制幅度，则增大并联谐振器的有效机电耦合系数的初值，减小串联谐振器的有效机电耦合系数的初值，并且保持两个所述初值之和为固定值。
  2. 根据权利要求1所述的方法，其特征在于，所述滤波器中，每个并联谐振器均连有接地电感，所述接地电感的电感值小于预设值。
  3. 根据权利要求1所述的方法，其特征在于，调整串联谐振器和并联谐振器的压电层的厚度的步骤包括：
    将串联谐振器和并联谐振器制造在不同的晶圆上，分别调整两块晶圆上压电层的厚度，以使串联谐振器和并联谐振器的压电层的厚度不同。
  4. 根据权利要求1所述的方法，其特征在于，并联谐振器的有效机电耦合系数的初值比串联谐振器的有效机电耦合系数的初值大1％～2％，两者之和为滤波器相对带宽的4-5倍。
  5. 根据权利要求1所述的方法，其特征在于，所述指定值为30dB。
  6. 根据权利要求1所述的方法，其特征在于，所述步骤A或步骤B中，并联谐振器的有效机电耦合系数的初值和串联谐振器的有效机电耦合系数的初值增大或减小0.5％。
  7. 根据权利要求2所述的方法，其特征在于，预设值为0.5nH。
  8. 一种滤波器，其特征在于，包括上晶圆、下晶圆、多个串联谐振器和多个并联谐振器，全部并联谐振器设于上晶圆第一表面，全部串联谐振器设于下晶圆的第一表面；上晶圆和下晶圆叠加形成封装结构；
    在所述封装结构的内部，上晶圆的第一表面和下晶圆的第一表面平行相对设置，串联谐振器和并联谐振器通过对接管脚键合形成多级串并联的滤波器电路；
    其中，多个串联谐振器的压电层的厚度与并联谐振器的压电层的厚度不同，所述串联谐振器的压电层的厚度与并联谐振器的压电层的厚度采用权利要求1至7中任一项所述的方法调整得到，而且，并联谐振器的有效机电耦合系数大于串联谐振器的有效机电耦合系数，滤波器的谐波区域低频抑制幅度和高频抑制幅度相等且大于指定值。
  9. 根据权利要求8所述的滤波器，其特征在于，所述滤波器电路还包括接地电感，接地电感的第一端连接并联谐振器，第二端接地；
    该接地电感的电感值小于预设值。
  10. 一种双工器，其特征在于，包括如权利要求8或9所述的滤波器。
  11. 一种通信设备，其特征在于，包括如权利要求8或9所述的滤波器。
PCT/CN2021/114004 2020-08-24 2021-08-23 滤波器带外抑制优化方法和滤波器、多工器、通信设备 WO2022042463A1 (zh)

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116722837A (zh) * 2023-05-31 2023-09-08 锐石创芯(重庆)科技有限公司 体声波滤波器组件、射频前端模块及电子设备
WO2024207720A1 (zh) * 2023-04-06 2024-10-10 华南理工大学 一种体声波谐振器优化设计方法、装置及存储介质

Families Citing this family (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN112073028B (zh) * 2020-08-24 2021-08-10 诺思(天津)微系统有限责任公司 滤波器带外抑制优化方法和滤波器、多工器、通信设备
CN114567287A (zh) * 2022-03-21 2022-05-31 苏州汉天下电子有限公司 多工器

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2018093388A (ja) * 2016-12-05 2018-06-14 太陽誘電株式会社 フィルタおよびマルチプレクサ
CN111010143A (zh) * 2019-11-20 2020-04-14 天津大学 体声波滤波器及其制造方法以及双工器
CN111313862A (zh) * 2020-02-26 2020-06-19 诺思(天津)微系统有限责任公司 调整滤波器电路的方法和滤波器、多工器、通讯设备
CN111327296A (zh) * 2020-02-27 2020-06-23 诺思(天津)微系统有限责任公司 体声波滤波器元件及其形成方法、多工器及通讯设备
CN112073028A (zh) * 2020-08-24 2020-12-11 诺思(天津)微系统有限责任公司 滤波器带外抑制优化方法和滤波器、多工器、通信设备

Family Cites Families (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP6819789B2 (ja) * 2017-08-09 2021-01-27 株式会社村田製作所 弾性波装置、マルチプレクサ、高周波フロントエンド回路及び通信装置
KR102066958B1 (ko) * 2018-07-10 2020-01-16 삼성전기주식회사 필터

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2018093388A (ja) * 2016-12-05 2018-06-14 太陽誘電株式会社 フィルタおよびマルチプレクサ
CN111010143A (zh) * 2019-11-20 2020-04-14 天津大学 体声波滤波器及其制造方法以及双工器
CN111313862A (zh) * 2020-02-26 2020-06-19 诺思(天津)微系统有限责任公司 调整滤波器电路的方法和滤波器、多工器、通讯设备
CN111327296A (zh) * 2020-02-27 2020-06-23 诺思(天津)微系统有限责任公司 体声波滤波器元件及其形成方法、多工器及通讯设备
CN112073028A (zh) * 2020-08-24 2020-12-11 诺思(天津)微系统有限责任公司 滤波器带外抑制优化方法和滤波器、多工器、通信设备

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2024207720A1 (zh) * 2023-04-06 2024-10-10 华南理工大学 一种体声波谐振器优化设计方法、装置及存储介质
CN116722837A (zh) * 2023-05-31 2023-09-08 锐石创芯(重庆)科技有限公司 体声波滤波器组件、射频前端模块及电子设备
CN116722837B (zh) * 2023-05-31 2024-07-02 锐石创芯(重庆)科技有限公司 体声波滤波器组件、射频前端模块及电子设备

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