RU2503019C1 - Method to measure nominal frequency of sinusoidal signals and device for its realisation - Google Patents

Abstract

FIELD: measurement equipment.SUBSTANCE: method to measure nominal frequency of sinusoidal signals contemplates realisation of tuning of measured nominal frequency with a phase changer controlled by a sawtooth voltage generator. Tuning is carried out until equality of phases with a frequency that arrives directly to the second input of the comparator, the actuation time of which is proportionate to the number of pulses measured by a counter and processed by a microcontroller. At the same time the phase changer comprises RC-links, in which the role of capacitance C is played by capacitor diodes, and the microcontroller comprises a program that provides for the possibility to calibrate different types of sensors for linearisation of dependences of values of physical parameters on frequency. Results of measurements are displayed on an indicator. The device for measurement of nominal frequency of sinusoidal signals comprises a generator of reference frequency, a key, an "AND" circuit, a pulse counter, an indication unit, a microcontroller, the input of which is connected to the output of the pulse counter, and the output - with the indicator, a phase comparator, a gate multivibrator, which starts the generator of sawtooth voltage, which controls the phase changer until equality of phases on the comparator.EFFECT: provision of high reliability, accuracy of the method, efficiency and universality of application.2 cl, 3 dwg

Description

The invention relates to measuring equipment and automation and can be used to work with various converters of non-electric quantities to the frequency of sinusoidal signals in information-measuring devices for monitoring and control of technological processes and in other industries for precision measurement of frequency deviations from the nominal value in a certain frequency range.

A known digital frequency counter (Ornatsky P.P. Automatic measurements and instruments. Kiev: Vishcha school, 1981. P.329-330), containing a pulse shaper of a controlled frequency, a pulse frequency divider that controls the duration of a rectangular pulse generator, the output of which is fed to one from the control inputs of the key. The command for controlling the other key input is generated from the clock pulse generator after dividing it by one hundred and forming the duration of the rectangular pulses by the corresponding generator. The key transmits the clock pulses for the time difference between the periods of the control rectangular pulses in each measurement cycle with a digital counter showing the percentage deviation of the frequency of the nominal value. Such a technical solution allowed to increase the measurement performance by 10 times compared to using conventional frequency meters and to produce industrial devices F5035.

A known frequency meter of nominal values (Shlyandin V.M. Digital measuring devices. M .: Higher school, 1981. P.150-153), in which the pulses of the measured frequency after the amplifier-driver and filling the digital counter to a certain number corresponding to the nominal value, they control through the trigger the opening of the key for the time the ratio of the nominal frequency to the measured one. During the same time, the reversible counter from the initial reference value set before measurement, equal to twice the product of the nominal frequency by the ratio of the reference frequency to the nominal one, subtracts the reference frequency pulses from the pulse generator and generates readings proportional to the measured frequency.

The disadvantages of the analogues are the small measurement range, which is not suitable for working with various converters of non-electric quantities to frequency, and a large methodological measurement error.

The closest in technical essence is a method of measuring frequency deviations from the nominal value (A.S. 336612 USSR, MKI G01R 23/10 of 04/21/1972. Publish. 10/19/1972. Bull. No. 14), based on the calculation of the number of periods of the reference frequency during one period of the measured frequency, in decreasing total from the initial value proportional to the nominal value of the period of the signal of the measured frequency, in which, to simplify the measurement process, change the counter status indication to inverse when receiving zero and continue to count pulses in decreasing total until the end of the period.

Closest to the proposed device is a digital meter deviation of the measured frequency from the nominal (A. with. 300833 USSR, MKI G01R 17/00 from 04/07/1971. Publ. 10.06.1971. Bull. No. 13), containing the pulse generator of the measured frequency, keys , a reference frequency generator, a control unit and a pulse counter, in which a counter with a preset, direct and reverse code comparison schemes, a matching circuit, and an integrator with pos research transfer, the output of which is connected to the input of the counter with a preset, the output of the code of which is connected to the first inputs of the direct and reverse codes comparison circuit, the second inputs of which are connected to the outputs of the direct and reverse codes of the integrator control counter with serial transfer, the outputs of which are connected via the matching circuit to the input of the installation of the initial state of the counter with a preset.

The main significant disadvantage of the method of measuring frequency deviations from the nominal value and the digital meter deviations of the measured frequency from the nominal is the low speed and accuracy, complexity, a large number of operations for processing sinusoidal signals and the need to subtract from the current frequency value its initial value corresponding to the zero value of the measured parameter, which requires the inclusion of an additional device, entailing additional complication and lower reliability STI device.

The task of the invention is to improve the accuracy and speed of measuring parameters from analog sensors with a frequency output in physical units and to develop a device for its implementation using a minimum set of standard functional units, which will provide high reliability.

The problem is solved by implementing a method for measuring the nominal frequency of sinusoidal signals from sensors with a frequency output, according to which a phase shifter, electronically controlled by a sawtooth voltage generator, consisting of RC links, in which the role of capacitance C is performed by varicaps connected through an amplifier to the first input of the phase comparator, adjustment of the measured nominal frequency to phase equality with the frequency fed directly to the second input of the comparator, the response time of which is It is proportional to the number of pulses measured by a counter and processed by a microcontroller, the program of which is supplied with calibration characteristics of various types of sensors to linearize the dependences of the values of physical parameters on frequency, the result of which is fed to the indicator.

The problem is also solved by a device for measuring the nominal frequency of sinusoidal signals, containing a reference frequency generator, a key, an “I” circuit, a pulse counter and an indication unit, characterized in that it is equipped with a microcontroller, the input of which is connected to the output of the pulse counter, and the output - an indicator, a phase comparator, at the first input of which a sinusoidal signal from the output of the sensor comes through an electronically controlled phase shifter and amplifier, and to the second - it comes directly from the sensor from frequencies output, a single-shot, starting a sawtooth voltage generator, which controls the phase shifter until the phases are equal on the comparator.

In addition, the essence of the technical solutions is illustrated by drawings, where:

- figure 1 presents the Converter circuit three-pole structure;

- figure 2 is a schematic diagram of an electronically controlled phase shifter on varicaps;

- figure 3 is a structural diagram of a frequency meter of nominal values.

Essence: the method is implemented using phase-locked loop of the nominal frequency (FANCH) of the analog signal using an electronically controlled phase shifter (EUV), which increases accuracy, because there is no frequency mismatch between the measured and balanced signals at the time of measurement, and speed, and also eliminates the methodological error of measurement (see Radio receivers / Ed. by A.P. Zhukovsky. - M .: Higher school, 1989. P.195). To expand the measurement range, as a EUV, a three-pole chain structure (CTS) is used, consisting of n / 2 RC links, where the role of capacitors C is performed by varicaps, practically inertialess elements (see Novitsky P.V., Knorring V.G. , Gutnikov BC Digital Devices with Frequency Sensors (Leningrad: Energy, 1970, p. 80). The measurement of the rated frequency without intermediate conversions greatly simplifies the circuit, which increases the reliability of the device.

Known traditional research methods do not allow obtaining analytical expressions that relate the measurement range, the quasi-resonance frequency, the signal attenuation from the number n / 2 RC - EUV units, especially those consisting of nonlinear elements (varicaps) and, therefore, solve the actual problem.

Using the method of transformation functions (FP) allowed to eliminate this gap (see Gulin A.I. Diagnostics of measuring transducers and communication devices with an inhomogeneous chain structure // Control. Diagnostics. 2010. No. 11. P.69-72).

FP K _n converter of a three-pole chain structure (DTS), formally, the reciprocal of the traditional transmission coefficient (Figure 1), which is the ratio of the input active quantity U ₀ to output V _n (voltage U _n or current I _n ) is described by the expression for an even number of arms n

$$K_n=\mathrm{one}+{\displaystyle \sum _{\begin{array}{l}i=\mathrm{one}\\ k=i+\mathrm{one}\end{array}}^{\begin{array}{l}n\\ n-\mathrm{one}\end{array}}{Z}_i{Y}_k+{\displaystyle \sum _{\begin{array}{l}i=\mathrm{one}\\ k=i+\mathrm{one}\end{array}}^{\begin{array}{l}n-2\\ n-3\end{array}}{\displaystyle \sum _{\begin{array}{l}p=k+\mathrm{one}\\ q=p+\mathrm{one}\end{array}}^{\begin{array}{l}n\\ n-\mathrm{one}\end{array}}{Z}_i{Y}_k{Z}_p{Y}_q+...,(\mathrm{one})}}}$$

where i = 2b-1;

b = 1,2,3, ..., 0,5n,

and for chain structures (CS) with an odd number of shoulders n

$$K_n={\displaystyle \sum _{i=\mathrm{one}}^n{Z}_i+{\displaystyle \sum _{\begin{array}{l}i=\mathrm{one,}\\ k=i+\mathrm{one}\end{array}}^{\begin{array}{l}n-\mathrm{one}\\ n-2\end{array}}{\displaystyle \sum _{p=k+\mathrm{one}}^n{Z}_i{Y}_k{Z}_p+...,(2)}}}$$

where b = 1,2,3, ..., 0,5 (n + 1) for a CA with an odd number of shoulders n.

Relations (1) and (2) lead to a recurrence formula for calculating the phase transition

K _n = T _n K _n-1 + K _n-2 ,

where T _{i is the} immitance of the i-th arm (resistance Z for odd i and conductivity Y for even i).

The initial conditions of the calculation algorithm K _n are the values K ₀ = 1 for n = 0 and K ₁ = T ₁ for n = 1.

The recommended electrical circuit of the EUV is shown in Figure 2, in which the required minimum number of capacitors (varicaps) must be at least three (see Gulin A.I. Design of multi-link regenerators // Izv. Universities "Instrument Engineering" 2012. V.15. No. 1 (41). S.14-118). Consider, for example, a six-armed PZT, the expression of which for which, according to (1),

K ₆ = 1 + Z ₁ Y ₂ + Z ₁ Y ₄ + Z ₁ Y ₆ + Z ₃ Y ₄ + Z ₃ Y ₆ + Z ₅ Y ₆ + Z ₁ Y ₂ Z ₃ Y ₄ +

+ Z ₁ Y ₂ Z ₃ Y ₆ + Z ₁ Y ₂ Z ₅ Y ₆ + Z ₁ Y ₄ Z ₅ Y ₆ + Z ₃ Y ₄ Z ₅ Y ₆ + Z ₁ Y ₂ Z ₃ Y ₄ Z ₅ Y ₆ .

For EUV (Figure 2), where Z ₁ = Z ₃ = Z ₅ = R, and Y ₂ = Y ₄ = Y ₆ = jωC, the AF will be

K ₆ = -jω ³ C ³ R ³ -5ωC ² R ² + j6ωCR + 1.

To determine the attenuation introduced by the EUV and the upper frequency of the initial tuning of its range (quasi-resonance frequency), we write the phase transition in the components of the real and imaginary parts, which has the form

K ₆ = ReK ₆ + Im K ₆ .

From the imaginary part of the phase transition Im K ₆ , equating it to zero, we determine the upper frequency of the initial tuning □ ₀ at which the EUV carries out a phase shift of 180 °, i.e.

Im K ₆ = -yω ³ ₀ С ³ R ³ + j6ω ₀ CR = 0,

where from $${\omega }_0=\frac{\sqrt{6}}{RC}(3)$$

From the real part of the FP ReK ₆ , substituting the value □ ₀ from (3), we determine the amount of attenuation introduced by the EUV, and which should be compensated by the amplifier (Figure 3)

Re K ₆ = -5ω ₀ C ² R ² + 1 = -29.

A minus sign indicates a phase rotation of the EUV 180 °. A single-stage amplifier with a gain of approximately 29 also rotates the phase of the measured signal by 180 °, realizing phase equality on the comparator at the time of measuring the nominal frequency.

The calculations for calculating the frequencies of quasi-resonances for an arbitrary number of links n / 2 are, as it turned out, reduced to determining the coefficient k _n from the expression

$${\omega }_0=\frac{k_n}{RC}.(\mathrm{four})$$

As a result of the analytical analysis, the formula was first obtained, which determines the coefficient k _n for PZT from any number of RC links from equations of the form

$${\displaystyle \sum _{i=0.1...}^P{(-\mathrm{one})}^i{k}_n^{2i+\mathrm{one}}{C}_{0.5n+\mathrm{one}+2\mathrm{<figure-callout\; id="2"\; label="i"\; filenames="00000015.png"\; state="\{\{state\}\}">i</figure-callout>}}^{2+\mathrm{four}i}=0(5)}$$

where p = 0.25n-1 - for even 0.5n;

p = 0.25 (n + 2) -1 - for odd 0.5n.

), так как использование других значений, удовлетворяющих условию (5), приведет к сдвигу фаз на 2π радиан и более.Of all the real positive roots of equation (5), it is necessary to use the smallest value (for six shoulders - PZT it is equal to $$\sqrt{6}$$

), since the use of other values satisfying condition (5) will lead to a phase shift of 2π radians or more.

The table shows the values of the conversion functions at the frequency of the quasi-resonance ReK _n and the coefficients k _{n of} PZT for the number of arms n from 6 to 40.

Table The values of the conversion functions at the frequency of the quasi-resonance ReK _n and the coefficients k _n of the number of shoulders n of the DTT n k _n Re _n 6 2,446 29th 8 1,195 24.70 10 0.739 23.46 12 0.509 22.77 fourteen 0.373 22.26 16 0.286 21.55 eighteen 0.227 20.58 twenty 0.185 20.11 22 0.153 19.57 24 0.129 18.93 26 0.11 18.48 28 0,095 17.91 thirty 0,083 17.44 32 0,073 17.11 34 0,065 16.76 36 0.058 16.51 38 0,052 16.29 40 0,047 16.09

To calculate more complex DTCs, you can use the program (see Gulin A.I., Sukhinets Zh.A. et al. Calculation of the frequency of quasi-resonance and transmission coefficient of multi-link RC structures // Certificate of official registration of a computer program No. 2003611147 / 05.16.2003 Rospatent. Moscow. 2003).

.It should be noted that the phase transition Re K _n at the quasi-resonance frequencies with increasing the number of arms n from six to infinity decreases and tends from Re K ₆ = -29 to Re K _n = -11.6, i.e. $$\underset{n\to \infty }{\lim }\left|K_n\right|=11.6$$

.

The use of varicaps as voltage-controlled sensitive inertia-free capacitors in phase shifters of a PZT type RC (Fig. 2) allows to obtain qualitatively and quantitatively new control characteristics not achieved in such schemes with linear elements, namely, an increase in the control ranges in automatic frequency and phase adjustment systems . The nature of the change in the dependence C = f (U) is determined by the design dimensions and technological features of the semiconductor. If we maintain the value of the control voltage (bias) on the capacitance 4–5 times the amplitude of the high-frequency oscillations, we can assume that the capacitance will mainly be determined only by the bias voltage. And since the inverse resistance of the transition is more than 1 MΩ, then practically the bias voltage on all varicaps is the same in view of the negligibly small current distribution along the vertical arms - the conductivities. High resistance R _{R is} necessary to prevent the input signal from being shunted by the control voltage source. The capacity of a varicap (see Berman L.S. Introduction to the physics of varicaps. - L .: Nauka, 1968. P.30) is determined from the expression

, $$C=C_{\mathrm{AT}}{(\frac{U_{\mathrm{at}PR}+{\varphi }_k}{U_{\mathrm{AT}}})}^{-b}$$

,

where C _In , U _In - the capacitance and bias voltage of the varicap, corresponding to the upper tuning frequency;

U _control - voltage control bias on varicaps;

φ _k is the contact potential difference of the pn junction, lying within 0.4 ÷ 0.7 V;

b is a coefficient depending on the distribution of impurities in the transition, equal to 0.5 for varicaps with a sharp p-n junction.

Therefore, the expression (4) when using varicaps will take the form

$$f_0=\frac{{\mathrm{<figure-callout\; id="2"\; label="k\; n"\; filenames="00000015.png"\; state="\{\{state\}\}">k\; </figure-callout>}}_n}{2\pi R{C}_{\mathrm{AT}}}\sqrt{\frac{U_{\mathrm{AT}}}{U_{\mathrm{at}PR}+{\varphi }_k}}$$

If the frequency f _{x the} input measured voltage U _{0 is} not equal to the frequency of the quasi-resonance of the phase shifter f ₀ , then the phase angle is

$$\varphi =\pi \sqrt{f/f}$$

For EUV (Figure 2), in which the phase change at the output is achieved by changing the value of the capacitance of varicaps With voltage U _CPR , from GPN and therefore by changing the value of f ₀ , the expression for the phase angle will be

$$\phi =\mathrm{<figure-callout\; id="2"\; label="\pi "\; filenames="00000015.png"\; state="\{\{state\}\}">\pi </figure-callout>}\sqrt{\frac{2\pi RA{f}_x}{{(U_{\mathrm{at}PR}+{\phi }_k)}^b{k}_n}},(5)$$

where k _n is determined from the expression (4);

A is the coefficient of proportionality, depending on the concentration of impurities and the area pn of the junction of the semiconductor device.

Knowing the range of variation of the output frequency of the sensor Δf equal to

$$\Delta f=f_{\max }-f_{\min }=\frac{{\mathrm{<figure-callout\; id="2"\; label="k\; n"\; filenames="00000015.png"\; state="\{\{state\}\}">k\; </figure-callout>}}_n}{2\pi R{C}_{\mathrm{AT}}}-\frac{{\mathrm{<figure-callout\; id="2"\; label="k\; n"\; filenames="00000015.png"\; state="\{\{state\}\}">k\; </figure-callout>}}_n}{2\pi R{C}_{\max }}=\frac{{\mathrm{<figure-callout\; id="2"\; label="k\; n"\; filenames="00000015.png"\; state="\{\{state\}\}">k\; </figure-callout>}}_n}{2\pi R}\times \frac{C_{\max }-C_{\mathrm{AT}}}{C_{\mathrm{AT}}{C}_{\max }},$$

where C _max - the maximum capacity of the varicap corresponding to the lower frequency of the EUV tuning, we obtain the expression for determining the coefficient k _n

$$k_n=2\pi R\Delta f\frac{C_{\mathrm{AT}}{C}_{\max }}{C_{\max }-C_{\mathrm{AT}}}$$

In the table we find the corresponding value of the coefficient k _n , by which we determine the number of links (varicaps) of the EUV that meets the measurement range, and the corresponding value of the gain Re K _n for the amplifier. In case of mismatch of the calculated coefficient with the table value, select the nearest lower value of k _n and Re K _n .

A device for measuring the nominal frequency of sinusoidal signals contains (Figure 3) an electronically controlled phase shifter 1 of n / 2 RC - links in which the role of capacitance C is performed by varicaps, connected through amplifier 2 to the first input of the phase comparator 3, the second input of which is measured the frequency goes directly, and the output of the comparator via key 4 is connected to the first input of the sawtooth generator 5 (GPN), the second input of which is connected to a single vibrator 6, and the output of the GPN 5 is connected to the control input of the phase shifter 1. Key output 4 is also connected to the first input of AND element 7, the second input of which is connected to a reference frequency generator (GOCH) 8, and the output through the counter 9 and microcontroller 10 is connected to a digital readout device 11.

The microcontroller program is provided with calibration characteristics of various types of sensors.

The measurement of the physical parameter from sensors with an analog frequency output using the proposed device is as follows. A sinusoidal signal from the output of the sensor is supplied through an electronically controlled phase shifter 1 and amplifier 2 to the first input of the phase comparator 3, to the second input of which it directly goes. When the frequency meter is turned on, the one-shot 6 starts the sawtooth voltage generator 5 (GPN), controlling the phase shifter 1 until the phases are equal on the comparator 3, issuing commands through the key 4 to the GPN, stopping its further change, and to the And 7 element, blocking the passage of pulses from the generator reference frequency 8 to the counter 9, the number of which is functionally proportional to the measured physical parameter. The microcontroller 10 linearizes the dependence of the value of the physical parameter on the frequency, which is displayed on the digital reading device 11 in units of the measured value. The microcontroller programmatically provides for setting calibration values of various types of sensors.

So, the claimed invention allows directly without additional transformations to measure physical parameters using various sensors with a frequency analog output, which ensures high reliability, accuracy of the method, speed and versatility of use.

Claims (2)

1. A method of measuring the nominal frequency of sinusoidal signals from sensors with a frequency output, characterized in that the phase shifter, electronically controlled by a sawtooth generator, consisting of RC links, in which the role of capacitance C is performed by varicaps, connected through an amplifier to the first input of the phase comparator, adjustment of the measured nominal frequency to phase equality with the frequency supplied directly to the second input of the comparator, the response time of which is proportional to the number of pulses measured tchikom and processed by the microcontroller, a program which is provided with a calibration characteristics of different types of sensors for physical parameters linearization dependencies of frequency values, the result of which is fed to the indicator. 2. A device for measuring the nominal frequency of sinusoidal signals, containing a reference frequency generator, a key, an “I” circuit, a pulse counter and an indication unit, characterized in that it is equipped with a microcontroller, the input of which is connected to the output of the pulse counter, and the output to the indicator, a phase comparator, at the first input of which a sinusoidal signal from the sensor output is fed through an electronically controlled phase shifter and amplifier, and to the second it comes directly from the sensor with a frequency output, single-shot, start-up a sawtooth voltage generator that controls the phase shifter until the phases are equal on the comparator.

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