[go: up one dir, main page]
More Web Proxy on the site http://driver.im/
You seem to have javascript disabled. Please note that many of the page functionalities won't work as expected without javascript enabled.
 
 
Sign in to use this feature.

Years

Between: -

Subjects

remove_circle_outline
remove_circle_outline
remove_circle_outline

Journals

Article Types

Countries / Regions

Search Results (6)

Search Parameters:
Keywords = sharp roll-off

Order results
Result details
Results per page
Select all
Export citation of selected articles as:
12 pages, 5292 KiB  
Article
A Wide Passband Frequency-Selective Surface with a Sharp Roll-Off Band Using the Filtering Antenna-Filtering Antenna Method
by Yanfei Ren, Zhenghu Xi, Qinqin Liu, Jiayi Gong, Zhiwei Sun and Boyu Sima
Materials 2024, 17(24), 6131; https://doi.org/10.3390/ma17246131 - 15 Dec 2024
Viewed by 470
Abstract
Frequency-selective surfaces (FSSs) have attracted great attention owing to their unique feature to manipulate transmission performance over the frequency domain. In this work, a filtering antenna-filtering antenna (FA-FA) FSS with a wide passband and double-side sharp roll-off characteristics is presented by inter-using the [...] Read more.
Frequency-selective surfaces (FSSs) have attracted great attention owing to their unique feature to manipulate transmission performance over the frequency domain. In this work, a filtering antenna-filtering antenna (FA-FA) FSS with a wide passband and double-side sharp roll-off characteristics is presented by inter-using the filtering antenna and receiving–transmitting metasurface methods. First, a dual-polarized filtering antenna element was designed by employing a parasitic band-stop structure with an L-probe feed. Then, the FA-FA-based FSS unit was constructed by placing two such filtering antennas back to back, with their feedings connected through metallic vias. Finally, the FSS with a wide passband and high selectivity was realized by arraying the FA-FA units periodically. The full-wave simulation results demonstrated that the designed FA-FA-based FSS had a wide passband from 13.06 GHz to 14.46 GHz with a flat in-band frequency response. The lower and upper roll-off bandwidths were sharp, reaching 1% and 1.2% of the center frequency. The proposed FA-FA-based FSS was fabricated and measured, achieving the coincident performance according to the theoretical prediction. The wideband band-pass FSS obtained a sharp double-side roll-off feature, which can be applied in various studies such as an antenna array, metasurface, communication, etc. Full article
Show Figures

Figure 1

Figure 1
<p>Schematic design of (<b>a</b>) the AFA-based FSS and (<b>b</b>) the FA-FA based FSS.</p>
Full article ">Figure 2
<p>L-shaped probe feed patch antenna: (<b>a</b>) schematic structure and (<b>b</b>) equivalent circuit diagram.</p>
Full article ">Figure 3
<p>Three-dimensional view of (<b>a</b>) the filtering antenna and (<b>b</b>) FA-FA-based FSS unit.</p>
Full article ">Figure 4
<p>Unit structure of FA-FA-based FSS: (<b>a</b>) top view of first layer, (<b>b</b>) second layer, (<b>c</b>) fourth layer, and (<b>d</b>) side view.</p>
Full article ">Figure 5
<p>Simulated magnitude of the transmission of the proposed FA-FA-based FSS unit under normal incidence.</p>
Full article ">Figure 6
<p>(<b>a</b>) Partial structure of the FSS and (<b>b</b>) the corresponding simplified equivalent transmission line model.</p>
Full article ">Figure 6 Cont.
<p>(<b>a</b>) Partial structure of the FSS and (<b>b</b>) the corresponding simplified equivalent transmission line model.</p>
Full article ">Figure 7
<p>S-parameters for different incidence angles under (<b>a</b>) TE polarization and (<b>b</b>) TM polarization.</p>
Full article ">Figure 8
<p>(<b>a</b>) The 3D diagram of the reconfigurable AFA FSS unit structure and (<b>b</b>) the top view of the second layer.</p>
Full article ">Figure 9
<p>Simulated magnitude of the transmission of the proposed reconfigurable AFA FSS unit under normal incidence.</p>
Full article ">Figure 10
<p>The fabrication photograph of the proposed FA-FA-based FSS prototype. (<b>a</b>) Front view of the upper plate. (<b>b</b>) Front and partial enlarged view of the middle plate. (<b>c</b>) Front view of the lower plate. (<b>d</b>) Front and partial enlarged view of the assembled prototype. (<b>e</b>) Side view of the assembled prototype. (<b>f</b>) Back view of the assembled prototype.</p>
Full article ">Figure 11
<p>Measurement setup: (<b>a</b>) prototype in the measurement window and (<b>b</b>) the whole test environment.</p>
Full article ">Figure 12
<p>Comparison of the simulated and measured transmission results of the proposed FSS: (<b>a</b>) TE and (<b>b</b>) TM.</p>
Full article ">
13 pages, 6986 KiB  
Communication
Integrated Broadband Filter with Sharp Transition Edges Based on SiN and SiON Composite Waveguide Coupler
by Xiao Ma, Qiongchan Shao, Jiamei Gu, Tingting Lang, Xiang Guo and Jian-Jun He
Photonics 2023, 10(11), 1285; https://doi.org/10.3390/photonics10111285 - 20 Nov 2023
Cited by 1 | Viewed by 1499
Abstract
Broadband filters with sharp transition edges are important elements in diverse applications, including Raman and fluorescence spectral analysis, wideband wavelength-division multiplexing (WDM), and multi-octave interferometry. While the multi-layer thin-film interference broadband filter has been widely applied in various free-space optical systems, its integrated [...] Read more.
Broadband filters with sharp transition edges are important elements in diverse applications, including Raman and fluorescence spectral analysis, wideband wavelength-division multiplexing (WDM), and multi-octave interferometry. While the multi-layer thin-film interference broadband filter has been widely applied in various free-space optical systems, its integrated counterpart is still far from mature, which is also highly desired for constructing chip-scale miniature optical modules. In this article, we design, fabricate, and characterize an integrated broadband filter with sharp transition edges. An adiabatic coupler based on silicon nitride (SiN) and silicon oxynitride (SiON) composite waveguide is employed here. Long-pass, short-pass, band-pass, and band-stop filters can be realized in a single design of the composite waveguide coupler for a specific wavelength range, with only a difference in the SiN taper waveguide width. Experimental results with a roll-off value of larger than 0.7 dB nm−1 and a 15 dB extinction ratio (ER) are presented. Full article
Show Figures

Figure 1

Figure 1
<p>The structural diagrams of single-core SiN waveguide (top left), single-core SiON waveguide (bottom left), three-segment SiN waveguide (top right) and five-segment SiN waveguide (bottom right).</p>
Full article ">Figure 2
<p>Dispersion curves of single-core SiN waveguide (1.2 μm × 0.25 μm), single-core SiON waveguide (3 μm × 1.8 μm) and (<b>a</b>) 60–90% duty cycle 3-segment SiN waveguides; (<b>b</b>) 60–90% duty cycle 5-segment SiN waveguides.</p>
Full article ">Figure 3
<p>(<b>a</b>) Three-dimensional schematic diagram and (<b>b</b>) top view of the broadband filter with an adiabatic coupler.</p>
Full article ">Figure 4
<p>Dispersion curves of SiON and SiN waveguides with different widths.</p>
Full article ">Figure 5
<p>Supermode filed distributions for the SiON and SiN composite waveguide with different SiN widths at a specific wavelength. In the figure, the above is a SiON waveguide with a width of 3 μm and a height of 1.8 μm. The below is a SiN waveguide with a height of 0.25 μm.</p>
Full article ">Figure 6
<p>Transmittance spectra of the (<b>a</b>) SiN waveguide and (<b>b</b>) SiON waveguide with different W<sub>11</sub> when light is injected into SiN waveguide. In all situations, W<sub>12</sub> equals W<sub>11</sub> + 0.2 μm.</p>
Full article ">Figure 7
<p>Simulated transmittance spectra of the adiabatic coupler when the SiN taper widths range from 0.9 μm to 2.1 μm. Light is injected from the SiN waveguide.</p>
Full article ">Figure 8
<p>(<b>a</b>) The relationship between ER and taper length L<sub>1</sub> when R<sub>2</sub> is set to 500,000 μm and (<b>b</b>) the relationship between ER and R<sub>2</sub> when L<sub>1</sub> is set to 2000 μm. W<sub>11</sub> and W<sub>12</sub> of the taper waveguide are 1 μm and 1.2 μm, respectively, and the gap between the two waveguides is 0.9 μm. The red diamond is the measured discrete points and the blue dotted line is the connecting line between them.</p>
Full article ">Figure 9
<p>Transmittance spectra of the SiN and SiON waveguides in the 785 nm band.</p>
Full article ">Figure 10
<p>Flow process diagram of the entire fabrication process.</p>
Full article ">Figure 11
<p>Cross-section of the composite waveguides in the approaching area after SiON etching process.</p>
Full article ">Figure 12
<p>Relationship between the refractive index of the developed SiON and the N<sub>2</sub>O flux when the SiH<sub>4</sub> and NH<sub>3</sub> fluxes is kept as 17.5 sccm and 190 sccm, respectively.</p>
Full article ">Figure 13
<p>Micrographs of the SiN waveguide after SiO<sub>2</sub> and SiON layers have been deposited when (<b>a</b>) most of the SiN film on the wafer has been etched; (<b>b</b>) only SiN film near the waveguide has been etched.</p>
Full article ">Figure 14
<p>Micrograph of the fabricated filter array with different parameters.</p>
Full article ">Figure 15
<p>SEM images of the adiabatic coupler at different positions after SiON etching.</p>
Full article ">Figure 16
<p>Normalized transmittance spectra in the output ends of the SiN and SiON waveguides in two adjacent filters.</p>
Full article ">
16 pages, 5093 KiB  
Article
Design and Modelling of a Compact Triband Passband Filter for GPS, WiMAX, and Satellite Applications with Multiple Transmission Zero’s
by Abdul Basit, Amil Daraz, Muhammad Irshad Khan, Farid Zubir, Salman A. AlQahtani and Guoqiang Zhang
Fractal Fract. 2023, 7(7), 511; https://doi.org/10.3390/fractalfract7070511 - 28 Jun 2023
Cited by 5 | Viewed by 1781
Abstract
Designing microwave filters with high selectivity and sharp roll-off between the stop and pass bands can be challenging due to the complex nature of the R.F. signals and the requirements for achieving high performance in a limited physical space. To achieve a high [...] Read more.
Designing microwave filters with high selectivity and sharp roll-off between the stop and pass bands can be challenging due to the complex nature of the R.F. signals and the requirements for achieving high performance in a limited physical space. To achieve a high selectivity and sharp roll-off rate, this paper presents a compact filter with a triple passband response. The two different passbands at 1.57 GHz and 3.5 GHz are achieved using a step impedance resonator (SIR) with metallic slots perturbation added to the lower corner of the high impedance section of the SIRs, which helps to enhance the filter’s selectivity and size reduction greatly. The embedded L-shaped structure originates a third passband at 4.23 GHz, resulting in a triband response with eight transmission zeros below and above the passbands at 1.22/1.42/1.98/3.18/3.82/3.98/4.38/4.53 GHz, respectively. The prototype has low signal attenuation of <1.2 dB and high signal reflection of >25 dB for the three passbands. The fractional bandwidths achieved are 2.54%, 4.2%, and 1.65% at 1.57/3.57/4.23 GHz, respectively, with rejection levels in the stopband greater than 15 dB. Lastly, the structure is fabricated on RO-4350B PCB and observed good matching between experimental and measured results. This demonstrates that the prototype can be successfully implemented in real-world applications such as GPS, WiMAX, and Satellite systems. The area occupied by the filter on a substrate or in a circuit is 0.31 λg × 0.24 λg, where λg is the guided wavelength of the material calculated at the lowest frequency. Full article
(This article belongs to the Special Issue Advances in Fractal Antennas: Design, Modeling and Applications)
Show Figures

Figure 1

Figure 1
<p>Placement of bandpass filter in communication systems [<a href="#B6-fractalfract-07-00511" class="html-bibr">6</a>].</p>
Full article ">Figure 2
<p>Configuration of resonators (<b>a</b>) Stepped impedance resonator (<b>b</b>) dual band filter topology (<b>c</b>) Equivalent structure of (<b>b</b>).</p>
Full article ">Figure 3
<p>(<b>a</b>) Proposed triband geometry. (<b>b</b>) Equivalent circuit model.</p>
Full article ">Figure 4
<p>Variations in cut-off frequencies w.r.t electrical length θ<sub>1</sub>.</p>
Full article ">Figure 5
<p>Variations in cut-off frequencies w.r.t electrical length θ<sub>4</sub>.</p>
Full article ">Figure 6
<p>Realization of the third passband by embedded L-shaped resonator.</p>
Full article ">Figure 7
<p>Response of filter without L-shaped structure.</p>
Full article ">Figure 8
<p>Variation in third pass band w.r.t stub length L<sub>4</sub>.</p>
Full article ">Figure 9
<p>Location of T.Z.s at different frequencies with sharpness factor.</p>
Full article ">Figure 10
<p>Effect of G parameter coupling coefficient on three passbands.</p>
Full article ">Figure 11
<p>Effect of G parameter external quality factor on three passbands.</p>
Full article ">Figure 12
<p>Effect of S parameter coupling coefficient on three passbands.</p>
Full article ">Figure 13
<p>Effect of T<sub>2</sub> parameter external quality factor on three passbands.</p>
Full article ">Figure 14
<p>Group delay response of the three passbands.</p>
Full article ">Figure 15
<p>Nature of the current distribution of the triband filter (<b>a</b>) surface current density due to the frequency ratio K. (<b>b</b>) surface current density due to L-shaped resonator at 4.23 GHz.</p>
Full article ">Figure 16
<p>The magnitude of the electric field intensity (<b>a</b>) E due to the frequency ratio K. (<b>b</b>) E due to the L-shaped resonator at 4.23 GHz.</p>
Full article ">Figure 17
<p>Simulated and measured S<sub>11</sub> and S<sub>21</sub> plots of the proposed geometry.</p>
Full article ">
14 pages, 5775 KiB  
Article
Role of Surface Topography in the Superhydrophobic Effect—Experimental and Numerical Studies
by Samih Haj Ibrahim, Tomasz Wejrzanowski, Bartłomiej Przybyszewski, Rafał Kozera, Xabier García-Casas and Angel Barranco
Materials 2022, 15(9), 3112; https://doi.org/10.3390/ma15093112 - 25 Apr 2022
Cited by 12 | Viewed by 2208
Abstract
Within these studies, the effect of surface topography for hydrophobic coatings was studied both numerically and experimentally. Chemically modified polyurethane coating was patterned by application of a laser beam. A set of patterns with variously distant linear peaks and grooves was obtained. The [...] Read more.
Within these studies, the effect of surface topography for hydrophobic coatings was studied both numerically and experimentally. Chemically modified polyurethane coating was patterned by application of a laser beam. A set of patterns with variously distant linear peaks and grooves was obtained. The cross section of the pattern showed that the edges of the peaks and grooves were not sharp, instead forming a rounded, rectangle-like shape. For such surfaces, experimental studies were performed, and in particular the static contact angle (SCA), contact angle hysteresis (CAH), and roll-off angle (ROA) were measured. Profilometry was used to create a numerical representation of the surface. Finite volume method was then applied to simulate the behavior of the water droplets. The model developed herewith enabled us to reproduce the experimental results with good accuracy. Based on the verified model, the calculation was extended to study the behavior of the water droplet on the simulated patterns, both spiked and rectangular. These two cases, despite a similar SCA of the water droplet, have shown extremely different ROA. Thus, more detailed studies were dedicated to other geometrical features of such topography, such as the size and distance of the surface elements. Based on the results obtained herewith, the future design of superhydrophobic and/or icephobic topography is discussed. Full article
(This article belongs to the Section Materials Simulation and Design)
Show Figures

Figure 1

Figure 1
<p>SEM images of modified polyurethane coatings after laser patterning: (<b>a</b>,<b>c</b>) sample ‘5by5’, (<b>b</b>,<b>d</b>) sample ‘20by20’.</p>
Full article ">Figure 2
<p>Example of the generated mesh for further calculations.</p>
Full article ">Figure 3
<p>Volume fraction of fluid (red—water, blue—air) presenting: (<b>a</b>) Initial droplet configuration and (<b>b</b>) droplet after stabilization.</p>
Full article ">Figure 4
<p>Measurement of contact angle: (<b>a</b>) initial volume fraction image (red—water, blue—air), (<b>b</b>) grayscale volume fraction image (black—water, white—air), and (<b>c</b>) contact angle measured with drop analysis plugin.</p>
Full article ">Figure 5
<p>Volume fraction of the fluid in the grooves: blue—water and red—air. Visible air pockets trapped in the grooves.</p>
Full article ">Figure 6
<p>Volume fraction of the fluid in the grooves: blue—air and red—water. (<b>a</b>) Water phase does not reach the next bulge on circular pattern, (<b>b</b>) Water-air interface with the spiked pattern.</p>
Full article ">
19 pages, 6988 KiB  
Article
Estimation and Analysis of Higher-Order Harmonics in Advanced Integrated Circuits to Implement Noise-Free Future-Generation Micro- and Nanoelectromechanical Systems
by Muhammad Imran Khan, Ahmed S. Alshammari, Badr M. Alshammari and Ahmed A. Alzamil
Micromachines 2021, 12(5), 541; https://doi.org/10.3390/mi12050541 - 10 May 2021
Cited by 5 | Viewed by 2986
Abstract
This work deals with the analysis of spectrum generation from advanced integrated circuits in order to better understand how to suppress the generation of high harmonics, especially in a given frequency band, to design and implement noise-free systems. At higher frequencies, the spectral [...] Read more.
This work deals with the analysis of spectrum generation from advanced integrated circuits in order to better understand how to suppress the generation of high harmonics, especially in a given frequency band, to design and implement noise-free systems. At higher frequencies, the spectral components of signals with sharp edges contain more energy. However, current closed-form expressions have become increasingly unwieldy to compute higher-order harmonics. The study of spectrum generation provides an insight into suppressing higher-order harmonics (10th order and above), especially in a given frequency band. In this work, we discussed the influence of transistor model quality and input signal on estimates of the harmonic contents of switching waveforms. Accurate estimates of harmonic contents are essential in the design of highly integrated micro- and nanoelectromechanical systems. This paper provides a comparative analysis of various flip-flop/latch topologies on different process technologies, i.e., 130 and 65 nm. An FFT plot of the simulated results signifies that the steeper the spectrum roll-off, the lesser the content of higher-order harmonics. Furthermore, the results of the comparison illustrate the improvement in the rise time, fall time, clock-Q delay and spectrum roll-off on the better selection of slow-changing input signals and more accurate transistor models. Full article
(This article belongs to the Special Issue MEMS Devices for Nanomanufacturing)
Show Figures

Figure 1

Figure 1
<p>Verilog-A code for input signal generation.</p>
Full article ">Figure 2
<p>Inverter output plot (one cycle) in MATLAB.</p>
Full article ">Figure 3
<p>Spectrum plot of the inverter switching waveform.</p>
Full article ">Figure 4
<p>Output plot of the inverter in MATLAB: the first cycle of the waveform contains 5000 equidistant data points.</p>
Full article ">Figure 5
<p>Inverter output plots (one cycle) in MATLAB.</p>
Full article ">Figure 6
<p>Spectrum plot of the inverter’s switching waveform.</p>
Full article ">Figure 7
<p>Schematic of the PowerPC 603 master–slave latch.</p>
Full article ">Figure 8
<p>Schematic of the modified C²MOS latch.</p>
Full article ">Figure 9
<p>Schematic of the hybrid-latch flip-flop (HLFF).</p>
Full article ">Figure 10
<p>Spectrum roll-off of HLFF using the 65 nm process.</p>
Full article ">Figure 11
<p>Spectrum roll-off of HLFF using the 130 nm process.</p>
Full article ">Figure 12
<p>Spectrum roll-off analysis using 65 nm process technology and various types of input signals.</p>
Full article ">Figure 13
<p>Spectrum roll-off analysis using 130 nm process technology and various types of input signals.</p>
Full article ">Figure 14
<p>Spectrum roll-off analysis using 65 and 130 nm process design kits and arctan(sin) as the input signal.</p>
Full article ">Figure 15
<p>Total power dissipation in the PowerPC 603 master–slave latch.</p>
Full article ">Figure 16
<p>Total power dissipation in the modified C²MOS latch.</p>
Full article ">Figure 17
<p>Total power dissipation in the hybrid-latch flip-flop (HLFF).</p>
Full article ">Figure 18
<p>Total power dissipation analysis using 65 nm process technology and various types of input signals.</p>
Full article ">Figure 19
<p>Total power dissipation analysis using 130 nm process technology and various types of input signals.</p>
Full article ">Figure 20
<p>Power Analysis of various flip-flop/latch topologies using 65 and 130 nm process technologies.</p>
Full article ">Figure 21
<p>Power delay product (PDP).</p>
Full article ">
15 pages, 14514 KiB  
Article
Compact Differential-Fed Planar Filtering Antennas
by Emadeldeen Hassan, Denys Martynenko, Eddie Wadbro, Gunter Fischer and Martin Berggren
Electronics 2019, 8(11), 1241; https://doi.org/10.3390/electronics8111241 - 30 Oct 2019
Cited by 3 | Viewed by 3662
Abstract
This paper proposes novel low-profile differential-fed planar antennas with embedded sharp frequency selectively. The antennas are compact and easy to integrate with differential devices without matching baluns. The antenna design is formulated as a topology optimization problem, where requirements on impedance bandwidth, directivity, [...] Read more.
This paper proposes novel low-profile differential-fed planar antennas with embedded sharp frequency selectively. The antennas are compact and easy to integrate with differential devices without matching baluns. The antenna design is formulated as a topology optimization problem, where requirements on impedance bandwidth, directivity, and filtering are used as the design objectives. The optimized antennas operate over the frequency band 6.0–8.5 GHz. The antennas have reflection coefficients below −15 dB, cross-polarization levels below −42 dB, a maximum gain of 6.0 ± 0.5 dB, and a uniform directivity over more than 130° beamwidth angle in the frequency band of interest. In addition, the antennas exhibit sharp roll-off between the operational band and frequencies around the 5.8 GHz WiFi band and the 10 GHz X-band. One antenna has been fabricated with a good match between simulation and measurement results. Full article
(This article belongs to the Section Microwave and Wireless Communications)
Show Figures

Figure 1

Figure 1
<p>The design domain <math display="inline"><semantics> <mrow> <mo>Ω</mo> <mo>=</mo> <msub> <mo>Ω</mo> <mn>1</mn> </msub> <mo>∪</mo> <msub> <mo>Ω</mo> <mn>2</mn> </msub> </mrow> </semantics></math> is connected to a source <math display="inline"><semantics> <msub> <mi>V</mi> <mi>s</mi> </msub> </semantics></math>, with an internal impedance <math display="inline"><semantics> <msub> <mi>Z</mi> <mi>s</mi> </msub> </semantics></math>, through a differential-fed strip line at the center of the left side. Each of the domains <math display="inline"><semantics> <msub> <mo>Ω</mo> <mn>1</mn> </msub> </semantics></math> and <math display="inline"><semantics> <msub> <mo>Ω</mo> <mn>2</mn> </msub> </semantics></math> has an area <math display="inline"><semantics> <mrow> <mi>h</mi> <mo>×</mo> <mi>l</mi> </mrow> </semantics></math> where a metallic material (copper) is to be distributed to achieve the design objectives.</p>
Full article ">Figure 2
<p>Development of the physical design of the reference antenna over iterations 1, 34, 68, and 136. Black color indicates a good conductor and white color indicates a good dielectric.</p>
Full article ">Figure 3
<p>Reference antenna: (<b>a</b>) The shape of the antenna optimized to radiate over the frequency band 6.0–8.5 GHz. (<b>b</b>) Reflection coefficient of the antenna. The operational band is marked by the green dashed lines.</p>
Full article ">Figure 4
<p>Reference antenna: (<b>a</b>) Realized gain in the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math>). (<b>b</b>) Realized gain in the <math display="inline"><semantics> <mrow> <mi>y</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>). The maximum cross-polarization level, not plotted here for brevity, is less than 57 dB and 45 dB for <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>, respectively.</p>
Full article ">Figure 5
<p>Surface current distribution on the reference antenna.</p>
Full article ">Figure 6
<p>Development of the physical design of Antenna II over iterations 1, 34, 68, and 129. Black color indicates a good conductor and white color indicates a good dielectric.</p>
Full article ">Figure 7
<p>Antenna II: (<b>a</b>) The shape of the antenna optimized to radiate over the frequency band 6.0–8.5 GHz and to filter the frequencies 3.5–6.0 GHz and 8.5–11.0 GHz. (<b>b</b>) Photo of a manufactured prototype. (<b>c</b>) The measured and computed antenna reflection coefficient. The operational band is marked by the green dashed lines.</p>
Full article ">Figure 8
<p>Antenna II: Simulated (<b>a</b>) and measured (<b>b</b>) realized gain in the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math>). Simulated (<b>c</b>) and measured (<b>d</b>) realized gain in the <math display="inline"><semantics> <mrow> <mi>y</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>). The maximum simulated cross-polarization level, not plotted here for brevity, is less than 54 dB and 42 dB for <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>, respectively. (<b>e</b>) Setup of the antenna gain measurements.</p>
Full article ">Figure 9
<p>Surface current distribution on Antenna II.</p>
Full article ">Figure 10
<p>Antenna III: (<b>a</b>) The shape of the antenna optimized to radiate over the frequency band 6.0–8.5 GHz and to filter the frequencies 3.5–6.0 GHz and 8.5–11.0 GHz. (<b>b</b>) The antenna reflection coefficient. The operational band is marked by the green dashed lines.</p>
Full article ">Figure 11
<p>Antenna III (<b>a</b>) Realized gain in the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math>). (<b>b</b>) Realized gain in the <math display="inline"><semantics> <mrow> <mi>y</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>). The maximum cross-polarization level is less than 49 dB and 41 dB for <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>, respectively.</p>
Full article ">Figure 12
<p>Surface current distribution on Antenna III.</p>
Full article ">Figure 13
<p>Antenna IV: (<b>a</b>) The shape of the antenna optimized to radiate over the frequency band 6.0–8.5 GHz and to filter the frequencies 3.5–6.0 GHz and 8.5–11.0 GHz. (<b>b</b>) The antenna reflection coefficient. The operational band is marked by the green dashed lines.</p>
Full article ">Figure 14
<p>Antenna IV: (<b>a</b>) Realized gain in the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math>). (<b>b</b>) Realized gain in the <math display="inline"><semantics> <mrow> <mi>y</mi> <mi>z</mi> </mrow> </semantics></math> plane (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>). The maximum cross-polarization level is less than 52 dB and 44 dB for <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>0</mn> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <mn>90</mn> </mrow> </semantics></math>, respectively.</p>
Full article ">Figure 15
<p>Surface current distribution on Antenna IV.</p>
Full article ">
Back to TopTop