[go: up one dir, main page]
More Web Proxy on the site http://driver.im/
You seem to have javascript disabled. Please note that many of the page functionalities won't work as expected without javascript enabled.
 
 
Sign in to use this feature.

Years

Between: -

Subjects

remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline

Journals

Article Types

Countries / Regions

remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline
remove_circle_outline

Search Results (606)

Search Parameters:
Keywords = 3D antenna array

Order results
Result details
Results per page
Select all
Export citation of selected articles as:
19 pages, 3943 KiB  
Article
Quad-Beam 4 × 2 Array Antenna for Millimeter-Wave 5G Applications
by Parveez Shariff Bhadravathi Ghouse, Tanweer Ali, Pallavi R. Mane, Sameena Pathan, Sudheesh Puthenveettil Gopi, Bal S. Virdee, Jaume Anguera and Prashant M. Prabhu
Electronics 2025, 14(5), 1056; https://doi.org/10.3390/electronics14051056 - 6 Mar 2025
Viewed by 97
Abstract
This article presents the design of a novel, compact, 4 × 2 planar-array antenna that provides quad-beam radiation in the broadside direction, and it enhances coverage and serviceability for millimeter-wave applications. The antenna utilizes a corporate (parallel) feed network to deliver equal power [...] Read more.
This article presents the design of a novel, compact, 4 × 2 planar-array antenna that provides quad-beam radiation in the broadside direction, and it enhances coverage and serviceability for millimeter-wave applications. The antenna utilizes a corporate (parallel) feed network to deliver equal power and phase to all elements. Non-uniform element spacing in the two orthogonal planes, exceeding 0.5λ1 (λ1 being the wavelength at 30 GHz), results in a quad-beam radiation pattern. Two beams are formed in the xz-plane and two in the yz-plane, oriented at angles of θ=±54°. However, this spacing leads to null radiation at the center and splits the radiation energy, reducing the overall gain. The measured half-power beamwidth (HPBW) is 30° in the xz-plane and 35° in the yz-plane, with X-polarization levels of −20.5 dB and −26 dB, respectively. The antenna achieves a bandwidth of 28.5–31.1 GHz and a peak gain of 10.6 dBi. Furthermore, increasing the aperture size enhances the gain and narrows the beamwidth by replicating the structure and tuning the feed network. These features make the proposed antenna suitable for 5G wireless communication systems. Full article
Show Figures

Figure 1

Figure 1
<p>Initial design of the antenna (Ant1) (<b>a</b>), and its surface current representation (<b>b</b>).</p>
Full article ">Figure 2
<p>Final design of the single-element antenna (Ant1). The dimensions in mm are: EL = 7.3, Eh1 = 3.74, Ev1 = 1.7, Eh2 = 3, Ev2 = 1, Fl1 = 1.8, Fw1 = 0.4, Fl2 = 1, Fw2 = 0.78, SW1 = 8, and SL1 = 7.</p>
Full article ">Figure 3
<p>Surface current distribution of the final single-element antenna (Ant1) in (<b>a</b>), and (<b>b</b>) represents its reflection coefficient of a comparison of the initial elliptical antenna and the final elliptical ring (figure-eight) antenna.</p>
Full article ">Figure 4
<p>(<b>a</b>) The 3D radiation pattern and peak gain of Ant1, and (<b>b</b>) a 2D representation with co- and X-polarization in the xz- and yz-planes.</p>
Full article ">Figure 5
<p>A Poynting vector representation of Ant1.</p>
Full article ">Figure 6
<p>Modeling of the single-element antenna with the transmission line theory.</p>
Full article ">Figure 7
<p>(<b>a</b>) Comparison of the reflection coefficient |S11| obtained from the electromagnetic solver and the circuit simulator. (<b>b</b>) The impedance comparison of the numerical method, electromagnetic solver, and circuit simulator.</p>
Full article ">Figure 8
<p>Effect on the reflection coefficient of the single-element antenna from the variations in the major (<span class="html-italic">Eh</span>1) and minor (<span class="html-italic">Ev</span>1) axes of an elliptical patch in (<b>a</b>) and (<b>b</b>), respectively. Likewise, the variations in the resonance frequency are due to variations in the major (<span class="html-italic">Eh</span>2) and minor (<span class="html-italic">Ev</span>2) axes of the elliptical slot in (<b>c</b>) and (<b>d</b>), respectively.</p>
Full article ">Figure 9
<p>(<b>a</b>) The <math display="inline"><semantics> <mrow> <mn>2</mn> <mo>×</mo> <mn>1</mn> </mrow> </semantics></math> array antenna (Ant2) with elements separated by a distance of Es1 = 6 mm, where SW2 = 8 mm and SL2 = 10 mm. (<b>b</b>) A pictorial representation of the wave formation by the <math display="inline"><semantics> <mrow> <mn>2</mn> <mo>×</mo> <mn>1</mn> </mrow> </semantics></math> Ant2 due to the Es1 spacing.</p>
Full article ">Figure 10
<p>(<b>a</b>) The resulting 3D radiation pattern of Ant2. (<b>b</b>) A 2D representation of the <math display="inline"><semantics> <mrow> <mn>2</mn> <mo>×</mo> <mn>1</mn> </mrow> </semantics></math> array (Ant2) antenna.</p>
Full article ">Figure 11
<p>The <math display="inline"><semantics> <mrow> <mn>2</mn> <mo>×</mo> <mn>2</mn> </mrow> </semantics></math> array (Ant3) with Es1 = 6 mm, Es2 = 10 mm, SW3 = 18 mm, and SL3 = 10 mm.</p>
Full article ">Figure 12
<p>A demonstration of the 3D radiation pattern formation of Ant3 (<b>a</b>) and its respective 2D pattern (<b>b</b>).</p>
Full article ">Figure 13
<p>The proposed <math display="inline"><semantics> <mrow> <mn>4</mn> <mo>×</mo> <mn>2</mn> </mrow> </semantics></math> array antenna (Ant4), with Es3 = 4 mm, SL4 = 20 mm, and SW4 = 18 mm.</p>
Full article ">Figure 14
<p>Comparison of the 3D radiation obtained from (<b>a</b>) the equation, and (<b>b</b>) the simulated radiation pattern from Ant4 at 30.2 GHz.</p>
Full article ">Figure 15
<p>The simulated radiation pattern of Ant4 representing the co- and X-polarizations in (<b>a</b>) xz-plane and (<b>b</b>) yz-plane.</p>
Full article ">Figure 16
<p>(<b>a</b>) A comparison of the proposed Ant4 reflection coefficient |S11| with its previous antenna structures. (<b>b</b>) The simulated gain and total efficiency of Ant4.</p>
Full article ">Figure 17
<p>(<b>a</b>) Prototype fabrication of the Ant4 antenna. (<b>b</b>) Measurement setup of the |S-parameter| and radiation pattern in the anechoic chamber.</p>
Full article ">Figure 18
<p>The simulated and measured reflection coefficient |S11| of the proposed Ant4 antenna.</p>
Full article ">Figure 19
<p>The simulated and measured radiation patterns of Ant4 at 30 GHz and 30.2 GHz. (<b>a</b>) Pattern in the xz-plane and (<b>b</b>) in the yz-plane.</p>
Full article ">Figure 20
<p>Plot of the gain and total efficiency of Ant1 compared with Ant4.</p>
Full article ">Figure 21
<p>Figure-of-merit plot comparing the proposed antenna Ant4 performance with other planar arrays.</p>
Full article ">
10 pages, 2974 KiB  
Article
A New Observation in Decoupling and Sequential Rotation Array Configurations Using Loop Radiation Elements
by Kazuhide Hirose, Koki Nishino and Hisamatsu Nakano
J 2025, 8(1), 9; https://doi.org/10.3390/j8010009 - 1 Mar 2025
Viewed by 259
Abstract
Using the method of moments, we analyze three array antennas for low cross-polarized radiation. Each antenna comprises two dual-loop elements connected to a feedline horizontal to the ground plane. First, a feedline end is excited with an unbalanced source as a reference antenna. [...] Read more.
Using the method of moments, we analyze three array antennas for low cross-polarized radiation. Each antenna comprises two dual-loop elements connected to a feedline horizontal to the ground plane. First, a feedline end is excited with an unbalanced source as a reference antenna. Next, the feedline center is excited with a balanced source, after the transformation of a decoupling array configuration. It is found that the antenna exhibits a cross-polarized radiation lower by 12 dB than the reference antenna. Last, the decoupling antenna is modified to have an unbalanced source without a complicated balun circuit design. It is pointed out that the modified antenna is an array of four loop elements, sequentially rotated by 90º. Full article
Show Figures

Figure 1

Figure 1
<p>Reference antenna with a horizontal feedline <span class="html-italic">F</span>–<span class="html-italic">T</span>, excited at one end <span class="html-italic">F</span> via a vertical wire <span class="html-italic">F</span>–<span class="html-italic">F</span>′. (<b>a</b>) Perspective view. (<b>b</b>) Top view. (<b>c</b>) Side view.</p>
Full article ">Figure 2
<p>Present antenna with a horizontal feedline <span class="html-italic">F</span>–<span class="html-italic">T</span> excited at the center <span class="html-italic">C</span>. (<b>a</b>) Perspective view. (<b>b</b>) Top view. (<b>c</b>) Side view.</p>
Full article ">Figure 3
<p>Modified antenna with a horizontal feedline <span class="html-italic">F</span>–<span class="html-italic">T</span> excited at one end <span class="html-italic">F</span> via a vertical wire <span class="html-italic">F</span>–<span class="html-italic">F</span>′. (<b>a</b>) Perspective view. (<b>b</b>) Top view. (<b>c</b>) Side view.</p>
Full article ">Figure 4
<p>Simulated radiation patterns of a reference antenna at <span class="html-italic">f</span><sub>0</sub>.</p>
Full article ">Figure 5
<p>Simulated radiation patterns of the present antenna at <span class="html-italic">f</span><sub>0</sub>.</p>
Full article ">Figure 6
<p>Simulated frequency responses of axial ratio and gain of present and reference antennas.</p>
Full article ">Figure 7
<p>Simulated radiation patterns of the modified antenna at <span class="html-italic">f</span><sub>0</sub>.</p>
Full article ">Figure 8
<p>Simulated frequency responses of the axial ratio, gain, and VSWR of the modified antenna.</p>
Full article ">Figure 9
<p>Photographs of a prototype for a modified antenna. (<b>a</b>) Perspective view. (<b>b</b>) Side view.</p>
Full article ">Figure 10
<p>Radiation patterns of a modified antenna at <span class="html-italic">f</span><sub>0</sub>. (<b>a</b>) simulated results. (<b>b</b>) experimental results.</p>
Full article ">Figure 11
<p>Frequency responses of axial ratio, gain, and VSWR of a modified antenna.</p>
Full article ">
26 pages, 3719 KiB  
Article
Design of Multi-Sourced MIMO Multiband Hybrid Wireless RF-Perovskite Photovoltaic Energy Harvesting Subsystems for IoTs Applications in Smart Cities
by Fanuel Elias, Sunday Ekpo, Stephen Alabi, Mfonobong Uko, Sunday Enahoro, Muhammad Ijaz, Helen Ji, Rahul Unnikrishnan and Nurudeen Olasunkanmi
Technologies 2025, 13(3), 92; https://doi.org/10.3390/technologies13030092 - 1 Mar 2025
Viewed by 451
Abstract
Energy harvesting technology allows Internet of Things (IoT) devices to be powered continuously without needing battery charging or replacement. In addressing existing and emerging massive IoT energy supply challenges, this paper presents the design of multi-sourced multiple input and multiple output (MIMO) multiband [...] Read more.
Energy harvesting technology allows Internet of Things (IoT) devices to be powered continuously without needing battery charging or replacement. In addressing existing and emerging massive IoT energy supply challenges, this paper presents the design of multi-sourced multiple input and multiple output (MIMO) multiband hybrid wireless RF-perovskite photovoltaic energy harvesting subsystems for IoT application. The research findings evaluate the efficiency and power output of different RF configurations (1 to 16 antennas) within MIMO RF subsystems. A Delon quadruple rectifier in the RF energy harvesting system demonstrates a system-level power conversion efficiency of 51%. The research also explores the I-V and P-V characteristics of the adopted perovskite tandem cell. This results in an impressive array capable of producing 6.4 V and generating a maximum power of 650 mW. For the first time, the combined mathematical modelling of the system architecture is presented. The achieved efficiency of the combined system is 90% (for 8 MIMO) and 98% (for 16 MIMO) at 0 dBm input RF power. This novel study holds great promise for next-generation 5G/6G smart IoT passive electronics. Additionally, it establishes the hybrid RF-perovskite energy harvester as a promising, compact, and eco-friendly solution for efficiently powering IoT devices in smart cities. This work contributes to the development of sustainable, scalable, and smart energy solutions for IoT integration into smart city infrastructures. Full article
Show Figures

Graphical abstract

Graphical abstract
Full article ">Figure 1
<p>Common types of ambient energy harvesting.</p>
Full article ">Figure 2
<p>RF energy harvesting block diagram.</p>
Full article ">Figure 3
<p>Proposed RF−perovskite multi-source energy harvesting block diagram.</p>
Full article ">Figure 4
<p>MIMO system in RF energy harvesting.</p>
Full article ">Figure 5
<p>Schematic diagram of common PSC architectures: 2-terminal (<b>A</b>) and 4-terminal (<b>B</b>).</p>
Full article ">Figure 6
<p>Delon quadruple rectifier used in the RF energy harvester.</p>
Full article ">Figure 7
<p>Single−diode PV cell equivalent circuit.</p>
Full article ">Figure 8
<p>PSC array equivalent circuit.</p>
Full article ">Figure 9
<p>I-V and P-V characteristic curve of the perovskite-on-Si tandem solar cell used on this study.</p>
Full article ">Figure 10
<p>The proposed rectifier’s output voltage measured at the node (Vdc), shown in <a href="#technologies-13-00092-f006" class="html-fig">Figure 6</a>, and current at different input RF power levels.</p>
Full article ">Figure 11
<p>The rectifier’s output power measured at the node (Vdc) (refer to <a href="#technologies-13-00092-f006" class="html-fig">Figure 6</a>) and efficiency for different input RF power levels.</p>
Full article ">Figure 12
<p>The output voltage of a single–antenna RF energy harvester across various loads and diverse levels of RF input power.</p>
Full article ">Figure 13
<p>The efficiency of a single–antenna RF energy harvester under different loads and RF input power levels.</p>
Full article ">Figure 14
<p>The MIMO RF-EH output voltage at various RF input power levels.</p>
Full article ">Figure 15
<p>The output power of the MIMO RF-EH at different RF input power.</p>
Full article ">Figure 16
<p>The MIMO RF-EH output current at varying input RF power levels.</p>
Full article ">Figure 17
<p>Efficiency of the MIMO of RF-EH at different levels of input RF power.</p>
Full article ">Figure 18
<p>I-V and P-V characteristic curve of the perovskite-on-Si tandem solar cell with ADS-based simulation.</p>
Full article ">Figure 19
<p>I-V and P-V characteristic curve of the perovskite-on-Si tandem solar array with MATLAB simulation.</p>
Full article ">Figure 20
<p>I-V and P-V characteristic curve of the perovskite-on-Si tandem solar array with ADS-based simulation.</p>
Full article ">Figure 21
<p>The power output of hybrid RF-PSC configurations varies across different levels of RF input power, particularly at the peak power point of the PSC array under irradiation of 1000 W/m<sup>2</sup>.</p>
Full article ">
16 pages, 5587 KiB  
Article
Flat Emission Silicon Nitride Grating Couplers for Lidar Optical Antennas
by Thenia Prousalidi, Georgios Syriopoulos, Evrydiki Kyriazi, Roel Botter, Charalampos Zervos, Giannis Poulopoulos and Dimitrios Apostolopoulos
Photonics 2025, 12(3), 214; https://doi.org/10.3390/photonics12030214 - 28 Feb 2025
Viewed by 247
Abstract
Light detection and ranging (Lidar) is a key enabling technology for autonomous vehicles and drones. Its emerging implementations are based on photonic integrated circuits (PICs) and optical phased arrays (OPAs). In this work, we introduce a novel approach to the design of OPA [...] Read more.
Light detection and ranging (Lidar) is a key enabling technology for autonomous vehicles and drones. Its emerging implementations are based on photonic integrated circuits (PICs) and optical phased arrays (OPAs). In this work, we introduce a novel approach to the design of OPA Lidar antennas based on Si3N4 grating couplers. The well-established TriPleX platform and the asymmetric double stripe waveguide geometry with full etching are employed, ensuring low complexity and simple fabrication combined with the low-loss advantages of the platform. The design study aims to optimize the performance of the grating coupler-based radiators as well as the OPA, thus enhancing the overall capabilities of Si3N4-based Lidar. Uniform and non-uniform grating structures are considered, achieving θ and φ angle divergences of 0.9° and 32° and 0.54° and 25.41°, respectively. Also, wavelength sensitivity of 7°/100 nm is achieved. Lastly, the fundamental OPA parameters are investigated, and 35 dBi of peak directivity is achieved for an eight-element OPA. Full article
Show Figures

Figure 1

Figure 1
<p>Schematic of an optical antenna in OPA configuration based on grating couplers. The <span class="html-italic">θ</span> and <span class="html-italic">φ</span> angles and the distance d between adjacent GC elements are noted.</p>
Full article ">Figure 2
<p>(<b>a</b>) The OPA schematic to indicate the cross-sectional planes. (<b>b</b>) Schematic of the yz-plane cross section of the standard TriPleX ADS waveguide. The different regions (Si<sub>3</sub>N<sub>4</sub> waveguide, SiO<sub>2</sub> top oxide layer (TOX) and bottom oxide layer (BOX) and air top cladding) are marked with different colors. (<b>c</b>) Schematic of the sideview (xz-plane cross section) of a periodic grating structure. The grating pitch is denoted with <span class="html-italic">Λ</span> and the filling factor with <span class="html-italic">FF</span>. The effective index of the etched part is n<sub>0</sub> and of the unetched part is n<sub>1</sub>.</p>
Full article ">Figure 3
<p>Sideview of the uniform GC, showing the constant pitch, <span class="html-italic">FF</span> and width across the direction of propagation.</p>
Full article ">Figure 4
<p>Simulated <span class="html-italic">θ</span> and <span class="html-italic">φ</span> angle divergences, (<b>a</b>) varying the grating width for a fixed length of 50 μm and (<b>b</b>) varying the grating length for a fixed width of 2 μm.</p>
Full article ">Figure 5
<p>(<b>Left</b>): Top view of the emission profile of the uniform grating for a width of 2 μm and a length of 100 μm. The Ez component field distribution is shown with a color scale. (<b>Right)</b>: A 1D plot of the emission profile along the dashed line of the left figure.</p>
Full article ">Figure 6
<p>Calculated emission angle <span class="html-italic">θ</span> of the far-field profile, varying the wavelength in the range of 1.5–1.6 μm for a uniform tooth profile and a pitch of 926 nm. The electric field intensity is shown with a color scale.</p>
Full article ">Figure 7
<p>(<b>Left</b>) Top view and (<b>Right</b>) sideview of the investigated non-uniform grating design with a varying width and <span class="html-italic">FF</span>.</p>
Full article ">Figure 8
<p>Calculated effective refractive index of the TE<sub>0</sub> mode varying the waveguide width.</p>
Full article ">Figure 9
<p><span class="html-italic">n<sub>eff−grating</sub></span> of the fundamental supported mode calculated via FDE simulations, varying the waveguide width and <span class="html-italic">FF</span>. The black lines are the contour lines of the plots along which the <span class="html-italic">n<sub>eff−grating</sub></span> has a constant value. The selected contour line is marked with pink stars.</p>
Full article ">Figure 10
<p>The calculated coupling constant <span class="html-italic">k</span> for the different width values across the grating for three of the contour lines.</p>
Full article ">Figure 11
<p>Simulated <span class="html-italic">θ</span> and <span class="html-italic">φ</span> angle divergences varying the grating length for width–<span class="html-italic">FF</span> pairs calculated from the same contour line.</p>
Full article ">Figure 12
<p>(<b>Left</b>): Top view of the emission profile of the non-uniform grating for the selected geometrical parameters. The Ez component field distribution is shown with a color scale. (<b>Right</b>): A 1D plot of the emission profile along the dashed line of the left figure.</p>
Full article ">Figure 13
<p>Calculated emission angle <span class="html-italic">θ</span> of the far-field profile, varying the wavelength in the range of 1.5–1.6 μm, for the non-uniform tooth profile and a pitch of 926 nm. The electric field intensity is shown with a color scale.</p>
Full article ">Figure 14
<p>3D directivity plots (in dBi) produced with the Sensor Array Analyzer app, varying the number (<span class="html-italic">NA</span>) of grating elements and the distance (<span class="html-italic">d</span>) between them. The axis information is mentioned in the first subplot and is the same for all the subplots. The colorbar shows the directivity in dBi.</p>
Full article ">Figure 15
<p>Elevation cut for azimuth angle = 0° for two of the directivity plots of <a href="#photonics-12-00214-f014" class="html-fig">Figure 14</a>. On the other hand, the divergence of the <span class="html-italic">φ</span> angle is affected by the OPA topology, both by the number of elements <span class="html-italic">NA</span> and by their distance <span class="html-italic">d</span>. Increasing the <span class="html-italic">NA</span> reduces the <span class="html-italic">φ</span> divergence of the main lobe. Also, increasing the distance between adjacent elements reduces the main lobe <span class="html-italic">φ</span> divergence. It can also be seen in <a href="#photonics-12-00214-f014" class="html-fig">Figure 14</a> that increasing the number of elements (for a fixed distance) will result in the appearance of more side lobes with lower peak directivities, while the width of the main lobe is reduced. A similar effect is observed when the number of elements is kept constant and their distance is increased. In this case, more side lobes appear, and their peak directivity is also increased. Lastly, from the directivity plots, the peak directivity value can be extracted. This is 34.8 dBi for <span class="html-italic">NA</span> = 4 and <span class="html-italic">d</span> = 1.5<span class="html-italic">λ</span> and 35 dBi for <span class="html-italic">NA</span> = 8 and <span class="html-italic">d</span> = 1.5<span class="html-italic">λ</span>.</p>
Full article ">Figure 16
<p>Calculated <span class="html-italic">L</span><sub>10</sub> varying the distance d between two adjacent waveguides, with widths of 1 μm, 1.5 μm and 2 μm.</p>
Full article ">
14 pages, 8868 KiB  
Article
Design of a Compact Unified SIW Cavity Filtenna Module for Antenna Array Application
by Andrey Altynnikov, Roman Platonov, Alexey Sosunov, Tatyana Legkova, Andrey Komlev and Andrey B. Kozyrev
Micromachines 2025, 16(3), 285; https://doi.org/10.3390/mi16030285 - 28 Feb 2025
Viewed by 158
Abstract
The design of a multilayer SIW cavity-fed filtenna is presented. The proposed filtenna can be used as a unified module in an antenna array structure. It consists of three-pole bandpass filter with slot antenna positioned centrally within the top module surface. The modules [...] Read more.
The design of a multilayer SIW cavity-fed filtenna is presented. The proposed filtenna can be used as a unified module in an antenna array structure. It consists of three-pole bandpass filter with slot antenna positioned centrally within the top module surface. The modules aperture dimensions of λ0/2×λ0/2 in conjunction with an SMA feeding port located on the bottom filtenna surface allow implementation of an antenna array of different configurations. This approach allows greatly simplifying the feeding and matching scheme of the array. This module is designed to operate at a 2.655 GHz central frequency with a 70 MHz bandwidth. The procedure of the filtenna design is described in detail. The proposed filtenna was fabricated and tested. The simulation and measurement results show a good agreement. The measurements demonstrate that the maximum measured gain of the prototype is 3.64 dBi with a small variation in the passband. Full article
Show Figures

Figure 1

Figure 1
<p>An equivalent circuit of a three-pole bandpass filter.</p>
Full article ">Figure 2
<p>Model of three-pole SIW filter prototype.</p>
Full article ">Figure 3
<p>Schematic of filter proyotype layers’ topology: (<b>a</b>)—bottom, (<b>b</b>)—middle and (<b>c</b>)—top substrate.</p>
Full article ">Figure 4
<p>Simulation results of filter prototype.</p>
Full article ">Figure 5
<p>Equivalent circuit of the filtenna based on SIW filter.</p>
Full article ">Figure 6
<p>Schematic view of the (<b>a</b>)—top, (<b>b</b>)—middle and (<b>c</b>)—bottom substrate.</p>
Full article ">Figure 7
<p>Top view—(<b>a</b>) and side view—(<b>b</b>) of filtenna model.</p>
Full article ">Figure 8
<p>Comparison of the simulation results of the SIW filter prototype—(<b>a</b>) and filtenna module—(<b>b</b>).</p>
Full article ">Figure 9
<p>Simulation results of cross coupling of the two filtenna modules oriented in a row— (<b>a</b>) and in parallel—(<b>b</b>).</p>
Full article ">Figure 10
<p>Comparison of the radiation pattern of single module and 16-element array.</p>
Full article ">Figure 11
<p>Photos of the assembled filtenna module prototype in the anechoic chamber: (<b>a</b>)—back view and (<b>b</b>)—front view of the module.</p>
Full article ">Figure 12
<p>Comparison of the experimental results of the filtenna’s gain and reflection coefficient on the frequency with simulated ones.</p>
Full article ">Figure 13
<p>Comparison of the experimental and simulation results of filtenna’s gain at different frequencies.</p>
Full article ">
13 pages, 6463 KiB  
Article
Design of an Aperiodic Optical Phased Array Based on the Multi-Strategy Enhanced Particle Swarm Optimization Algorithm
by Zhuangzhuang Zang, Junjie Wu and Qingzhong Huang
Photonics 2025, 12(3), 210; https://doi.org/10.3390/photonics12030210 - 27 Feb 2025
Viewed by 247
Abstract
We have proposed a multi-strategy enhanced particle swarm optimization (PSO) algorithm to optimize the antenna spacing distribution of an optical phased array (OPA). The global search capability is improved by incorporating circle chaotic mapping initialization and an updated strategy based on adaptive inertia [...] Read more.
We have proposed a multi-strategy enhanced particle swarm optimization (PSO) algorithm to optimize the antenna spacing distribution of an optical phased array (OPA). The global search capability is improved by incorporating circle chaotic mapping initialization and an updated strategy based on adaptive inertia weights and dynamic learning factors. We used the peak side-lobe level (PSLL) at different steering angles as the fitness function, which effectively suppresses the rapid degradation of PSLL during scanning. Based on this approach, 32- and 64-channel aperiodic OPAs were designed with a scanning range of ±60°, with improvements of the PSLL of 1.94 and 2.05 dB at 60°, respectively. In addition, the analytical and numerical simulation results are in good agreement. We also analyzed the influence of spacing deviations on PSLL and found that the obtained OPAs exhibit sufficient robustness. Full article
(This article belongs to the Section Lasers, Light Sources and Sensors)
Show Figures

Figure 1

Figure 1
<p>Schematic diagram of a 1D OPA with an aperiodically spaced antenna array.</p>
Full article ">Figure 2
<p>Schematic diagrams of (<b>a</b>) a uniform antenna array and (<b>b</b>) an aperiodic antenna array.</p>
Full article ">Figure 3
<p>(<b>a</b>) The spacing distribution, (<b>b</b>) normalized far-field pattern, and (<b>c</b>) 2D far-field pattern of the 16-channel uniform OPA. (<b>d</b>) The spacing distribution, (<b>e</b>) normalized far-field pattern, and (<b>f</b>) 2D far-field pattern of the 16-channel aperiodic OPA.</p>
Full article ">Figure 4
<p>Initialization distribution of (<b>a</b>) random function and (<b>b</b>) circle chaotic mapping strategy.</p>
Full article ">Figure 5
<p>Adaptive inertia weight and dynamic learning factors as functions of the iteration numbers.</p>
Full article ">Figure 6
<p>Optimization process of antenna spacing distribution based on the enhanced PSO algorithm.</p>
Full article ">Figure 7
<p>PSLL versus steering angle for the (<b>a</b>) 32- and (<b>b</b>) 64-channel aperiodic OPAs optimized with different steering angles. (<b>c</b>) Spacing distribution for the 32-channel aperiodic OPA optimized at the steering angle of 20°. (<b>d</b>) Spacing distribution for the 64-channel aperiodic OPA optimized at the steering angle of 40°.</p>
Full article ">Figure 8
<p>Objective function values for (<b>a</b>) 32- and (<b>b</b>) 64-channel aperiodic OPAs. PSLL versus steering angle for the (<b>c</b>) 32- and (<b>d</b>) 64-channel aperiodic OPAs.</p>
Full article ">Figure 9
<p>The far-field patterns of 64-channel aperiodic OPAs: (<b>a</b>) with the traditional PSO algorithm and Fitness<sub>0</sub>° and (<b>b</b>) with the enhanced PSO algorithm and Fitness<sub>40°</sub>.</p>
Full article ">Figure 10
<p>The far-field patterns of the 32-channel aperiodic OPA at 0° for (<b>a</b>) the analytical results and (<b>b</b>) the numerical results. (<b>c</b>) The 2D far-field pattern of the 32-channel aperiodic OPA at 0°. The far-field patterns of the 32-channel aperiodic OPA at 60° for (<b>d</b>) the analytical results and (<b>e</b>) the numerical results. (<b>f</b>) The 2D far-field pattern of the 32-channel aperiodic OPA at 60°.</p>
Full article ">Figure 11
<p>The far-field patterns of the 64-channel aperiodic OPA at 0° for (<b>a</b>) the analytical results and (<b>b</b>) the numerical results. (<b>c</b>) The 2D far-field pattern of the 64-channel aperiodic OPA at 0°. The far-field patterns of the 64-channel aperiodic OPA at 60° for (<b>d</b>) the analytical results and (<b>e</b>) the numerical results. (<b>f</b>) The 2D far-field pattern of the 64-channel aperiodic OPA at 60°.</p>
Full article ">Figure 12
<p>PSLL versus the wavelength at steering angles of 0°, 30°, and 60° for the aperiodic OPAs with (<b>a</b>) 32 channels and (<b>b</b>) 64 channels.</p>
Full article ">Figure 13
<p>Influence of spacing deviations on PSLL.</p>
Full article ">
26 pages, 9151 KiB  
Article
Beam-Switching Antennas Using a Butler Matrix with a Five-Element Configuration
by Wei-Heng Peng and Yen-Sheng Chen
Electronics 2025, 14(5), 959; https://doi.org/10.3390/electronics14050959 - 27 Feb 2025
Viewed by 200
Abstract
Beam-switching technology is critical for fifth-generation (5G) Frequency Range 1 (FR1) base stations, yet existing odd-number Butler matrix designs often struggle to achieve compact size, low complexity, and efficient performance. Although a few studies have investigated 5 × 5 Butler matrices, their reliance [...] Read more.
Beam-switching technology is critical for fifth-generation (5G) Frequency Range 1 (FR1) base stations, yet existing odd-number Butler matrix designs often struggle to achieve compact size, low complexity, and efficient performance. Although a few studies have investigated 5 × 5 Butler matrices, their reliance on waveguide structures or multilayer implementations leads to larger footprints and greater fabrication complexity. This work introduces a novel 5 × 5 Butler matrix integrated with a five-element dipole antenna array for 3.3–3.7 GHz applications, offering notable improvements in beam-switching efficiency and overall system design. The proposed matrix generates five distinct beams at −144°, −72°, 0°, 72°, and 144° by employing precise phase progression, while eliminating crossovers and power dividers to simplify the layout. With a compact footprint of 2.67 × 0.80 × 0.02 cubic wavelength—94.4% smaller than waveguide-based designs—the matrix achieves a bandwidth of 3.32–3.62 GHz and consistently covers the target beams. The integrated system attains measured gains up to 11.4 dBi and half-power beamwidths ranging from 7.96° to 23.66°, with sidelobe levels comparable to those of state-of-the-art configurations. By employing a low-loss substrate, the gain can be further enhanced by as much as 6.81 dB, highlighting the potential for additional performance gains. These innovations establish the proposed design as a compact, low-complexity, and high-performance solution for 5G base station applications. Full article
Show Figures

Figure 1

Figure 1
<p>(<b>a</b>) Geometry and (<b>b</b>) photograph of the proposed single antenna element.</p>
Full article ">Figure 1 Cont.
<p>(<b>a</b>) Geometry and (<b>b</b>) photograph of the proposed single antenna element.</p>
Full article ">Figure 2
<p>Parametric studies of (<b>a</b>) <span class="html-italic">D<sub>LW</sub></span> and (<b>b</b>) <span class="html-italic">R<sub>W</sub></span>.</p>
Full article ">Figure 3
<p>Current distribution of the dipole at (<b>a</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span>/4, (<b>b</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span>/2, (<b>c</b>) <span class="html-italic">t</span> = 3<span class="html-italic">T</span>/4, and (<b>d</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span> at 3.5 GHz.</p>
Full article ">Figure 3 Cont.
<p>Current distribution of the dipole at (<b>a</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span>/4, (<b>b</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span>/2, (<b>c</b>) <span class="html-italic">t</span> = 3<span class="html-italic">T</span>/4, and (<b>d</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span> at 3.5 GHz.</p>
Full article ">Figure 4
<p>Simulated and measured reflection coefficients of the single antenna element.</p>
Full article ">Figure 5
<p>Simulated and measured antenna efficiency of the single antenna element.</p>
Full article ">Figure 6
<p>Simulated and measured antenna patterns on (<b>a</b>) E- and (<b>b</b>) H-plane at 3.5 GHz.</p>
Full article ">Figure 7
<p>(<b>a</b>) Layout and (<b>b</b>) photograph of the proposed 5 × 5 Butler matrix.</p>
Full article ">Figure 8
<p>Simulated array factor achieved using the proposed 5 × 5 Butler matrix.</p>
Full article ">Figure 9
<p>Detailed configurations of the 10 units implemented in the Butler matrix.</p>
Full article ">Figure 10
<p>The comparison of measured and simulated amplitudes for the output of (<b>a</b>) port 6, (<b>b</b>) port 7, (<b>c</b>) port 8, (<b>d</b>) port 9, and (<b>e</b>) port 10.</p>
Full article ">Figure 11
<p>The comparison of measured and simulated phases for the input of (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, (<b>d</b>) port 4, and (<b>e</b>) port 5.</p>
Full article ">Figure 12
<p>(<b>a</b>) Top and (<b>b</b>) bottom view of the fabricated prototype. (<b>c</b>) Simulated beam-switching patterns at 3.5 GHz, obtained using an HFSS model where each port is excited independently with wave ports. The simulation replicates the fabricated prototype while excluding coaxial cables and support structures to focus on electromagnetic behavior.</p>
Full article ">Figure 12 Cont.
<p>(<b>a</b>) Top and (<b>b</b>) bottom view of the fabricated prototype. (<b>c</b>) Simulated beam-switching patterns at 3.5 GHz, obtained using an HFSS model where each port is excited independently with wave ports. The simulation replicates the fabricated prototype while excluding coaxial cables and support structures to focus on electromagnetic behavior.</p>
Full article ">Figure 13
<p>(<b>a</b>) Configuration and (<b>b</b>) photograph of the fabricated antenna fed by the 5 × 5 Butler matrix.</p>
Full article ">Figure 13 Cont.
<p>(<b>a</b>) Configuration and (<b>b</b>) photograph of the fabricated antenna fed by the 5 × 5 Butler matrix.</p>
Full article ">Figure 14
<p>Simulated and measured impedance matching at (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, (<b>d</b>) port 4, and (<b>e</b>) port 5.</p>
Full article ">Figure 14 Cont.
<p>Simulated and measured impedance matching at (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, (<b>d</b>) port 4, and (<b>e</b>) port 5.</p>
Full article ">Figure 15
<p>Simulated and measured radiation pattern fed by (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, (<b>d</b>) port 4, and (<b>e</b>) port 5 at 3.3 GHz.</p>
Full article ">Figure 16
<p>Simulated and measured radiation pattern fed by (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, (<b>d</b>) port 4, and (<b>e</b>) port 5 at 3.5 GHz.</p>
Full article ">Figure 17
<p>Simulated and measured radiation pattern fed by (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, (<b>d</b>) port 4, and (<b>e</b>) port 5 at 3.7 GHz.</p>
Full article ">Figure 18
<p>Ideal layout of the proposed antenna that integrates antenna elements and Butler matrix on the same substrate.</p>
Full article ">Figure 19
<p>Radiation pattern of the ideal layout fed at (<b>a</b>) port 4 and (<b>b</b>) port 5.</p>
Full article ">Figure 19 Cont.
<p>Radiation pattern of the ideal layout fed at (<b>a</b>) port 4 and (<b>b</b>) port 5.</p>
Full article ">
14 pages, 9498 KiB  
Article
Electromagnetic Absorber-Embedded Ka-Band Double-Layer Tapered Slot Antenna for the Reduced Radar Cross Section at X-Band
by Wonkyo Kim, Youngwan Kim, Hee-Duck Chae, Jihan Joo, Jun-Beom Kwon and Ick-Jae Yoon
Appl. Sci. 2025, 15(5), 2507; https://doi.org/10.3390/app15052507 - 26 Feb 2025
Viewed by 204
Abstract
An electromagnetic (EM) absorber-embedded Ka-band double-layer tapered slot antenna (DLTSA) is proposed in this work. The EM absorber is placed on both sides of the tapered radiating slots as a means of achieving the reduced monostatic radar cross section (RCS) at the X-band. [...] Read more.
An electromagnetic (EM) absorber-embedded Ka-band double-layer tapered slot antenna (DLTSA) is proposed in this work. The EM absorber is placed on both sides of the tapered radiating slots as a means of achieving the reduced monostatic radar cross section (RCS) at the X-band. A conventional tapered slot antenna (TSA) with EM absorbers at the same position suffers from the distorted current distribution from the feedline to the radiating slots and causes a degraded radiation performance with a tilted beam. In contrast, the DLTSA with EM absorbers maintains the impedance and radiation characteristics of the antenna without the EM absorbers, while achieving the reduced monostatic RCS for the cross-polarized incident wave. The functionality of the reduced RCS is verified with the 4-by-4 DLTSA array design. The 4-by-4 array prototype with FGM-125 EM absorbers is matched at the Ka-band with a 14.7 dBi boresight gain at 35 GHz. The monostatic RCS is measured in an indoor environment, showing 6.5 dB monostatic RCS reduction at the X-band on average, verifying the computed expectations. This work validates the possible use of EM absorbers at the front side of a missile seeker composed of end-fire radiating elements. Full article
(This article belongs to the Special Issue Multi-Band/Broadband Antenna Design, Optimization and Measurement)
Show Figures

Figure 1

Figure 1
<p>Proposed double-layer TSA (DLTSA). (<b>a</b>) Dissemble view. (<b>b</b>) Side view of the assembled DLTSA without EM absorbers. (<b>c</b>) Side view of the assembled DLTSA with EM absorbers.</p>
Full article ">Figure 2
<p>Simulation results for the DLTSA according to the presence of the EM absorber (FGM-125 on <a href="#applsci-15-02507-f001" class="html-fig">Figure 1</a>b,c). (<b>a</b>) Reflection coefficients. (<b>b</b>) The <math display="inline"><semantics> <mrow> <mi>θ</mi> </mrow> </semantics></math>-sweep radiation patterns at 35 GHz (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> </mrow> </semantics></math> = 90°).</p>
Full article ">Figure 3
<p>Simulation results. (<b>a</b>) Surface current distribution of a conventional TSA with the EM absorber (FGM-125) at 35 GHz. (<b>b</b>) Reflection coefficients of the conventional TSA according to the presence of the EM absorber. (<b>c</b>) The <math display="inline"><semantics> <mrow> <mi>θ</mi> </mrow> </semantics></math>-sweep radiation patterns at 35 GHz (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> </mrow> </semantics></math> = 90°) for the conventional TSA. (<b>d</b>) Surface current distribution of the DLTSA with the EM absorber at 35 GHz.</p>
Full article ">Figure 4
<p>Simulation results for the conventional TSA and proposed DLTSA with the EM absorber (<a href="#applsci-15-02507-f003" class="html-fig">Figure 3</a>a,d). (<b>a</b>) Reflection coefficients. (<b>b</b>) The <math display="inline"><semantics> <mrow> <mi>θ</mi> </mrow> </semantics></math>-sweep radiation patterns at 35 GHz (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> </mrow> </semantics></math> = 90°).</p>
Full article ">Figure 5
<p>Simulation results of the DLTSA according to the thickness (<span class="html-italic">L<sub>4</sub></span>) of the EM absorber. (<b>a</b>) Reflection coefficients. (<b>b</b>) The <math display="inline"><semantics> <mrow> <mi>θ</mi> </mrow> </semantics></math>-sweep radiation pattern (<math display="inline"><semantics> <mrow> <mi>ϕ</mi> </mrow> </semantics></math> = 90°). (<b>c</b>) Radiation efficiency. (<b>d</b>) Cross-polarized monostatic RCS results.</p>
Full article ">Figure 6
<p>Simulated results of the 1-by-4 power divider. (<b>a</b>) Power divider design. (<b>b</b>) Simulated S-parameter results. (<b>c</b>) Surface current distribution at 35 GHz.</p>
Full article ">Figure 7
<p>Simulated 4-by-4 array DLTSA. (<b>a</b>) Array design with the proposed DLTSAs with FGM-125. (<b>b</b>) Array design with the proposed DLTSAs without FGM-125. (<b>c</b>) Reflection coefficients. (<b>d</b>) <span class="html-italic">θ</span>-sweep radiation patterns (<span class="html-italic">ϕ</span> = 90°).</p>
Full article ">Figure 8
<p>4-by-4 array DLTSA. (<b>a</b>) Photo of the built prototype. (<b>b</b>) Radiation pattern measurement environment (CATR). (<b>c</b>) Simulated and measured reflection coefficients. (<b>d</b>) Simulated and measured <span class="html-italic">θ</span>-sweep radiation patterns (<span class="html-italic">ϕ</span> = 90°). (<b>e</b>) Measured <span class="html-italic">θ</span>-sweep radiation patterns (<span class="html-italic">ϕ</span> = 90°).</p>
Full article ">Figure 9
<p>RCS measurement environment.</p>
Full article ">Figure 10
<p>RCS results. (<b>a</b>) Simulation. (<b>b</b>) Measurement.</p>
Full article ">Figure 11
<p>Beam steering simulation results. (<b>a</b>) Phase excitation by 90°. (<b>b</b>) Simulated 3D pattern at 35 GHz.</p>
Full article ">
17 pages, 5727 KiB  
Article
Development and Implementation of High-Gain, and High-Isolation Multi-Input Multi-Output Antenna for 5G mmWave Communications
by Mahmoud Shaban
Telecom 2025, 6(1), 14; https://doi.org/10.3390/telecom6010014 - 25 Feb 2025
Viewed by 210
Abstract
This work introduces a high-performance multi-input multi-output (MIMO) antenna design to operate at the 28 GHz band. The proposed four-port MIMO antenna, in which each port comprises a 1 × 8 series-fed array, achieves peak gains of 13 dBi along with bandwidths of [...] Read more.
This work introduces a high-performance multi-input multi-output (MIMO) antenna design to operate at the 28 GHz band. The proposed four-port MIMO antenna, in which each port comprises a 1 × 8 series-fed array, achieves peak gains of 13 dBi along with bandwidths of 1 GHz. Enhanced antenna performance is achieved through the optimal spacing of antenna elements and a decoupling methodology comprising a well-designed metamaterial unit cell, leading to reduced interference between antenna arrays. The design shows significantly suppressed mutual coupling to be less than −40 dB, a diversity gain that is very close to 10 dB, an envelope correlation coefficient of 0.00012, and a channel capacity loss of 0.147 bit/s/Hz, at 28 GHz. The experimental assessments confirmed these developments, endorsing the suggested design as a robust contender for 5G mmWave communications. Full article
(This article belongs to the Special Issue Advances in Wireless Communication: Applications and Developments)
Show Figures

Figure 1

Figure 1
<p>(<b>a</b>) Front, (<b>b</b>) back view of single-port antenna; and (<b>c</b>) dimensions of slots created in each single-patch.</p>
Full article ">Figure 2
<p>Simulated reflection coefficient of the single-port antenna as a function of frequency. The highlighted area shows bandwidth range.</p>
Full article ">Figure 3
<p>A parametric analysis of the inset length for the proposed antenna design.</p>
Full article ">Figure 4
<p>Frequency dependence of peak gain of single-array antenna.</p>
Full article ">Figure 5
<p>Radiation patterns of 1 × 8 series-fed array; (<b>a</b>) E-plane, (<b>b</b>) H-plane computed at 28.8 GHz.</p>
Full article ">Figure 6
<p>(<b>a</b>) Top, (<b>b</b>) bottom view of four-port MIMO antenna; and (<b>c</b>) simulated s-parameters versus frequency.</p>
Full article ">Figure 7
<p>(<b>a</b>) Top, (<b>b</b>) bottom views of MTM unit cell and its computed (<b>c</b>) ε<sub>r</sub>, (<b>d</b>) µ<sub>r</sub>, and (<b>e</b>) n parameters.</p>
Full article ">Figure 8
<p>Computed frequency dependence of (<b>a</b>) ECC and (<b>b</b>) DG of four-port MIMO antenna.</p>
Full article ">Figure 9
<p>Radiation patterns of the 4-port MIMO antenna at 28.8 GHz: (<b>a</b>) E-plane and (<b>b</b>) H-plane.</p>
Full article ">Figure 10
<p>Measurement setup of (<b>a</b>) evaluating the reflection coefficients of individual ports, (<b>b</b>,<b>c</b>) assessing mutual coupling coefficients among the ports, and (<b>d</b>) displaying a sample result from the measurement of S<sub>11</sub>.</p>
Full article ">
8 pages, 17999 KiB  
Article
4 × 4 Wideband Slot Antenna Array Fed by TE440 Mode Based on Groove Gap Waveguide
by Yuanjun Shen, Tianling Zhang, Liangqin Luo, Honghuan Zhu and Lei Chen
Electronics 2025, 14(4), 813; https://doi.org/10.3390/electronics14040813 - 19 Feb 2025
Viewed by 270
Abstract
A 4 × 4 wideband millimeter-wave (mmWave) slot array antenna excited by the TE440 mode based on the groove gap waveguide is presented in this paper. A vertical waveguide located in the center of the cavity and two ridges are used to [...] Read more.
A 4 × 4 wideband millimeter-wave (mmWave) slot array antenna excited by the TE440 mode based on the groove gap waveguide is presented in this paper. A vertical waveguide located in the center of the cavity and two ridges are used to excite the TE440 mode. In addition, a pair of corrugations acting as the soft surface are added on the top of the array antenna to improve the gain. A 4 × 4 prototype is fabricated and measured. The measured and simulated results are in great agreement. The measured results show that the proposed array antenna achieved an impedance bandwidth (|S11| < −10 dB) of 26.7% from 26.14 to 34.2 GHz, and the maximum gain is 17.7 dBi. The proposed array antenna avoids the complicated feeding network, allowing us to reduce the manufacturing cost. Full article
(This article belongs to the Special Issue Antenna and Array Design for Future Sensing and Communication System)
Show Figures

Figure 1

Figure 1
<p>The geometry of the 4 × 4 slot array antenna introduced using TE<sub>440</sub>.</p>
Full article ">Figure 2
<p>The dispersion diagram of the unit cell.</p>
Full article ">Figure 3
<p>The geometry of the metal plates (in mm): (<b>a</b>) top plate and (<b>b</b>) bottom plate.</p>
Full article ">Figure 4
<p>The simulated E-field distribution within the GGW cavity without slots at 31.62 GHz.</p>
Full article ">Figure 5
<p>The evolution process of the newly introduced wideband 4 × 4 slot array antenna.</p>
Full article ">Figure 6
<p>The simulated results of the proposed 4 × 4 wideband slot array antenna after Steps 1–3: (<b>a</b>) reflection coefficient and (<b>b</b>) gain.</p>
Full article ">Figure 7
<p>The simulated maximum E-field distribution over the antenna’s top surface at 29 GHz: (<b>a</b>) without ridges and corrugations, (<b>b</b>) with ridges but without corrugations, and (<b>c</b>) with both ridges and corrugations.</p>
Full article ">Figure 8
<p>Photographs of the prototype: (<b>a</b>) from the oblique top, (<b>b</b>) from the oblique top (inside), and (<b>c</b>) from the bottom.</p>
Full article ">Figure 9
<p>The measured and simulated reflection coefficients of the newly introduced 4 × 4 wideband slot array antenna.</p>
Full article ">Figure 10
<p>The measured and simulated radiation patterns of the proposed 4 × 4 wideband slot array antenna: (<b>a</b>) 27 GHz, (<b>b</b>) 30.5 GHz, and (<b>c</b>) 34 GHz.</p>
Full article ">Figure 11
<p>The measured and simulated gains of the newly introduced slot array antenna.</p>
Full article ">
22 pages, 5289 KiB  
Article
Design of the New Dual-Polarized Broadband Phased Array Feed Antenna for the Sardinia Radio Telescope
by Paolo Maxia, Giovanni Andrea Casula, Alessandro Navarrini, Tonino Pisanu, Giuseppe Valente, Giacomo Muntoni and Giorgio Montisci
Electronics 2025, 14(4), 807; https://doi.org/10.3390/electronics14040807 - 19 Feb 2025
Viewed by 226
Abstract
High-sensitivity and large-scale surveys are essential in advancing radio astronomy, enabling detailed exploration of the universe. A Phased Array Feed (PAF) installed in the focal plane of a radio telescope significantly enhances mapping efficiency by increasing the instantaneous Field of View (FoV) and [...] Read more.
High-sensitivity and large-scale surveys are essential in advancing radio astronomy, enabling detailed exploration of the universe. A Phased Array Feed (PAF) installed in the focal plane of a radio telescope significantly enhances mapping efficiency by increasing the instantaneous Field of View (FoV) and improving sky sampling capabilities. This paper presents the design and optimization of a novel C-Band Phased Array Feed antenna for the Sardinia Radio Telescope (SRT). The system features an 8 × 8 array of dual-polarized elements optimized to achieve a uniform beam pattern and an edge taper of approximately 5 dB for single radiating elements within the 3.0–7.7 GHz frequency range. The proposed antenna addresses key efficiency limitations identified in the PHAROS 2 (PHased Arrays for Reflector Observing Systems) system, including the under-illumination of the Sardinia Radio Telescope’s primary mirror caused by narrow sub-array radiation patterns. By expanding the operational bandwidth and refining the radiation characteristics, this new design enables significantly improved performance across the broader frequency range of 3.0–7.7 GHz, enhancing the telescope’s capability for wide-field, high-resolution observations. Full article
(This article belongs to the Special Issue Microwave Devices: Analysis, Design, and Application)
Show Figures

Figure 1

Figure 1
<p>Simplified layout of the 64-element array, showing the central 32 active elements highlighted in orange.</p>
Full article ">Figure 2
<p>Diagram of a typical Linear Tapered Slot Antenna with key parameters, such as taper length (<span class="html-italic">L<sub>t</sub></span>), opening angle α, taper width (<span class="html-italic">W<sub>t</sub></span>), and feed slot width (<span class="html-italic">W<sub>s</sub></span>), clearly marked.</p>
Full article ">Figure 3
<p>Spacing between adjacent elements of the array.</p>
Full article ">Figure 4
<p>(<b>a</b>) Unit cell; (<b>b</b>) parameters of the feeding network.</p>
Full article ">Figure 5
<p>Implementation of the feeding probe integrated in the antenna.</p>
Full article ">Figure 6
<p>(<b>a</b>) 3D view of the 128-element array; (<b>b</b>) antenna feeding scheme for edge taper adjustment (in red color the active zone): the antenna numbered with 93 and highlighted in green is the only one that is powered.</p>
Full article ">Figure 7
<p>H-plane 5 dB edge taper: variation of taper length <span class="html-italic">L<sub>t</sub></span>.</p>
Full article ">Figure 8
<p>E-plane 5 dB edge taper: variation of taper length <span class="html-italic">L<sub>t</sub></span>.</p>
Full article ">Figure 9
<p>Single Linear Tapered Slot Antenna H-plane-normalized radiated field.</p>
Full article ">Figure 10
<p>Single Linear Tapered Slot Antenna E-plane-normalized radiated field.</p>
Full article ">Figure 11
<p>The −5 dB H-plane edge taper comparison.</p>
Full article ">Figure 12
<p>The −5 dB E-plane edge taper comparison.</p>
Full article ">Figure 13
<p>(<b>a</b>) Overall dimensions of the optimized unit cell, and (<b>b</b>) optimized parameters of the “L-Shaped” slot: all dimensions are in millimeters.</p>
Full article ">Figure 14
<p>S-parameters of the dual-polarized unit cell.</p>
Full article ">Figure 15
<p>(<b>a</b>) Finite array model with additional boundary elements, and (<b>b</b>) the antenna feeding scheme for the final configuration of the Phased Array antenna: the 16 passive additional elements are highlighted in blue color. In red color is the active zone.</p>
Full article ">Figure 16
<p>Comparison between the Active Reflection Coefficients of the single antenna of the unit cell and the antenna labelled with number 93 in the finite array.</p>
Full article ">Figure 17
<p>Simulated mutual coupling between some elements having the same polarization.</p>
Full article ">Figure 18
<p>Simulated mutual coupling between some elements having cross-polarization.</p>
Full article ">
21 pages, 5859 KiB  
Article
Genetic Algorithm-Based Acoustic Array Optimization for Estimating UAV DOA Using Beamforming
by Nathan Itare, Jean-Hugh Thomas and Kosai Raoof
Drones 2025, 9(2), 149; https://doi.org/10.3390/drones9020149 - 18 Feb 2025
Viewed by 264
Abstract
The localization of unmanned aerial vehicles is an important topic due to several threats near sensitive sites. Localization based on their sounds has been a particular point of interest in past studies for many years. It requires the use of a microphone array. [...] Read more.
The localization of unmanned aerial vehicles is an important topic due to several threats near sensitive sites. Localization based on their sounds has been a particular point of interest in past studies for many years. It requires the use of a microphone array. The positioning of the various microphones making up an antenna defines the intrinsic directivity of the array. In this study, a genetic algorithm is used to determine the microphone positions that optimize directivity in a focus direction and for a frequency, by favoring the narrowness of the main lobe and the reduction of the secondary lobes. The optimization leads to several antennas with a 3D structure similar to that designed in a previous study. A method estimating the direction of arrival of a drone was also presented in that study making use of its acoustic signature to enhance the signal-to-noise ratio and thus improving the estimations. In this paper, an improvement to the method is proposed for tracking the drone’s trajectory. Measurements were conducted to compare the drone locations given by the first designed antenna and the one optimized by the genetic algorithm. Performance on the direction of arrival found is characterized in terms of mean error, standard deviation and root mean square error relative to the GPS reference onboard the UAV. An experiment with the optimized antenna has also been conducted with the drone at a great distance to the antenna to characterize the maximal distance for possible estimations of the direction of arrival. Results show that the method used for the direction of arrival estimation can give a mean error below 10° in azimuth and 5° in elevation. The maximum distance between the antenna and the drone for which the method is able to give estimations is between 240 and 340 m. Full article
(This article belongs to the Special Issue Technologies and Applications for Drone Audition)
Show Figures

Figure 1

Figure 1
<p>Spectrogram of the signal acquired by the first microphone of the array.</p>
Full article ">Figure 2
<p>Directivity factor of the initial array for 175 Hz and (−45°, 20°) focus direction (<b>a</b>) and for 350 Hz and (45°, 45°) focus direction (<b>b</b>).</p>
Full article ">Figure 3
<p>Genetic algorithm process.</p>
Full article ">Figure 4
<p>Directivity factor of (<b>a</b>) the initial array for 350 Hz frequency and direction (45°, 45°), the optimized array with (<b>b</b>) 100 generations, (<b>c</b>) 200 generations.</p>
Full article ">Figure 5
<p>(<b>a</b>) Microphone arrangement for the initial array and the optimized arrays with (<b>b</b>) 100 generations and (<b>c</b>) 200 generations.</p>
Full article ">Figure 6
<p>Maps highlighting the directivity factor parameters (lobe widths, MSL) for the initial array and optimized arrays with 100 and 200 generations.</p>
Full article ">Figure 7
<p>Synopsis of the time–frequency representation method.</p>
Full article ">Figure 8
<p>Spectrogram of the signal acquired by the first microphone with different bandwidths corresponding to different quality factors Q.</p>
Full article ">Figure 9
<p>Spectrogram of the signal acquired by the first microphone of the array during the drone’s circular trajectory using the maximum criterion in SHC. The red points are candidate frequencies whose harmonics are selected in the TFR.</p>
Full article ">Figure 10
<p>Azimuth and elevation vs. time for the circular trajectory with conventional beamforming and TFR method using two criteria for candidate selection in the SHC: maximum criterion (2nd line) and continuity criterion (last line).</p>
Full article ">Figure 11
<p>Energy maps for the TFR method at t = 9.125 s with the corresponding SHC (first column), at t = 9.25 s with the corresponding SHC using the maximum (second column) or the continuity criteria (third column). Crosses and circles correspond, respectively, to estimations obtained with the TFR and conventional beamforming.</p>
Full article ">Figure 12
<p>Trajectory performed by the drone for antenna comparisons.</p>
Full article ">Figure 13
<p>Spectrograms of the signals acquired by the first microphone of the array for the four measurements. The red dots are candidate frequencies whose harmonics are selected in the TFR.</p>
Full article ">Figure 14
<p>Azimuth vs. time for the TFR method in the four configurations of measurements.</p>
Full article ">Figure 15
<p>Elevation vs. time for the TFR method in the four configurations of measurements.</p>
Full article ">Figure 16
<p>Examples of energy maps given by conventional beamforming (first line) and by the TFR method (second line) for two instants and two antenna configurations where the drone is in front of the antennas. Circles correspond to DOA estimations given by conventional beamforming and crosses those given by the TFR method.</p>
Full article ">Figure 17
<p>Trajectory performed by the drone for the long-distance measurement.</p>
Full article ">Figure 18
<p>Spectrogram of the signal acquired by the first microphone of the optimized array during the long-distance measurement (<b>a</b>). Red points are candidate frequencies whose harmonics are selected in the TFR method. Areas representing car acceleration are circled. Azimuth (<b>b</b>) and elevation (<b>c</b>) vs. time for the TFR method. Evolution of the distance over time between the drone and the antenna according to GPS data embedded on the drone (<b>d</b>).</p>
Full article ">
14 pages, 10104 KiB  
Article
A Compact and Wideband Beam-Scanning Antenna Array Based on SICL Butler Matrix
by Zhu Hua, Chuang Gao, Jiejun Peng, Shuting Fan and Zhengfang Qian
Electronics 2025, 14(4), 757; https://doi.org/10.3390/electronics14040757 - 15 Feb 2025
Viewed by 269
Abstract
A compact and wideband beamforming antenna array based on a substrate-integrated coaxial line (SICL) Butler matrix at 60 GHz is proposed in this paper. The cavity-backed patch antenna loading double-ridged horn antenna is designed to enhance a gain of 5.4 dB and a [...] Read more.
A compact and wideband beamforming antenna array based on a substrate-integrated coaxial line (SICL) Butler matrix at 60 GHz is proposed in this paper. The cavity-backed patch antenna loading double-ridged horn antenna is designed to enhance a gain of 5.4 dB and a bandwidth of 2.7 GHz. Different phase centers of double-ridged horn elements are formed into a non-uniform array to reduce sidelobes by −7.9 dB. By introducing the defected ground structure (DGS) for a broadband coupler, a rotationally symmetric SICL Butler matrix is designed with a 55–70 GHz bandwidth and compact dimensions of 63 × 65 × 0.512 mm3. To validate the design, a prototype was fabricated and measured. The experimental results show a wideband −10 dB impedance bandwidth of 23.3% (55.4–70 GHz) with measured gains ranging from 15 to 16.1 dBi at 62 GHz. The one-dimensional beam scanning covers ±32°. The simulation and measurement results are in good agreement. Full article
(This article belongs to the Special Issue Antennas and Microwave/Millimeter-Wave Applications)
Show Figures

Figure 1

Figure 1
<p>The configuration of SICL: (<b>a</b>) side view of SICL; (<b>b</b>) vertical views of SIW and SICL.</p>
Full article ">Figure 2
<p>The simulated |<span class="html-italic">S</span><sub>21</sub>| of SICL.</p>
Full article ">Figure 3
<p>The cavity-backed ridged hybrid horn antenna element: (<b>a</b>) 3D model; (<b>b</b>) top metal layer of substrate 1; (<b>c</b>) top metal layer of substrate 3; (<b>d</b>) bottom metal layer of substrate 3.</p>
Full article ">Figure 4
<p>Simulated result: (<b>a</b>) |<span class="html-italic">S</span><sub>11</sub>|; (<b>b</b>) simulated radiation patterns; (<b>c</b>) gain curve with frequency.</p>
Full article ">Figure 4 Cont.
<p>Simulated result: (<b>a</b>) |<span class="html-italic">S</span><sub>11</sub>|; (<b>b</b>) simulated radiation patterns; (<b>c</b>) gain curve with frequency.</p>
Full article ">Figure 5
<p>Configuration of the 4 × 4 Butler matrix: (<b>a</b>) traditional Butler matrix; (<b>b</b>) rotationally symmetric Butler matrix.</p>
Full article ">Figure 6
<p>Planar 3 dB three-branch-line coupler with DGS: (<b>a</b>) 3D model; (<b>b</b>) signal line; (<b>c</b>) Top/bottom grounds.</p>
Full article ">Figure 6 Cont.
<p>Planar 3 dB three-branch-line coupler with DGS: (<b>a</b>) 3D model; (<b>b</b>) signal line; (<b>c</b>) Top/bottom grounds.</p>
Full article ">Figure 7
<p>Simulation results of SICL 3 dB coupler: (<b>a</b>) S-parameter; (<b>b</b>) phase difference.</p>
Full article ">Figure 8
<p>The waveguide (WR-15)-to-SICL transition: (<b>a</b>) configuration; (<b>b</b>) simulated |<span class="html-italic">S</span><sub>11</sub>| and |<span class="html-italic">S</span><sub>12</sub>|.</p>
Full article ">Figure 9
<p>Configuration of SICL Butler matrix.</p>
Full article ">Figure 10
<p>Simulated results of SICL Butler matrix: (<b>a</b>) |<span class="html-italic">S</span><sub>11</sub>|, |<span class="html-italic">S</span><sub>22</sub>|, |<span class="html-italic">S</span><sub>33</sub>|, and |<span class="html-italic">S</span><sub>44</sub>|; (<b>b</b>) |<span class="html-italic">S</span><sub>51</sub>|, |<span class="html-italic">S</span><sub>61</sub>|, |<span class="html-italic">S</span><sub>71</sub>|, |<span class="html-italic">S</span><sub>81</sub>|;|<span class="html-italic">S</span><sub>52</sub>|, |<span class="html-italic">S</span><sub>62</sub>|, |<span class="html-italic">S</span><sub>72</sub>|, and |<span class="html-italic">S</span><sub>82</sub>|; (<b>c</b>) phase difference.</p>
Full article ">Figure 11
<p>Non-uniform double-ridged horn antenna array: (<b>a</b>) 3D view; (<b>b</b>) perspective view; (<b>c</b>) configuration.</p>
Full article ">Figure 12
<p>Simulated normalized radiation pattern of non-uniform and uniform antenna arrays.</p>
Full article ">Figure 13
<p>Fabricated prototype: (<b>a</b>) top view; (<b>b</b>) bottom view; (<b>c</b>) compensated compact range (CCR) antenna measurement environment.</p>
Full article ">Figure 13 Cont.
<p>Fabricated prototype: (<b>a</b>) top view; (<b>b</b>) bottom view; (<b>c</b>) compensated compact range (CCR) antenna measurement environment.</p>
Full article ">Figure 14
<p>Simulated and measured results of proposed beam-scanning antenna array: (<b>a</b>) S-parameter; (<b>b</b>) radiation patterns at 62 GHz.</p>
Full article ">Figure 14 Cont.
<p>Simulated and measured results of proposed beam-scanning antenna array: (<b>a</b>) S-parameter; (<b>b</b>) radiation patterns at 62 GHz.</p>
Full article ">
12 pages, 2891 KiB  
Article
Dual-Band Multi-Layer Antenna Array with Circular Polarization and Gain Enhancement for WLAN and X-Band Applications
by Bal S. Virdee, Tohid Aribi and Tohid Sedghi
Micromachines 2025, 16(2), 203; https://doi.org/10.3390/mi16020203 - 10 Feb 2025
Viewed by 564
Abstract
This paper presents a novel multi-layer, dual-band antenna array designed for WLAN and X-band applications, incorporating several innovative features. The design employs a pentagon-shaped radiating element with parasitic strips to enable dual-band operation. A dual-transformed feed network with chamfered feed strip corners minimizes [...] Read more.
This paper presents a novel multi-layer, dual-band antenna array designed for WLAN and X-band applications, incorporating several innovative features. The design employs a pentagon-shaped radiating element with parasitic strips to enable dual-band operation. A dual-transformed feed network with chamfered feed strip corners minimizes radiation distortion and cross-polarization while introducing orthogonal phase shifts to achieve circular polarization (CP) at the X-band. A Fabry–Pérot structure, strategically placed above the array, enhances gain in the WLAN band. The antenna demonstrates an impedance bandwidth of 1.8 GHz (S11 < −10 dB) at the WLAN band, with 36% fractional bandwidth, and 4.3 GHz at the X-band, with 43% fractional bandwidth. Measured peak gains are 7 dBi for the WLAN band and 6.8 dBi for the X-band, with favourable S11 levels, omni-directional radiation patterns, and consistent gain across both bands. Circular polarization is achieved within 8.5–10.4 GHz. Experimental results confirm the array’s significant advancements in multi-band performance, making it highly suitable for diverse wireless communication applications. Full article
(This article belongs to the Special Issue RF Devices: Technology and Progress)
Show Figures

Figure 1

Figure 1
<p>(<b>a</b>) Top layer with Fabry–Pérot array structure, (<b>b</b>) bottom layer of array with four parasitic resonators, and (<b>c</b>) the defected ground structure.</p>
Full article ">Figure 2
<p>(<b>a</b>) Reflection coefficient response of the various radiating elements and (<b>b</b>) configuration of the proposed parasitic resonator used in the multi-band antenna array.</p>
Full article ">Figure 3
<p>Steps used to realize the proposed antenna, (<b>a</b>) Triangular-shaped patch flanked by parasitic resonators with standard truncated ground plane, (<b>b</b>) Diamond-shaped patch flanked by conformal parasitic resonators with standard truncated ground plane, and (<b>c</b>) Pentagon-shaped patch flanked by conformal parasitic resonators and modified ground plane.</p>
Full article ">Figure 4
<p>Simulated S<sub>11</sub> response of the (<b>a</b>) Feeding network, and (<b>b</b>) 4 × 1 antenna arrays.</p>
Full article ">Figure 5
<p>Scattering parameter response of the Fabry–Pérot unit cell.</p>
Full article ">Figure 6
<p>(<b>a</b>) Fabricated antenna array layers, (<b>b</b>) Assembled antenna array, (<b>c</b>) Measured and simulated reflection coefficient response of the proposed dual-band Fabry–Pérot antenna array, (<b>d</b>) Measured gain of the proposed FP antenna array with and without Fabry–Perot (FB), (<b>e</b>) Axial ratio of the fabricated FP antenna array, (<b>f</b>) Measured normalized radiation patterns for CP FP antenna array at 5.7 GHz, and (<b>g</b>) Measured normalized radiation patterns for CP FP antenna array at 9.5 GHz.</p>
Full article ">Figure 6 Cont.
<p>(<b>a</b>) Fabricated antenna array layers, (<b>b</b>) Assembled antenna array, (<b>c</b>) Measured and simulated reflection coefficient response of the proposed dual-band Fabry–Pérot antenna array, (<b>d</b>) Measured gain of the proposed FP antenna array with and without Fabry–Perot (FB), (<b>e</b>) Axial ratio of the fabricated FP antenna array, (<b>f</b>) Measured normalized radiation patterns for CP FP antenna array at 5.7 GHz, and (<b>g</b>) Measured normalized radiation patterns for CP FP antenna array at 9.5 GHz.</p>
Full article ">Figure 6 Cont.
<p>(<b>a</b>) Fabricated antenna array layers, (<b>b</b>) Assembled antenna array, (<b>c</b>) Measured and simulated reflection coefficient response of the proposed dual-band Fabry–Pérot antenna array, (<b>d</b>) Measured gain of the proposed FP antenna array with and without Fabry–Perot (FB), (<b>e</b>) Axial ratio of the fabricated FP antenna array, (<b>f</b>) Measured normalized radiation patterns for CP FP antenna array at 5.7 GHz, and (<b>g</b>) Measured normalized radiation patterns for CP FP antenna array at 9.5 GHz.</p>
Full article ">
20 pages, 8736 KiB  
Article
High-Performance Series-Fed Array Multiple-Input Multiple-Output Antenna for Millimeter-Wave 5G Networks
by Nabeel Alsaab, Khaled Alhassoon, Fahd Alsaleem, Fahad Nasser Alsunaydih, Sayed O. Madbouly, Sherif A. Khaleel, Allam M. Ameen and Mahmoud Shaban
Sensors 2025, 25(4), 1036; https://doi.org/10.3390/s25041036 - 9 Feb 2025
Viewed by 577
Abstract
This research presents a high-performance design for a multiple-input multiple-output (MIMO) antenna intended for operation within the 28 GHz band. The four-port MIMO antenna configuration, featuring 1 × 8 series-fed arrays for each port, has demonstrated peak gains of 15.5 dBi and bandwidths [...] Read more.
This research presents a high-performance design for a multiple-input multiple-output (MIMO) antenna intended for operation within the 28 GHz band. The four-port MIMO antenna configuration, featuring 1 × 8 series-fed arrays for each port, has demonstrated peak gains of 15.5 dBi and bandwidths of 2 GHz. This improved antenna performance results from carefully optimized antenna spacing and a decoupling approach involving well-designed metamaterial cells, effectively minimizing interference between antenna elements. The system exhibits remarkably low mutual coupling, measuring below −40 dB, with envelope correlation coefficients of 0.00010, diversity gains nearing 10 dB, and a channel loss capacity of 0.11 bit/s/Hz across the frequency spectrum under investigation. Experimental evaluations have confirmed these improvements, establishing the proposed design as a robust candidate suitable for a wide range of millimeter-wave communication systems. Full article
Show Figures

Figure 1

Figure 1
<p>Single-element antenna: (<b>a</b>) front view, (<b>b</b>) back view, and (<b>c</b>) patch slot, inset, and stub dimensions.</p>
Full article ">Figure 2
<p>Parametric sweep of (<b>a</b>) L<sub>inset</sub> and (<b>a</b>) L<sub>stub</sub> of proposed antenna design.</p>
Full article ">Figure 3
<p>Computed reflection coefficient of single-port antenna versus frequency. The dashed line highlights the −10 dB borderline and the highlighted area represents the bandwidth range, in which the S11 parameter is less than −10 dB.</p>
Full article ">Figure 4
<p>Computed peak gain of single-port antenna versus frequency.</p>
Full article ">Figure 5
<p>(<b>a</b>) Front view and (<b>b</b>) back view of 4-Port MIMO antenna.</p>
Full article ">Figure 6
<p>Unit cell MTM in (<b>a</b>) top and (<b>b</b>) bottom views and its extracted parameters: (<b>c</b>) ε<sub>r</sub>, (<b>d</b>) µ<sub>r</sub>, (<b>e</b>) n. (<b>f</b>) Dispersive diagram exhibiting EM bandgaps detected in the unit cell structure and (<b>g</b>) current distribution resulting from excitation of port 3.</p>
Full article ">Figure 7
<p>Simulated S-parameters versus frequency of the proposed MIMO antenna.</p>
Full article ">Figure 8
<p>Variation in (<b>a</b>) ECC and (<b>b</b>) DG of 4-port MIMO antennas versus frequency.</p>
Full article ">Figure 9
<p>Radiation patterns of 4-port MIMO antenna: (<b>a</b>) E-plane, (<b>b</b>) H-plane at 28 GHz.</p>
Full article ">Figure 10
<p>(<b>a</b>) Measurement setup of reflection coefficient of individual antenna port, (<b>b</b>) measured reflection coefficient of two ports, and (<b>c</b>) setup of measuring mutual coupling between each port pair.</p>
Full article ">Figure 11
<p>Measured and simulated reflection coefficients for each MIMO port, S<sub>11</sub>, S<sub>22</sub>, S<sub>33</sub>, and S<sub>44</sub>.</p>
Full article ">Figure 12
<p>Simulated and measured S-parameters of 4-port MIMO antenna.</p>
Full article ">
Back to TopTop