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Search Results (580)

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20 pages, 22092 KiB  
Article
Design of Shared-Aperture Base Station Antenna with a Conformal Radiation Pattern
by Changpeng Ji, Xin Ning and Wei Dai
Electronics 2025, 14(2), 225; https://doi.org/10.3390/electronics14020225 - 7 Jan 2025
Viewed by 397
Abstract
Aiming at solving the problem of radiation pattern distortion caused by coupling between antennas in different frequency bands in traditional shared-aperture base station array antennas, a new shared-aperture array antenna integrating high-frequency filtering units and medium-frequency electromagnetic transparent antenna units is proposed. Without [...] Read more.
Aiming at solving the problem of radiation pattern distortion caused by coupling between antennas in different frequency bands in traditional shared-aperture base station array antennas, a new shared-aperture array antenna integrating high-frequency filtering units and medium-frequency electromagnetic transparent antenna units is proposed. Without adding additional decoupling structures, it is possible to effectively reduce the coupling of different frequencies, while weakening common-mode and scattering interferences, making the radiation pattern conformal. The array consists of an electromagnetic transparent antenna unit in the medium-frequency (1.71–2.70 GHz) band and four filtering antenna units in the high-frequency (3.30–3.70 GHz) band. The four high-frequency antenna units form two 2 × 1 linear arrays arranged on both sides of the medium-frequency antenna unit and share a reflector. The simulation and measurement results show that the voltage standing wave ratio (VSWR) in the working frequency band is less than 1.50, the average gain in the medium-frequency band is 8.80 dBi, the average gain in the high-frequency band is 12.20 dBi, and the radiation pattern is normal. It is suitable for the field of shared-aperture base station antennas. Full article
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Figure 1

Figure 1
<p>Schematic diagram of different frequency coupling methods.</p>
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<p>High-frequency antenna unit structure (unit: mm).</p>
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<p>High-frequency unit design process.</p>
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<p>Filtering effect comparison.</p>
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<p>Comparison of simulation results of antenna 3.</p>
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<p>Comparison of VSWR and isolation between antenna 3 and antenna 4.</p>
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<p>Current vector diagram at the center frequency of the filter band.</p>
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<p>High-frequency antenna unit measure.</p>
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<p>High-frequency antenna unit VSWR, isolation measure, and simulation results.</p>
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<p>High-frequency antenna unit gain simulation and measure results.</p>
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<p>Medium-frequency antenna unit structure (unit: mm).</p>
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<p>Antenna 1.</p>
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<p>Antenna electromagnetic transparent effect design process.</p>
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<p>RCS simulation result comparison.</p>
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<p>Electric field distribution of high-frequency unit.</p>
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<p>Current distribution.</p>
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<p>Medium-frequency antenna unit measure.</p>
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<p>Medium-frequency antenna unit VSWR, isolation measure, and simulation results.</p>
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<p>Medium-frequency antenna unit gain simulation and measure results.</p>
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<p>Antenna array measure.</p>
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<p>Current distribution of shared-aperture array.</p>
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<p>Array VSWR, isolation simulation, and measure results.</p>
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<p>Array gain and radiation efficiency.</p>
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<p>Comparison of medium-frequency radiation patterns.</p>
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<p>Comparison of high-frequency radiation patterns.</p>
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20 pages, 45658 KiB  
Article
Design and Modeling of a Reconfigurable Multiple Input, Multiple Output Antenna for 24 GHz Radar Sensors
by Mahmoud Shaban
Modelling 2025, 6(1), 2; https://doi.org/10.3390/modelling6010002 - 6 Jan 2025
Viewed by 403
Abstract
A frequency-reconfigurable MIMO antenna with high gain, low mutual coupling and highly suppressed side lobe level (SLL) for applications in 24 GHz ISM band sensing and automotive radar systems was designed, modeled, and simulated. The reconfigurability feature was modeled with the implementation of [...] Read more.
A frequency-reconfigurable MIMO antenna with high gain, low mutual coupling and highly suppressed side lobe level (SLL) for applications in 24 GHz ISM band sensing and automotive radar systems was designed, modeled, and simulated. The reconfigurability feature was modeled with the implementation of a varactor diode in the model to alter the frequency in a wide band around 24 GHz. The design features 2- and 4-port MIMO antenna each comprising a 1 × 8 microstrip patch array. At the core of achieving both a high gain of 16 dBi and high isolation of 38.4 dB at a resonance frequency of 24.120 GHz lies the integration of a metamaterial absorber, comprising an optimized split-ring unit cell to effectively mitigate interference among the MIMO elements. Noteworthy impedance bandwidths of the sensor antenna span from 23.8 to 24.3 GHz, catering to diverse frequency requirements. The proposed sensor antenna feature a half-power beamwidth of 74° in the E-plane and 11° in the H-plane and an SLL of −24 dB at 24.120 GHz showing its robust performance characteristics across multiple operational dimensions. Full article
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Figure 1

Figure 1
<p>(<b>a</b>) Antenna array geometry with SMA connector, and (<b>b</b>) dimensions of the array as given in <a href="#modelling-06-00002-t001" class="html-table">Table 1</a>.</p>
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<p>(<b>a</b>) Reflection coefficient of the antenna array, and (<b>b</b>) real and imaginary part of the input impedance of the antenna at the feeding port.</p>
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<p>Influence of varying varactor diode capacitance on (<b>a</b>) S<sub>11</sub> and (<b>b</b>) gain versus frequency.</p>
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<p>Simulated radiation patterns of (<b>a</b>) E-plane, (<b>b</b>) H-plane for the antenna array, and (<b>c</b>) 3D patterns.</p>
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<p>Simulated radiation patterns of (<b>a</b>) E-plane, (<b>b</b>) H-plane for the antenna array, and (<b>c</b>) 3D patterns.</p>
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<p>(<b>a</b>) Unit cell metamaterial and its exciting ports, top and bottom views, and (<b>b</b>) extracted cell parameters, ε, µ, and <span class="html-italic">n</span>.</p>
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<p>Comparison of S<sub>11</sub> and S<sub>21</sub> parameters for 2-port MIMO antennas with different decoupling techniques: (<b>a</b>) λ/2 separation between the two arrays, (<b>b</b>) 2 mm ground plane gap, (<b>c</b>) additional decoupling patches, and (<b>d</b>) split-ring MMT.</p>
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<p>Three-dimensional patterns of 4-port antenna with metamaterial spacer between the MIMO elements.</p>
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<p>Surface current distribution of the MIMO antenna exited from (<b>a</b>) port 1, (<b>b</b>) port 2, (<b>c</b>) port 3, and (<b>d</b>) port 4.</p>
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<p>Simulated S-parameter variations versus frequency of 4-port MIMO antenna with metamaterial decoupling structure.</p>
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<p>(<b>a</b>) ECC and (<b>b</b>) DG versus frequency for 4-port MIMO antenna.</p>
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14 pages, 12507 KiB  
Article
Broadband Millimeter-Wave Front-End Module Design Considerations in FD-SOI CMOS vs. GaN HEMTs
by Clint Sweeney, Donald Y. C. Lie, Jill C. Mayeda and Jerry Lopez
Appl. Sci. 2024, 14(23), 11429; https://doi.org/10.3390/app142311429 - 9 Dec 2024
Viewed by 617
Abstract
Millimeter-wave (mm-Wave) phased array systems need to meet the transmitter (Tx) equivalent isotropic radiated power (EIRP) requirement, and that depends mainly on the design of two key sub-components: (1) the antenna array and (2) the Tx power amplifier (PA) in the front-end-modules (FEMs). [...] Read more.
Millimeter-wave (mm-Wave) phased array systems need to meet the transmitter (Tx) equivalent isotropic radiated power (EIRP) requirement, and that depends mainly on the design of two key sub-components: (1) the antenna array and (2) the Tx power amplifier (PA) in the front-end-modules (FEMs). Simulations using an electromagnetic (EM) solver carried out in Cadence AWR with AXIEM suggest that for two uniform square patch antenna arrays at 24 GHz, the 4 element array has ~6 dB lower antenna gain and twice the half power beam width (HPBW) compared to the 16 element array. We also present measurements and post-layout parasitic-extracted (PEX) EM simulation data taken on two broadband mm-Wave PAs designed in our lab that cover the key portions of the fifth-generation (5G) FR2-band (i.e., 24.25–52.6 GHz) that lies between the super-high-frequency (SHF, i.e., 3–30 GHz) band and the extremely-high-frequency (EHF, i.e., 30–300 GHz) band: one designed in a 22 nm fully depleted silicon on insulator (FD-SOI) CMOS process, and the other in an advanced 40 nm Gallium Nitride (GaN) high-electron-mobility transistor (HEMT) process. The FD-SOI PA achieves saturated output power (POUT,SAT) of ~14 dBm and peak power-added efficiency (PAE) of ~20% with ~14 dB of gain and 3 dB bandwidth (BW) from ~19.1 to 46.5 GHz in measurement, while the GaN PA achieves measured POUT,SAT of ~24 dBm and peak PAE of ~20% with ~20 dB gain and 3 dB BW from ~19.9 to 35.2 GHz. The PAs’ measured data are in good agreement with the PEX EM simulated data, and 3rd Watt-level GaN PA design data are also presented, but with simulated PEX EM data only. Assuming each antenna element will be driven by one FEM and each phased array targets the same 65 dBm EIRP, millimeter wave (mm-Wave) antenna arrays using the Watt-level GaN PAs and FEMs are expected to achieve roughly 2× wider HPBW with 4× reduction in the array size compared with the arrays using Si FEMs, which shall alleviate the thorny mm-Wave line-of-sight (LOS)-blocking problems significantly. Full article
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Figure 1

Figure 1
<p>System diagram depicting a phased array transceiver system, with monolithic microwave-integrated circuit (MMIC) FEMs highlighted on the left.</p>
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<p>Various circuit and system simulation tool options and set-ups for antenna array and PA design available to us at TTU.</p>
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<p>Uniform square antenna array simulations in Cadence AWR showing the antenna radiation patterns for (<b>a</b>) 2 × 2 and (<b>b</b>) 4 × 4 patch antenna arrays with spacing = λ/2, designed to work at 24 GHz using PECs, a dielectric with <span class="html-italic">ξ<sub>R</sub></span> = 2.2, and an infinite ground plane.</p>
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<p>This figure shows the differential cascode/stacked Si CMOS PA’s (<b>a</b>) schematic and die micrograph along with its measured vs. PEX EM-simulated (<b>b</b>) S-parameters and (<b>c</b>) summarized large-signal performance using <span class="html-italic">V<sub>DD</sub></span> = 1.8 V for ~0.22 mA/μm current density on each of the devices.</p>
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<p>This figure shows the medium-power CS/2-stack GaN PA’s (<b>a</b>) schematic and die micrograph along with its measured vs. PEX EM-simulated (<b>b</b>) S-parameters and (<b>c</b>) summarized large-signal performance using <span class="html-italic">V<sub>D1</sub></span> = 6 V and <span class="html-italic">V<sub>D2</sub></span> = 20 V for ~0.22 mA/μm current density on the device M1 and ~0.36 mA/μm current density for devices M2 and M3.</p>
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<p>This figure shows the Watt-level power CS/2-stack GaN PA’s (<b>a</b>) schematic and die micrograph along with its PEX EM-simulated (<b>b</b>) S-parameters and (<b>c</b>) summarized large-signal performance using <span class="html-italic">V<sub>D1</sub>/V<sub>D2</sub></span> = 10/20 for ~0.19 mA/μm current density on the device M1 and ~0.08 mA/μm current density for devices M2 and M3.</p>
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<p>Summary plots showing (<b>a</b>) <span class="html-italic">P<sub>OUT,SAT</sub></span> and (<b>b</b>) PAE taken from CW large-signal measurements on the CMOS PA and the medium-power GaN PA are shown with PEX EM simulations taken on the Watt-level GaN PA.</p>
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<p>This figure shows (<b>a</b>) <span class="html-italic">OP<sub>1dB</sub></span> for the PAs designed in our lab and (<b>b</b>) the min. <span class="html-italic">P<sub>OUT,SAT</sub></span> of Tx FEMs using our PAs with a flat 1.5 dB estimated RF switch loss, plotted along with Si and GaN FEMs in the literature [<a href="#B25-applsci-14-11429" class="html-bibr">25</a>,<a href="#B26-applsci-14-11429" class="html-bibr">26</a>,<a href="#B28-applsci-14-11429" class="html-bibr">28</a>,<a href="#B33-applsci-14-11429" class="html-bibr">33</a>,<a href="#B34-applsci-14-11429" class="html-bibr">34</a>,<a href="#B35-applsci-14-11429" class="html-bibr">35</a>,<a href="#B36-applsci-14-11429" class="html-bibr">36</a>].</p>
Full article ">Figure 9
<p>Antenna array gain plotted with <span class="html-italic">P<sub>OUT,AVE</sub>/total</span> of an <span class="html-italic">N</span>-element <span class="html-italic">2-D</span> uniform square <math display="inline"><semantics> <mrow> <msqrt> <mi>N</mi> </msqrt> <mo>×</mo> <msqrt> <mi>N</mi> </msqrt> </mrow> </semantics></math> array assuming spacing between the elements is <span class="html-italic">d</span> = λ/2 with the beam pointing towards broadside plotted, targeting a Tx EIRP of 65 dBm (antenna gain per element = 5 dBi; one FEM per element).</p>
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<p>HPBW plotted with <span class="html-italic">P<sub>OUT,AVE</sub>/element</span> of an <span class="html-italic">N</span>-element <span class="html-italic">2-D</span> uniform square <math display="inline"><semantics> <mrow> <msqrt> <mi>N</mi> </msqrt> <mo>×</mo> <msqrt> <mi>N</mi> </msqrt> </mrow> </semantics></math> array assuming spacing between the elements is <span class="html-italic">d</span> = λ/2 with the beam pointing towards broadside plotted, targeting a Tx EIRP of 65 dBm (antenna gain per element = 5 dBi; one FEM per element).</p>
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13 pages, 8436 KiB  
Article
Series-Fed Microstrip Patch Antenna Array with Additive-Manufactured Foldable Honeycomb-Shaped Substrate
by Sima Noghanian, Yi-Hsiang Chang, Patricio Guerron and Reena Dahle
Micromachines 2024, 15(12), 1449; https://doi.org/10.3390/mi15121449 - 29 Nov 2024
Viewed by 619
Abstract
This paper presents a novel foldable S-band microstrip patch antenna array operating in the 2.4–2.45 GHz band. The substrate is designed to allow the array to be folded and arranged in tiles, forming a versatile, reconfigurable antenna array. Additive manufacturing is used to [...] Read more.
This paper presents a novel foldable S-band microstrip patch antenna array operating in the 2.4–2.45 GHz band. The substrate is designed to allow the array to be folded and arranged in tiles, forming a versatile, reconfigurable antenna array. Additive manufacturing is used to fabricate the substrate for ease of fabrication and flexibility in its design. The major challenge in this type of design is creating a proper method of feeding the elements while maintaining the array’s optimal performance. A novel hinge design that can hold a coaxial cable for the series-fed array is introduced. The hinge provides the capability of folding the array from a flat orientation into various folded orientations. In this paper, a 2 × 1 microstrip array unit is presented as proof of concept. The antenna was fabricated and measured, and the results of the measurements are in close agreement with the simulations. The antenna can provide a gain as high as 7.72 dBi in flat conditions. Full article
(This article belongs to the Special Issue Exploring the Potential of 5G and Millimeter-Wave Array Antennas)
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Figure 1

Figure 1
<p>A honeycomb shape is suitable for creating 2D arrays.</p>
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<p>Circular patch two-element series-fed array designed for 2.45 GHz center frequency, (<b>a</b>) overall design, (<b>b</b>) coaxial cable connection, and (<b>c</b>) hinge holding the coax.</p>
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<p>(<b>a</b>) Mechanical model of two-element antenna array, (<b>b</b>) fabricated antenna in flat orientation, (<b>c</b>) mechanical model of hinge and transmission line transition, and (<b>d</b>) fabricated antenna in 360° fold position.</p>
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<p>(<b>a</b>) Dimensions of honeycomb-shaped substrate and (<b>b</b>) dimensions of microstrip patch series-fed array and coaxial connection.</p>
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<p>Comparison of the reflection coefficient of the series-fed 2-element patch array on a hexagonal substrate.</p>
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<p>Simulated and measured radiated patterns at 2.24 GHz: (<b>a</b>) XOZ plane; (<b>b</b>) YOZ plane.</p>
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<p>Simulated 3D realized gain at 2.24 GHz.</p>
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<p>(<b>a</b>) Surface current distribution on the patches at 2.24 GHz; (<b>b</b>) radiation pattern φ and θ components on φ = 45°.</p>
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<p>Inset-fed slotted 2-element honeycomb patch array slot dimensions.</p>
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<p>Measured vs. simulated reflection coefficients of the inset feed slotted patch antenna array.</p>
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<p>(<b>a</b>) Fabricated slotted patch antenna placed on a 3D-printed platform for measurements under flat and bent conditions; (<b>b</b>) flat and bent condition measurements in the anechoic chamber.</p>
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<p>Simulated and measured radiation patterns of the inset-fed 2-element slotted patch at 2.4 GHz: (<b>a</b>) φ = 0°; (<b>b</b>) φ = 90°.</p>
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<p>Simulated and measured radiation patterns of the inset-fed 2-element slotted patch at 2.45 GHz: (<b>a</b>) φ = 0°; (<b>b</b>) φ = 90°.</p>
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<p>Measured radiation pattern at φ = 0° for various folding conditions. Dashed lines are the cross-polarization levels, and solid lines are the polarization patterns: (<b>a</b>) 2.4 GHz; (<b>b</b>) 2.45 GHz.</p>
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<p>Measured reflection coefficients for various folding angles.</p>
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<p>Measured (<b>a</b>) gain and (<b>b</b>) efficiency at 2.4 and 2.45 GHz vs. folding angle.</p>
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<p>Surface current distribution of the slotted array at (<b>a</b>) 2.4 GHz and (<b>b</b>) 2.45 GHz.</p>
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17 pages, 3976 KiB  
Article
Numerical Investigation and Design Curves for Thinned Planar Antenna Arrays for 5G and 6G
by Daniele Pinchera, Fulvio Schettino, Mario Lucido, Gaetano Chirico and Marco Donald Migliore
Electronics 2024, 13(23), 4711; https://doi.org/10.3390/electronics13234711 - 28 Nov 2024
Viewed by 610
Abstract
We numerically investigate the relationship between the main parameters of thinned antenna arrays using a specifically designed evolutionary algorithm, the Multi-Objective Pareto Evolution for Thinning (MOPET). We provide some useful results that allow for the assessment of the achievable performance of antenna arrays [...] Read more.
We numerically investigate the relationship between the main parameters of thinned antenna arrays using a specifically designed evolutionary algorithm, the Multi-Objective Pareto Evolution for Thinning (MOPET). We provide some useful results that allow for the assessment of the achievable performance of antenna arrays and help researchers and practitioners design radar, 5G, and 6G systems. In particular, our approach allows us to quantify the advantage of thinned arrays with respect to traditional equispaced arrays (EA); as an example, using the same number of radiators, we can obtain the same directivity of an EA with a reduction in the side-lobe level (SLL) of more than 10dB, or increase the directivity of a couple of dB maintaining the same SLL of the EA, or get a combination of the two improvements. Moreover, the advantage of thinned architectures with respect to standard EA seems to improve with the increase in the dimension of the array. Full article
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Figure 1

Figure 1
<p>Scheme of some popular antenna array architectures. (<b>a</b>) Clustered/subarray architecture, where groups (marked in yellow) of radiators (marked in red) are fed by a single T/R module. (<b>b</b>) Sparse architectures, where the position of the radiators on the plane is freely selectable. (<b>c</b>) Thinned architectures, where the position of the radiators is selected starting from a regular grid.</p>
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<p>Scheme of the planar antenna array on the <math display="inline"><semantics> <mrow> <mo>(</mo> <mi>x</mi> <mo>,</mo> <mi>y</mi> <mo>)</mo> </mrow> </semantics></math> plane and of the coordinate systems employed. Ellipses with different colors represent the radiators according to their activation (red: active, brown: non-active).</p>
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<p>Graphical representation of the grid centering. On the left side, we can see the binary grid describing the excitation of the thinned array before the centering. In the right grid, the group of “1” of the binary grid representing the turned-on elements is shifted in order to have such elements centered on the overall grid. The red color of the “1” is used to emphasize the shift.</p>
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<p>Graphical description of the Thinning-Rate-Preserving Crossover. A “XOR” operation is performed between two binary sequences (A,B) to identify their common bits (the position of the common bits is emphasized by means of cyan rectangles). A temporary trinary vector T is then generated (the “X” represents non-common bits of the two binary sequences to cross). Then, a random binary vector D, of the same length as the number of “X” of T, is generated; this sequence must have the same number of ones and zeros. Finally, the crossover vector C is obtained from T, substituting the “X” with the elements of D. Vertical arrows are used to show the origin of each bit.</p>
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<p>PB found with MOPET for the case of a rectangular antenna with 128 elements. Some reference solutions are compared to the Pareto curve.</p>
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<p>Analysis of Solution A from <a href="#electronics-13-04711-f005" class="html-fig">Figure 5</a>. (<b>a</b>) Antenna layout, with the red circles representing the 128 used elements. (<b>b</b>) Normalized radiation pattern represented in the <math display="inline"><semantics> <mrow> <mo>(</mo> <mi>u</mi> <mo>,</mo> <mi>v</mi> <mo>)</mo> </mrow> </semantics></math> plane.</p>
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<p>Analysis of Solution B from <a href="#electronics-13-04711-f005" class="html-fig">Figure 5</a>. (<b>a</b>) Antenna layout, with the red circles representing the 128 used elements. (<b>b</b>) Normalized radiation pattern represented in the <math display="inline"><semantics> <mrow> <mo>(</mo> <mi>u</mi> <mo>,</mo> <mi>v</mi> <mo>)</mo> </mrow> </semantics></math> plane.</p>
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<p>PB found with MOPET for the case of a rectangular antenna with 124 elements. Some reference solutions are compared to the Pareto curve.</p>
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<p>Analysis of Solution C from <a href="#electronics-13-04711-f008" class="html-fig">Figure 8</a>. (<b>a</b>) Antenna layout, with the red circles representing the 124 used elements. (<b>b</b>) Normalized radiation pattern represented in the <math display="inline"><semantics> <mrow> <mo>(</mo> <mi>u</mi> <mo>,</mo> <mi>v</mi> <mo>)</mo> </mrow> </semantics></math> plane.</p>
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<p>PBs for the square <math display="inline"><semantics> <mrow> <msub> <mi>M</mi> <mrow> <mi>o</mi> <mi>n</mi> </mrow> </msub> <mo>=</mo> <mn>64</mn> </mrow> </semantics></math> case, compared to the reference solution.</p>
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<p>PBs for the square <math display="inline"><semantics> <mrow> <msub> <mi>M</mi> <mrow> <mi>o</mi> <mi>n</mi> </mrow> </msub> <mo>=</mo> <mn>128</mn> </mrow> </semantics></math> case, compared to the reference solution.</p>
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<p>PBs for the square <math display="inline"><semantics> <mrow> <msub> <mi>M</mi> <mrow> <mi>o</mi> <mi>n</mi> </mrow> </msub> <mo>=</mo> <mn>256</mn> </mrow> </semantics></math> case, compared to the reference solution.</p>
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<p>PB curves for the <math display="inline"><semantics> <mrow> <msub> <mi>M</mi> <mrow> <mi>o</mi> <mi>n</mi> </mrow> </msub> <mo>=</mo> <mrow> <mo>{</mo> <mn>64</mn> <mo>,</mo> <mn>128</mn> <mo>,</mo> <mn>256</mn> <mo>}</mo> </mrow> </mrow> </semantics></math> broadside beam-only cases.</p>
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<p>PB curves for the <math display="inline"><semantics> <mrow> <msub> <mi>M</mi> <mrow> <mi>o</mi> <mi>n</mi> </mrow> </msub> <mo>=</mo> <mrow> <mo>{</mo> <mn>64</mn> <mo>,</mo> <mn>128</mn> <mo>,</mo> <mn>256</mn> <mo>}</mo> </mrow> </mrow> </semantics></math> large scanning cases.</p>
Full article ">Figure 15
<p>PB found with MOPET for the case of a rectangular antenna with 1024 elements. Some reference solutions are compared to the Pareto curve.</p>
Full article ">Figure 16
<p>Analysis of Solution D from <a href="#electronics-13-04711-f015" class="html-fig">Figure 15</a>. (<b>a</b>) Antenna layout, with the red circles representing the 1024 used elements. (<b>b</b>) Normalized radiation pattern represented in the <math display="inline"><semantics> <mrow> <mo>(</mo> <mi>u</mi> <mo>,</mo> <mi>v</mi> <mo>)</mo> </mrow> </semantics></math> plane.</p>
Full article ">Figure 17
<p>Analysis of Solution E from <a href="#electronics-13-04711-f015" class="html-fig">Figure 15</a>. (<b>a</b>) Antenna layout, with the red circles representing the 1024 used elements. (<b>b</b>) Normalized radiation pattern represented in the <math display="inline"><semantics> <mrow> <mo>(</mo> <mi>u</mi> <mo>,</mo> <mi>v</mi> <mo>)</mo> </mrow> </semantics></math> plane.</p>
Full article ">
14 pages, 2855 KiB  
Article
A Wide-Angle and PON Fully Polarimetric Retrodirective Array at the X Band
by Shuangdi Zhao, Lei Chen, Jicheng Pan and Tianling Zhang
Micromachines 2024, 15(12), 1418; https://doi.org/10.3390/mi15121418 - 26 Nov 2024
Viewed by 504
Abstract
A new type of fully polarimetric retrodirective array (RDA) using a PON-type structure is proposed in this paper. The fully polarimetric property is the result of the proposed phase conjugation circuits, which perform phase conjugation processing on the x, y, and z polarization [...] Read more.
A new type of fully polarimetric retrodirective array (RDA) using a PON-type structure is proposed in this paper. The fully polarimetric property is the result of the proposed phase conjugation circuits, which perform phase conjugation processing on the x, y, and z polarization electric field components of the incident wave when combined with a tri-polarized antenna array. It enables the retrodirective array to receive and retransmit an arbitrary polarized incident wave. The measured results of the monostatic radar cross-section (RCS) show that the −5 dB beam width of the array was greater than 95° at 9.6 GHz for different polarized incident waves. Furthermore, the proposed RDA has better retrodirectivity performance on arbitrary polarized incident waves when using a wide-beam antenna, and if we further incorporate modulation and demodulation into the circuits, it has the potential to be applied to the wireless communications field. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
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<p>The model of the fully Polarimetric retrodirective array system.</p>
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<p>Schematic of an element in the fully polarimetric RDA based on a PON-type structure.</p>
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<p>Structure of the tri-polarized antenna. (<b>a</b>) Top view. (<b>b</b>) Side view. (L1 = 19 mm, L2 = 7.8 mm, L3 = 6 mm, d1 = 2 mm, d2 = 0.51 mm, d3 = 0.5 mm, dv = 2 mm, s = 0.5 mm, W1 = 2.5 mm, h = 1.5 mm, hm = 7.5 mm, Lf = 3.5 mm).</p>
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<p>Photograph of the fully polarimetric phase conjugate circuit.</p>
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<p>Schematic of an element in the fully polarimetric RDA based on a PON-type structure.</p>
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<p>Measurement in the anechoic chamber.</p>
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<p>Incident plane wave with different polarizations.</p>
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<p>Measured (@9.61 GHz) monostatic RCS.</p>
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11 pages, 6399 KiB  
Article
A Ku-Band Compact Offset Cylindrical Reflector Antenna with High Gain for Low-Earth Orbit Sensing Applications
by Bashar A. F. Esmail, Dustin Isleifson and Lotfollah Shafai
Sensors 2024, 24(23), 7535; https://doi.org/10.3390/s24237535 - 26 Nov 2024
Viewed by 520
Abstract
The rise of CubeSats has unlocked opportunities for cutting-edge space missions with reduced costs and accelerated development timelines. CubeSats necessitate a high-gain antenna that can fit within a tightly confined space. This paper is primarily concerned with designing a compact Ku-band offset cylindrical [...] Read more.
The rise of CubeSats has unlocked opportunities for cutting-edge space missions with reduced costs and accelerated development timelines. CubeSats necessitate a high-gain antenna that can fit within a tightly confined space. This paper is primarily concerned with designing a compact Ku-band offset cylindrical reflector antenna for a CubeSat-based Earth Observation mission, with the goal of monitoring Arctic snow and sea ice. The development of a Ku-band offset cylindrical reflector, with a compact aperture of 110 × 149 mm2 (6.3λ × 8.5λ), is described alongside a patch array feed consisting of 2 × 8 elements. The patch array feed is designed using a lightweight Rogers substrate and is utilized to test the reflector. Adopting an offset configuration helped prevent gain loss due to feed blockage. Analyzing the reflector antenna, including the feed, thorough simulations and measurements indicates that achieving a gain of 25 dBi and an aperture efficiency of 52% at 17.2 GHz is attainable. The reflector’s cylindrical shape and compact size facilitate the design of a simple mechanism for reflector deployment, enabling the antenna to be stored within 1U. The array feed and reflector antenna have been fabricated and tested, demonstrating good consistency between the simulation and measurement outcomes. Full article
(This article belongs to the Section Remote Sensors)
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<p>6U CubeSat employing the proposed deployable cylindrical reflector antenna. The antenna is designed to fit within 1U (10 × 10 × 10 cm<sup>3</sup>) [<a href="#B24-sensors-24-07535" class="html-bibr">24</a>].</p>
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<p>The configuration of the 2 × 8 array feed.</p>
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<p>The fabricated array feed.</p>
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<p>The simulated and measured reflection coefficients of the feed.</p>
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<p>Radiation pattern measurement setup.</p>
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<p>The simulated and measured radiation patterns of the feed at 17.2 GHz: (<b>a</b>) E-plane and (<b>b</b>) H-plane.</p>
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<p>The offset cylindrical reflector system: (<b>a</b>) optical specifications for the offset-fed cylindrical reflector with the key parameters (the dimensions are <span class="html-italic">W</span><sub>r</sub> = 110 mm, <span class="html-italic">L</span><sub>r</sub> = 149 mm, <span class="html-italic">F</span>= 98 mm, and <span class="html-italic">H</span> = 13 mm) and (<b>b</b>) the cross-section of the reflector system.</p>
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<p>The normalized E- and H-plane radiation patterns at 17.2 GHz.</p>
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<p>(<b>a</b>) The antenna system prototype includes the cylindrical reflector and the array feed and (<b>b</b>) the reflector system in the antenna test facility.</p>
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<p>Measured and calculated radiation patterns of the cylindrical reflector at 17.2 GHz: (<b>a</b>) E-plane and (<b>b</b>) H-plane.</p>
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<p>The deployment transitions through the stages of (<b>a</b>) fully stowed, (<b>b</b>) and (<b>c</b>) mid-deployment, and (<b>d</b>) fully deployed.</p>
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14 pages, 5548 KiB  
Article
Phased Array Antenna Calibration Based on Autocorrelation Algorithm
by Xuan Luong Nguyen, Nguyen Trong Nhan, Thanh Thuy Dang Thi, Tran Van Thanh, Phung Bao Nguyen and Nguyen Duc Trien
Sensors 2024, 24(23), 7496; https://doi.org/10.3390/s24237496 - 24 Nov 2024
Viewed by 718
Abstract
The problem of calibrating phased array antennas in a noisy environment using an autocorrelation algorithm is investigated and a mathematical model of the autocorrelation calibration method is presented. The proposed calibration system is based on far-field scanning of the phased array antenna in [...] Read more.
The problem of calibrating phased array antennas in a noisy environment using an autocorrelation algorithm is investigated and a mathematical model of the autocorrelation calibration method is presented. The proposed calibration system is based on far-field scanning of the phased array antenna in an environment with internal noise and external interference. The proposed method is applied to a phased array antenna and compared with traditional rotating-element electric-field vector methods, which involve identifying the maximum and minimum vector–sum points (REVmax and REVmin, respectively). The proposed calibration system is verified for a phased array antenna at 3 GHz. Experimental verification of the mathematical model of the proposed method demonstrates that the autocorrelation method is more accurate than the rotating-element electric-field vector methods in determining the amplitude and phase shifts. The measured peak gain of the combined beam in the E-plane increased from 7.83 to 8.37 dB and 3.57 to 4.36 dB compared to the REVmax and REVmin methods, respectively, and the phase error improved from 47° to 55.48° and 19.43° to 29.16°, respectively. The proposed method can be considered an effective solution for large-scale phase calibration at both in-field and in-factory levels, even in the presence of external interference. Full article
(This article belongs to the Section Communications)
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<p>A Schematic diagram of the PAA calibration method based on the autocorrelation algorithm.</p>
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<p>The simulation environment of the calibration showing reference (orange), fixed (green), and calibrated (blue) antennas and interference source (red): 2D<sup>2</sup>/λ = far-field distance in meters; λ = wavelength of the radiating wave in meters.</p>
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<p>The radiation patterns of the combined beam in the E-plane when using different calibration methods: autocorrelation method—red; REVmin method—green; REVmax method—blue.</p>
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<p>The calibration system diagram for the REVmax, REVmin, and autocorrelation methods.</p>
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<p>The (<b>a</b>) signal processing block, (<b>b</b>) receiving antenna, and (<b>c</b>) reference (transmitting) antenna.</p>
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<p>The software graphical user interface for automatic calibration control.</p>
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<p>A diagram of the calibration system that produced the radiation patterns after performing the amplitude and phase shifts of the received signals using the REVmax, REVmin, and autocorrelation methods.</p>
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<p>The radiation patterns of the combined beam in the E-plane when using different calibration methods in the cases (<b>a</b>) <math display="inline"><semantics> <mrow> <mi>D</mi> <mo>=</mo> <mn>0.625</mn> <mi>λ</mi> </mrow> </semantics></math> and <span class="html-italic">φ<sub>tr</sub>.</span> = 0°; (<b>b</b>) <math display="inline"><semantics> <mrow> <mi>D</mi> <mo>=</mo> <mn>1.25</mn> <mi>λ</mi> </mrow> </semantics></math> and <span class="html-italic">φ<sub>tr</sub>.</span> = 0°; (<b>c</b>) <math display="inline"><semantics> <mrow> <mi>D</mi> <mo>=</mo> <mn>0.625</mn> <mi>λ</mi> </mrow> </semantics></math> and <span class="html-italic">φ<sub>tr</sub>.</span> = 240°; and (<b>d</b>) <math display="inline"><semantics> <mrow> <mi>D</mi> <mo>=</mo> <mn>1.25</mn> <mi>λ</mi> </mrow> </semantics></math> and <span class="html-italic">φ<sub>tr.</sub></span> = 240°. Autocorrelation method—red; REVmin method—green; REVmax method—blue.</p>
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12 pages, 5737 KiB  
Article
Modeling of 2-D Periodic Array of Dielectric Bars with a Low Reflection Angle for a Wind Tunnel High-Power Microwave Experiment
by Rong Bao, Yang Tao and Yongdong Li
Appl. Sci. 2024, 14(23), 10876; https://doi.org/10.3390/app142310876 - 24 Nov 2024
Viewed by 455
Abstract
Two-dimensional periodic dielectric bars have potential applications in high-power microwave (HPM) radiation effect experiments performed in wind tunnels. Such a bar is designed to consist of two types of dielectric materials, and two lined-up blocks can be considered as a period along the [...] Read more.
Two-dimensional periodic dielectric bars have potential applications in high-power microwave (HPM) radiation effect experiments performed in wind tunnels. Such a bar is designed to consist of two types of dielectric materials, and two lined-up blocks can be considered as a period along the bar. Under plane excitation, the theoretical period length of the beat wave pattern fits well with the simulation result, which requires modifying the previously presented field-matching method. The phase distribution on the cross-section can be non-uniform when two different guiding modes are excited independently and propagate along different materials. Directional reflection with a low reflection angle can be obtained by reasonably choosing the parameters of the dielectric array. The designed array can decrease the returned-back microwave power toward the microwave source by 6 dB according to the numerical simulation, which included the wind tunnel, the input antenna, the test target, and the reflect array in one model. Full article
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<p>Illustration of HPM radiation effect experiment system [<a href="#B1-applsci-14-10876" class="html-bibr">1</a>].</p>
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<p>Illustration of the 2-D dielectric bar reflect array.</p>
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<p>Incident microwave after reflection: (<b>a</b>) <span class="html-italic">E<sub>y</sub></span> component on the <span class="html-italic">x</span>o<span class="html-italic">y</span> plane, (<b>b</b>) power flow on the <span class="html-italic">x</span>o<span class="html-italic">y</span> plane and (<b>c</b>) simulation model of the test target.</p>
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<p>Propagation in the dielectric bar waveguide along the −<span class="html-italic">x</span> direction: (<b>a</b>) illustration of geometry and (<b>b</b>) simulation model using periodic boundary. <span class="html-italic">w</span> is the width of the bars in the <span class="html-italic">y</span> direction; <span class="html-italic">l</span><sub>1</sub> and <span class="html-italic">l</span><sub>2</sub> are the sizes of the high-permittivity and low-permittivity materials, respectively; and <span class="html-italic">ε</span><sub>l</sub> and <span class="html-italic">ε</span><sub>h</sub> are the permittivities of the materials and <span class="html-italic">ε</span><sub>l</sub> &lt; <span class="html-italic">ε</span><sub>h</sub>.</p>
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<p>Electric field distribution under plane wave excitation at different frequencies: (<b>a</b>) electric field at 2 GHz, (<b>b</b>) electric field at 4 GHz, (<b>c</b>) electric field at 6 GHz, (<b>d</b>) field strength of <span class="html-italic">E<sub>y</sub></span> component at 2 GHz, (<b>e</b>) field strength of <span class="html-italic">E<sub>y</sub></span> component at 4 GHz, and (<b>f</b>) field strength of <span class="html-italic">E<sub>y</sub></span> component at 6 GHz.</p>
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<p>Region division for the theoretical analysis.</p>
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<p>Simulated average norm of the electric field in the <span class="html-italic">x</span>o<span class="html-italic">z</span> plane (<b>a</b>) at 8.7 GHz, (<b>b</b>) at 9.2 GHz, and (<b>c</b>) at 9.7 GHz.</p>
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<p>Amplitude simulation model of (<b>a</b>) the array without metal wall and simulated power density distributions at (<b>b</b>) 7.7 GHz, (<b>c</b>) 8.7 GHz, and (<b>d</b>) 9.7 GHz.</p>
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<p>Simulated phase distribution using the model in <a href="#applsci-14-10876-f008" class="html-fig">Figure 8</a> at (<b>a</b>) 7.7 GHz, (<b>b</b>) 8.7 GHz, and (<b>c</b>) 9.7 GHz.</p>
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<p>Radiation pattern of structure in <a href="#applsci-14-10876-f008" class="html-fig">Figure 8</a> with (<b>a</b>) <span class="html-italic">h</span> = 40 mm, (<b>b</b>) <span class="html-italic">h</span> = 45 mm, and (<b>c</b>) <span class="html-italic">h</span> = 50 mm.</p>
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<p>Radiation pattern of the reflect-array structure: (<b>a</b>) simulation model of the reflect array, (<b>b</b>) radiation pattern of the electric field, (<b>c</b>) radiation pattern with ‘phi’ = 0°, (<b>d</b>) radiation pattern with ‘phi’ = 90°, (<b>e</b>) phase distribution on the interface when <span class="html-italic">g</span> is 13.5 mm, and (<b>f</b>) distribution of the amplitude of the <span class="html-italic">E<sub>y</sub></span> component on the interface when <span class="html-italic">g</span> is 17.5 mm.</p>
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<p>Simulation of the experimental setup in a wind tunnel: (<b>a</b>) simulation model, (<b>b</b>) outgoing power vs. frequency without reflect array, and (<b>c</b>) outgoing power vs. frequency with reflect array.</p>
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16 pages, 14961 KiB  
Article
A Sub-6 GHz 8 × 8 MIMO Antenna Array for 5G Metal-Frame Mobile Phone Applications
by Yu-Tung Chen and Hsin-Lung Su
Electronics 2024, 13(23), 4590; https://doi.org/10.3390/electronics13234590 - 21 Nov 2024
Viewed by 557
Abstract
This article introduces a broadband sub-6 GHz 8 × 8 MIMO (multi-input multi-output) antenna array for 5G (fifth-generation) metal-frame mobile phone applications. The unique advantage of this compact antenna design is its placement in the corners of the mobile phone, allowing for significant [...] Read more.
This article introduces a broadband sub-6 GHz 8 × 8 MIMO (multi-input multi-output) antenna array for 5G (fifth-generation) metal-frame mobile phone applications. The unique advantage of this compact antenna design is its placement in the corners of the mobile phone, allowing for significant PCB board space reduction. The proposed antenna’s 6 dB impedance bandwidth ranged from 3.3 to 6 GHz, covering the n77/78/79 and WiFi-5GHz bands. The main radiating element was an open-slot antenna coupled by a T-shaped structure connected to a 50-Ω transmission line. The size of the single-antenna element was 12.25 mm × 2.5 mm × 7 mm, and these antennas were symmetrical at four corners of the smartphone. A wide slot and neutral line were incorporated to reduce mutual coupling between adjacent antennas. The MIMO antenna array achieved isolation above 12 dB. The peak realized gain ranged from 2 to 5.28 dBi, and the total efficiency spanned 37% to 71%. The ECC (envelope correlation coefficient) was less than 0.34, and the CC (channel capacity) ranged from 33 and 41 bps/Hz. Full article
(This article belongs to the Special Issue Broadband Antennas and Antenna Arrays)
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<p>Arrangement of the proposed 8 × 8 MIMO antenna array in a mobile phone.</p>
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<p>The (<b>a</b>) front view, (<b>b</b>) bottom view, and (<b>c</b>) side view of these two adjacent antennas.</p>
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<p>(<b>a</b>) Reflection coefficients and (<b>b</b>) isolations with the neutral line of the proposed MIMO antenna array.</p>
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<p>(<b>a</b>) Design process of the antenna element and (<b>b</b>) the simulated reflection coefficients for Ref.1, Ref.2, and the proposed single-antenna element (prop.-ant. element).</p>
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<p>(<b>a</b>) Design process of the decoupling components. (<b>b</b>) Isolation and its corresponding (<b>c</b>) surface current distributions for the simulated Case1, Case2, and the proposed case.</p>
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<p>(<b>a</b>) Electric field, (<b>b</b>) surface current distribution at 3.3 GHz, (<b>c</b>) electric field, and (<b>d</b>) surface current distribution at 5 GHz of the proposed single-antenna element.</p>
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<p>(<b>a</b>) The geometry of and (<b>b</b>) variation in the reflection coefficient of parameter L1.</p>
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<p>(<b>a</b>) The geometry of and (<b>b</b>) variation in the reflection coefficient of parameter L2.</p>
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<p>(<b>a</b>) The geometry of and (<b>b</b>) variation in the reflection coefficient of parameter W1.</p>
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<p>(<b>a</b>) The geometry of and (<b>b</b>) variation in the isolation of parameter W2.</p>
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<p>The variations in the T-shape feeding structure in (<b>a</b>) step 1, step 2, and the proposed case; and (<b>b</b>) the variation in the input impedance and its corresponding (<b>c</b>) reflection coefficient.</p>
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<p>Photos of the manufactured MIMO antenna array: (<b>a</b>) top view and (<b>b</b>) back view.</p>
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<p>Simulated and measured reflection coefficients for Ant.1, Ant.2, Ant.3, and Ant.4.</p>
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<p>Simulated and measured isolations of the proposed antenna.</p>
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<p>Simulated and measured (<b>a</b>) efficiencies and (<b>b</b>) peak realized gains of the proposed antenna.</p>
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<p>Calculated ECC of the proposed antenna.</p>
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<p>Calculated channel capacity of the proposed antenna is the red line and the black dash line is the ideal CC value.</p>
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<p>The measured and simulated 2D radiation patterns of Ant.1 in the XY, YZ, and XY planes at 3.7, 4.7, and 5.5 GHz.</p>
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<p>The measured and simulated 2D radiation patterns of Ant.2 are in the XY, YZ, and XY planes at 3.7, 4.7, and 5.5 GHz.</p>
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11 pages, 8301 KiB  
Article
A 2-D Fully Polarized Van Atta Array Based on Wide-Beam Tri-Polarized Antennas
by Jicheng Pan, Lei Chen, Shuangdi Zhao and Tianling Zhang
Micromachines 2024, 15(11), 1400; https://doi.org/10.3390/mi15111400 - 20 Nov 2024
Cited by 1 | Viewed by 654
Abstract
This paper proposes a 2-D fully polarized Van Atta array, which consists of four tri-polarized antenna elements. The tri-polarized antenna element comprises a monopole antenna and a low-profile microstrip antenna that widens the beam by folding four electric walls. This configuration enables the [...] Read more.
This paper proposes a 2-D fully polarized Van Atta array, which consists of four tri-polarized antenna elements. The tri-polarized antenna element comprises a monopole antenna and a low-profile microstrip antenna that widens the beam by folding four electric walls. This configuration enables the Van Atta arrays to receive and transmit arbitrarily polarized incident waves over a wider range. The measurement results indicate that the proposed Van Atta array exhibits a −5 dB radar cross-section (RCS) greater than 95° when TE-polarized waves are incident and greater than 134° when TM-polarized waves are incident, significantly surpassing the 2-D dual-polarized array. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
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<p>Schematic diagram of the antenna loaded with electric walls: (<b>a</b>) current relationship between vertical electric walls and main patch; (<b>b</b>) electric wall loading method; (<b>c</b>) principle of beam spreading.</p>
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<p>Antenna loaded with parasitic patches: (<b>a</b>) 3D view; (<b>b</b>) top view.</p>
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<p>Current flow on the surface of the antenna loaded with folded electric walls: (<b>a</b>) 3D view; (<b>b</b>) top view; (<b>c</b>) side view.</p>
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<p>Comparison of simulated results of antenna pattern with or without a folding wall: (<b>a</b>) E-plane pattern; (<b>b</b>) H-plane pattern.</p>
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<p>Geometry of tri-polarized antenna: (<b>a</b>) 3D view; (<b>b</b>) top view; (<b>c</b>) side view (L1 = 19 mm, L2 = 7.8 mm, L3 = 6 mm, r1 = 2 mm, r2 = 0.51 mm, r3 = 0.5 mm, dv = 2 mm, S = 0.5 mm, W1 = 2.5 mm, h = 1.5 mm, hm = 7.5 mm, Lf = 3.5 mm).</p>
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<p>Simulated and measured S-parameters of tri-polarized antennas: (<b>a</b>) reflection coefficients; (<b>b</b>) isolations.</p>
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<p>Simulated and measured radiation patterns at 9.6 GHz of the proposed antenna.</p>
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<p>Prototype of the antenna array: (<b>a</b>) 3D view; (<b>b</b>) top view.</p>
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<p>Reflection coefficients of antenna array elements: (<b>a</b>) Ant. 1; (<b>b</b>) Ant. 2; (<b>c</b>) Ant. 3; (<b>d</b>) Ant. 4.</p>
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<p>Van Atta array: (<b>a</b>) prototype of the Van Atta array; (<b>b</b>) connections for fully polarized planar Van Atta array.</p>
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<p>Schematic of the measurement system for the monostatic RCS.</p>
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<p>Measured monostatic RCS with TM incident wave: (<b>a</b>) tri-polarized antenna; (<b>b</b>) dual-polarized antenna.</p>
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<p>Measured monostatic RCS with TE incident wave: (<b>a</b>) tri-polarized antenna; (<b>b</b>) dual-polarized antenna.</p>
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27 pages, 6231 KiB  
Review
A Review of Unmanned Aerial Vehicle Based Antenna and Propagation Measurements
by Venkat R. Kandregula, Zaharias D. Zaharis, Qasim Z. Ahmed, Faheem A. Khan, Tian Hong Loh, Jason Schreiber, Alexandre Jean René Serres and Pavlos I. Lazaridis
Sensors 2024, 24(22), 7395; https://doi.org/10.3390/s24227395 - 20 Nov 2024
Cited by 1 | Viewed by 1170
Abstract
This paper presents a comprehensive survey of state-of-the-art UAV–based antennas and propagation measurements. Unmanned aerial vehicles (UAVs) have emerged as powerful tools for in situ electromagnetic field assessments due to their flexibility, cost-effectiveness, and ability to operate in challenging environments. This paper highlights [...] Read more.
This paper presents a comprehensive survey of state-of-the-art UAV–based antennas and propagation measurements. Unmanned aerial vehicles (UAVs) have emerged as powerful tools for in situ electromagnetic field assessments due to their flexibility, cost-effectiveness, and ability to operate in challenging environments. This paper highlights various UAV applications, from testing large–scale antenna arrays, such as those used in the square kilometer array (SKA), to evaluating channel models for 5G/6G networks. Additionally, the review discusses technical challenges, such as positioning accuracy and antenna alignment, and it provides insights into the latest advancements in portable measurement systems and antenna designs tailored for UAV use. During the UAV–based antenna measurements, key contributors to the relatively small inaccuracies of around 0.5 to 1 dB are identified. In addition to factors such as GPS positioning errors and UAV vibrations, ground reflections can significantly contribute to inaccuracies, leading to variations in the measured radiation patterns of the antenna. By minimizing ground reflections during UAV–based antenna measurements, errors in key measured antenna parameters, such as HPBW, realized gain, and the front-to-back ratio, can be effectively mitigated. To understand the source of propagation losses in a UAV to ground link, simulations were conducted in CST. These simulations identified scattering effects caused by surrounding buildings. Additionally, by simulating a UAV with a horn antenna, potential sources of electromagnetic coupling between the antenna and the UAV body were detected. The survey concludes by identifying key areas for future research and emphasizing the potential of UAVs to revolutionize antenna and propagation measurement practices to avoid the inaccuracies of the antenna parameters measured by the UAV. Full article
(This article belongs to the Special Issue New Methods and Applications for UAVs)
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<p>Organization of the paper.</p>
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<p>Measurement configuration of the UAV system.</p>
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<p>Conventional elevated slant test range.</p>
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<p>UAV–based in situ measurement for a parabolic reflector antenna system.</p>
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<p>UAV–based measurement for a parabolic reflector antenna system placed on a ship.</p>
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<p>Vertical radiation pattern of a BASTA using a UAV.</p>
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<p>Horizontal radiation pattern of a BASTA using a UAV.</p>
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<p>Common errors in broadcasting systems.</p>
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<p>SixArms airborne measurements for broadcasting systems.</p>
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<p>(<b>a</b>) UAV–based measurements at 720 m and (<b>b</b>) UAV–based measurements at 2025 m.</p>
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<p>UAV–based measurements in radiating near field.</p>
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<p>UAV–based measurements for an array of HF wire biconical antennas.</p>
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<p>UAV with monopole flying over the LPDA array.</p>
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<p>Scattering effect in a semi–urban area simulated in CST.</p>
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<p>Hexacopter carrying cloverleaf wire antenna.</p>
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<p>Scattering effect in an urban area simulated in CST Studio Suite.</p>
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<p>Fields scattered by the UAV body.</p>
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12 pages, 5482 KiB  
Communication
Array Radar Three-Dimensional Forward-Looking Imaging Algorithm Based on Two-Dimensional Super-Resolution
by Jinke Dai, Weijie Sun, Xinrui Jiang and Di Wu
Sensors 2024, 24(22), 7356; https://doi.org/10.3390/s24227356 - 18 Nov 2024
Viewed by 625
Abstract
Radar imaging is a technology that uses radar systems to generate target images. It transmits radio waves, receives the signal reflected back by the target, and realizes imaging by analyzing the target’s position, shape, and motion information. The three-dimensional (3D) forward-looking imaging of [...] Read more.
Radar imaging is a technology that uses radar systems to generate target images. It transmits radio waves, receives the signal reflected back by the target, and realizes imaging by analyzing the target’s position, shape, and motion information. The three-dimensional (3D) forward-looking imaging of missile-borne radar is a branch of radar imaging. However, owing to the limitation of antenna aperture, the imaging resolution of real aperture radar is restricted. By implementing the super-resolution techniques in array signal processing into missile-borne radar 3D forward-looking imaging, the resolution can be further improved. In this paper, a 3D forward-looking imaging algorithm based on the two-dimensional (2D) super-resolution algorithm is proposed for missile-borne planar array radars. In the proposed algorithm, a forward-looking planar array with scanning beams is considered, and each range-pulse cell in the received data is processed one by one using a 2D super-resolution method with the error function constructed according to the weighted least squares (WLS) criterion to generate a group of 2D spectra in the azimuth-pitch domain. Considering the lack of training samples, the super-resolution spectrum of each range-pulse cell is estimated via adaptive iteration processing only with one sample, i.e., the cell under process. After that, all the 2D super-resolution spectra in azimuth-pitch are accumulated according to the changes in instantaneous beam centers of the beam scanning. As is verified by simulation results, the proposed algorithm outperforms the real aperture imaging method in terms of azimuth-pitch resolution and can obtain 3D forward-looking images that are of a higher quality. Full article
(This article belongs to the Special Issue Recent Advances in Radar Imaging Techniques and Applications)
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<p>Distribution of the array elements.</p>
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<p>Angular division of the observed region.</p>
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<p>Signal processing flow.</p>
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<p>Point targets distribution.</p>
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<p>Simulation results of the point targets. (<b>a</b>) Real beam method. (<b>b</b>) 2D-IAA.</p>
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<p>Simulation results in the range profile of <math display="inline"><semantics> <mrow> <mi>R</mi> <mo>=</mo> <mn>1000</mn> <mo> </mo> <mi mathvariant="normal">m</mi> </mrow> </semantics></math>. (<b>a</b>) Real beam method. (<b>b</b>) 2D-IAA.</p>
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<p>Scene for ship target imaging simulation.</p>
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<p>The 3D imaging results of the scene. (<b>a</b>) Real beam method. (<b>b</b>) 2D-OMP. (<b>c</b>) 2D-IAA.</p>
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<p>The projection of the 3D imaging results in x-y plane. (<b>a</b>) Real beam method. (<b>b</b>) 2D-OMP. (<b>c</b>) 2D-IAA.</p>
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<p>The section of the 3D imaging results at <math display="inline"><semantics> <mrow> <mi>x</mi> <mo>=</mo> <mo>−</mo> <mn>25</mn> <mo> </mo> <mi mathvariant="normal">m</mi> </mrow> </semantics></math>. (<b>a</b>) Real beam method. (<b>b</b>) 2D-OMP. (<b>c</b>) 2D-IAA.</p>
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18 pages, 8594 KiB  
Article
A Novel Ground Slot-Based Dual-Band Massive Multiple-Input Multiple-Output (MIMO) Antenna for n47 and n48 Smartphone Applications
by Muhammad Zahid, Muhammad Uzair Akbar, Nirman Bhowmike, Devi Prasanth Bolla, Asad Ullah Talib, Sultan Shoaib, Yasar Amin and Saleem Shahid
Electronics 2024, 13(21), 4296; https://doi.org/10.3390/electronics13214296 - 31 Oct 2024
Viewed by 796
Abstract
A novel ground slot-based massive multiple-input multiple-output (MIMO) system with 14 elements for dual-band application of next-generation smartphones is developed. Every single antenna in the array is placed on the borders of a smartphone’s PCB, operating within the 3.55 GHz to 3.7 GHz [...] Read more.
A novel ground slot-based massive multiple-input multiple-output (MIMO) system with 14 elements for dual-band application of next-generation smartphones is developed. Every single antenna in the array is placed on the borders of a smartphone’s PCB, operating within the 3.55 GHz to 3.7 GHz (n48) and 5.855 GHz to 5.925 GHz (n47) frequency bands. This study also further assesses the performance of this system with analytical results, in which the self-s-parameters are less than −6 dB for each antenna, whereas isolation between components is less than −10 dB. At the same time, the overall efficiency reaches 65.55% and higher in the whole band. The evaluation of MIMO performance parameters such as envelope correlation coefficient (ECC), and DG of the MIMO system is well performed. A proposed MIMO antenna system is deployed and analyzed, which is a massive MIMO antenna, and the simulation results are compared to the measurement. The results validate the fact that the design of this particular MIMO system suits it to be incorporated in the n47 and n48 frequency bands, and therefore the presented MIMO system is suitable to be an option for integration into the upcoming smartphone architecture. Full article
(This article belongs to the Section Microwave and Wireless Communications)
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<p>Single antenna element. (<b>a</b>) Front view [left]. (<b>b</b>) Rear view [center]. (<b>c</b>) Reflection coefficient of single antenna element [right].</p>
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<p>Current distribution of single antenna element. (<b>a</b>) For n48 [left]. (<b>b</b>) For n47 [right].</p>
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<p>Ground slot length analysis of single antenna element. (<b>a</b>) For n48 [left]. (<b>b</b>) For n47 [right].</p>
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<p>Ground slot width analysis of single antenna element (<b>a</b>) For n48 [left] (<b>b</b>) For n47 [right].</p>
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<p>Fourteen element MIMO antenna design. Top view [left], and bottom view [right].</p>
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<p>Simulated and measured reflection coefficients of n48 frequency band.</p>
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<p>Simulated and measured reflection coefficients of n47 frequency band.</p>
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<p>Simulated transmission coefficients of for dual-band proposed MIMO.</p>
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<p>Simulated ECC for dual-band MIMO antennas.</p>
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<p>Simulated and measured DG of n48 frequency band.</p>
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<p>Simulated and measured DG of n47 frequency band.</p>
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<p>Simulatio of 14-antenna massive MIMO system integrated with human parts. (<b>a</b>) Two hands. (<b>b</b>) Head with hand. (<b>c</b>) Only head.</p>
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<p>Simulated reflection and transmission coefficients for a human model of n48 frequency band.</p>
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<p>Simulated reflection and transmission coefficients for a human model of n47 frequency band.</p>
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<p>SAR values on all the ports for n48 frequency band.</p>
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<p>Fabricated prototype. (<b>a</b>) Front view [left]. (<b>b</b>) Rear view [center]. (<b>c</b>) Measurement setup [right].</p>
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<p>Simulated and measured e- (red lines) and h-planes (blue lines) for n48 frequency band.</p>
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<p>Simulated and measured e- (red lines) and h-planes (blue lines) for n47 frequency band.</p>
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17 pages, 34163 KiB  
Article
Analysis of 3D Printed Dielectric Resonator Antenna Arrays for Millimeter-Wave 5G Applications
by Siyu Li, Benito Sanz Izquierdo, Steven Gao and Zhijiao Chen
Appl. Sci. 2024, 14(21), 9886; https://doi.org/10.3390/app14219886 - 29 Oct 2024
Cited by 1 | Viewed by 834
Abstract
This paper explores the potential use of fused deposition modeling (FDM) technology for manufacturing microwave and millimeter-wave dielectric resonator antennas (DRAs) for 5G and beyond communication systems. DRAs operating at microwave and millimeter-wave (mmWave) frequency bands were simulated, fabricated, and analyzed in terms [...] Read more.
This paper explores the potential use of fused deposition modeling (FDM) technology for manufacturing microwave and millimeter-wave dielectric resonator antennas (DRAs) for 5G and beyond communication systems. DRAs operating at microwave and millimeter-wave (mmWave) frequency bands were simulated, fabricated, and analyzed in terms of manufacturing quality and radio frequency (RF) performance. Samples were manufactured using a 3D printer and PREPERM® ABS1000 filament, which offers a stable dielectric constant (εr = 10 ± 0.35) and low losses (tan δ = 0.003) over wide frequency and temperature ranges. Surface profile tests and microscope measurements revealed discrepancies in the dimensions in the xy-plane and along the z-axis, consistent with the observed shift in resonant frequency. Despite these variations, reasonably good agreement between RF-simulated and measured results was achieved, and the DRA array successfully covered the intended mmWave band. However, challenges in achieving high precision may restrict applications at higher mmWave bands. Nevertheless, compared with conventional methods, FDM techniques offer a highly accessible and flexible solution with a wide range of materials for home and micro-manufacturing of mmWave DRAs for modern 5G systems. Full article
(This article belongs to the Special Issue 5G and Beyond: Technologies and Communications)
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<p>Configuration of the FDM printed microwave DRA: (<b>a</b>) DRA on top of a metallic ground plane, (<b>b</b>) substrate and DRA made transparent to show the microstrip transmission line on the back of the substrate and the slot on the top ground plane.</p>
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<p>Zoom-in view of the top and bottom surfaces of the FDM-printed microwave DRA. (<b>a</b>) top view, (<b>b</b>) bottom view.</p>
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<p>2D cut surface profile of the FDM-printed DRA.</p>
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<p>3D side view and 2D cut surface profile of the FDM-printed DRA.</p>
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<p>Fabricated DRA. (<b>a</b>) top view, (<b>b</b>) bottom view.</p>
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<p>Simulated and measured <span class="html-italic">S</span><sub>11</sub> of the microwave DRA design. Adjusted dimensional parameters and permittivity were also taken into account based on the measured values.</p>
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<p>Configuration of the mmWave DRA array.</p>
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<p>Simulated efficiency and realized gain for mmWave DRA array.</p>
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<p>Beam scanning performance of the DRA array in (<b>a</b>) xz- and (<b>b</b>) yz-plane.</p>
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<p>Gain at broadside direction with different (<b>a</b>) radius, (<b>b</b>) height, (<b>c</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>ε</mi> </mrow> <mrow> <mi>r</mi> </mrow> </msub> </mrow> </semantics></math>, (<b>d</b>) tan <span class="html-italic">δ</span> of DRA element.</p>
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<p>Efficiency with different (<b>a</b>) radius, (<b>b</b>) height, (<b>c</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>ε</mi> </mrow> <mrow> <mi>r</mi> </mrow> </msub> </mrow> </semantics></math>, (<b>d</b>) tan <span class="html-italic">δ</span> of DRA element.</p>
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<p>Configuration of the DRA array with an in-phase feeding network.</p>
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<p>Photographs of the FDM 3D-printed DRA array with an in-phase feeding network. (<b>a</b>) top view, (<b>b</b>) bottom view.</p>
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<p>Zoom-in view of the top and bottom surfaces of the FDM 3D-printed mmWave DRA. (<b>a</b>) top view, (<b>b</b>) bottom view.</p>
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<p>2D cut surface profile of the FDM-printed mmWave DRA.</p>
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<p>3D side view and 2D cut surface profile of the FDM-printed mmWave DRA element.</p>
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<p>Simulated and measured matching performance of the FDM 3D-printed DRA array with in-phase feeding network.</p>
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<p>Simulated and measured radiation patterns of the FDM 3D-printed DRA array at 26.8 GHz. (<b>a</b>) xz- and (<b>b</b>) yz-plane.</p>
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<p>Simulated and measured realized gain at the broadside direction of the FDM 3D-printed DRA array.</p>
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<p>(<b>a</b>) Configuration of the DRA array, and (<b>b</b>) simulated efficiency and realized gain.</p>
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<p>Beam scanning performance of the cuboid DRA array in the (<b>a</b>) xz-, and (<b>b</b>) yz-planes.</p>
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<p>Top view of the FDM 3D-printed cuboid DRA element (<b>a</b>) before, and (<b>b</b>) after refining.</p>
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<p>Side view of the FDM 3D-printed cuboid DRA element (<b>a</b>) before, and (<b>b</b>) after refining.</p>
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